CONTROL AND DYNAMIC SYSTEMS
Advances in Theory and Applications Volume 74
CONTROL AND DYNAMIC SYSTEMS
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CONTROL AND DYNAMIC SYSTEMS
Advances in Theory and Applications Volume 74
CONTROL AND DYNAMIC SYSTEMS
Advances in Theory and Applications Volume 74
CONTRIBUTORS TO THIS VOLUME DE ABRE U-GAR CIA J. CHEN M UNTHER A. DAHLEH SARIT K. DAS ANTHONY N. MICHEL AHMAD A. MOHAMMED YAS UHIKO MUTOH PETER N. NIKIFORUK R. J. PATTON MILOJE S. RADENKOVIC P. K. RAJAGOPALAN TIELONG SHEN ROBERT E. SKELTON PETROS G. VOULGARIS G UOMING G. ZHU
J. A .
CONTROL AND DYNAMIC SYSq'EMS ADVANCES IN THEORY AND APPLICATIONS
Edited by
CORNELIUS T. LEONDES School of Engineering and Applied Science University of California, Los Angeles Los Angeles, California
VOLUME 74:
TECHNIQUES IN DISCRETE AND CONTINUOUS ROBUST SYSTEMS
ACADEMIC PRESS San Diego New York Boston London Sydney Tokyo Toronto
This book is printed on acid-free paper. ( ~ Copyright 9 1996 by ACADEMIC PRESS, INC. All Rights Reserved. No part of this publication may be reproduced or transmitted in any form or by any means, electronic or mechanical, including photocopy, recording, or any information storage and retrieval system, without permission in writing from the publisher.
Academic Press, Inc. A Division of Harcourt Brace & Company 525 B Street, Suite 1900, San Diego, California 92101-4495
United Kingdom Edition published by Academic Press Limited 24-28 Oval Road, London NW1 7DX
International Standard Serial Number: 0090-5267 International Standard Book Number: 0-12-012774-1
PRINTED IN THE UNITED STATES OF AMERICA 96 97 98 99 00 01 QW 9 8 7 6 5
4
3
2
1
CONTENTS
CONTRIBUTORS .................................................................................. PREFACE ................................................................................................
vii ix
Optimal and Robust Controllers for Periodic and Multirate Systems ......................................................................................
Petros G. Voulgaris and Munther A. Dahleh ................................
59
A Two-Riccati, Feasible Algorithm for Guaranteeing Output L~ Constraints ............................................................................
97
Discrete-Time Robust Adaptive Control Systems
Miloje S. Radenkovic and Anthony N. Michel
Guoming G. Zhu and Robert E. Skelton Techniques of Analysis and Robust Control via Zero-Placement of Periodically Compensated Discrete-Time Plants ................................... 133
Sarit K. Das and P. K. Rajagopalan Robust Fault Detection and Isolation (FDI) Systems
........................... 171
R. J. Patton and J. Chen Absolute Stability of Discrete Nonlinear Feedback Systems
............... 225
Yashuhiko Mutoh, Tielong Shen, and Peter N. Nikiforuk
vi
CONTENTS
Continuous Time and Discrete Time L y a p u n o v Equations" R e v i e w and N e w Directions ................................................................................ 253
Ahmad A. Mohammed and J. A. De Abreu-Garcia
I N D E X ..................................................................................................... 305
CONTRIBUTORS
Numbers in parentheses indicatethe pages on which the authors' contributions begin. J. A. De Abreu-Garcia (253), Department of Electrical Engineering, The
University of Akron, Akron, Ohio 44325 J. Chen (171), Division of Dynamics and Control, University of Strathclyde,
Glasgow, Gl lXJ, United Kingdom Munther A. Dahleh (1), Laboratory of Information and Decision Systems,
Massachusetts Institute of Technology, Cambridge, Massachusetts 02139 Sarit K. Das (133), Department of Electrical Engineering, Indian Institute
of Technology, Kharagpur 721302, India Anthony N. Michel (59), Department of Electrical Engineering, University
of Notre Dame, Notre Dame, Indiana 46556 Ahmad A. M o h a m m e d (253), Department of Electrical Engineering, The
University of Akron, Akron, Ohio 44325 Yasuhiko Mutoh (225), Department of Mechanical Engineering, Sophia Uni-
versity, Tokyo, Chiyoda-ku, Japan Peter N. Nikiforuk (225), Department of Mechanical Engineering, Univer-
sity of Saskatchewan, Saskatoon, Saskatchewan, Canada S7N OWO R. J. Patton (171), Department of Electronic Engineering, University of Hull,
Hull HU6 7RX, United Kingdom Miloje S. Radenkovic (59), Department of Electrical Engineering, Univer-
sity of Colorado at Denver, Denver, Colorado 80217 P. K. Rajagopalan (133), Department of Electrical Engineering, Indian In-
stitute of Technology, Kharagpur 721302, India vii
viii
CONTRIBUTORS
Tielong Shen (225), Department of Electrical Engineering, Sophia University, Tokyo, Chiyoda-ku, Japan Robert E. Skelton (97), Space Systems Control Laboratory, Purdue University, West Lafayette, Indiana 47907 Petros G. Voulgaris (1), Coordinated Science Laboratory, University of Illinois at Urbana-Champaign, Urbana, Illinois 61801 Guoming G. Zhu (97), Cummins Engine Company, Inc., Columbus, Indiana 47202
PREFACE Effective control concepts and applications date back over millennia. One very familiar example of this is the windmill. It was designed to derive maximum benefit from windflow, a simple but highly effective optimization technique. Harold Hazen's 1932 paper in the Journal of the Franklin Institute was one of the earlier reference points wherein an analytical framework for modem control theory was established. There were many other notable items along the way, including the MIT Radiation Laboratory series volume on servomechanisms, the Brown and Campbell book, Principles of Servomechanisms, and Bode's book, Network Analysis and Synthesis Techniques, all published shortly after mid-1945. However, it remained for Kalman's papers of the late 1950s (wherein a foundation for modem state-space techniques was established) and the tremendous evolution of digital computer technology (which was founded on the continuous giant advances in integrated electronics) to establish truly powerful control systems techniques for increasingly complex systems to be developed. Today we can look forward to a future that is rich in possibilities in many systems of major significance: manufacturing, electric power, robotics, aerospace, and others with significant economic, safety, cost, and reliability implications. Thus, this volume is devoted to the most timely theme of "Techniques in Discrete and Continuous Robust Systems." The first contribution to this volume is "Optimal and Robust Controllers for Periodic and Multirate Systems," by Petros G. Voulgaris and Munther A. Dahleh. The problems of optimal disturbance rejection and robust stability in periodic and multirate systems are treated in this contribution, and techniques for dealing with these in several major cases are presented. This contribution further notes that the problem of robust stabilization in periodic and multirate systems can be treated without introducing conservatism by considering the problem for the equivalent linear time invariant (LTI) system. Other important results and techniques are also discussed, and as such it is a most appropriate chapter to begin this volume. The second contribution is "Discrete-Time Robust Adaptive Control Systems," by Miloje S. Radenkovic and Anthony N. Michel. The robust adaptive control problem has been the topic of numerous studies, publications,
x
PREFACE
and discussions over the past decade, and it has been observed that algorithms designed for the case of perfect system models cannot provide global stability of the adaptive system in the presence of unmodeled dynamics and external disturbances. Techniques are presented for establishing the global stability of adaptive systems for unmodeled system dynamics and external disturbances which are unstructured, complex, and large at high frequencies. It is worth noting that results similar to those presented in this contribution can be established for the general delay case. The next contribution is "A Two-Riccati, Feasible Algorithm for Guaranteeing Output L~ Constraints," by Guoming G. Zhu and Robert E. Skelton. This contribution presents a rather powerful new and computationally efficient methodology for designing measurement or dynamic controllers that guarantee requisite output bounds for measures of bounded input disturbances while minimizing a weighted summation of measures of the upper bound of the outputs for each control channel. The substantial effectiveness of the results presented in this contribution is illustrated by several examples. Techniques for the design of a linear, discrete periodic controller structure that has the maximum possible degrees of freedom for its order are presented in "Techniques of Analysis and Robust Control via Zero-Placement of Periodically Compensated Discrete-Time Plants," by Sarit K. Das and R. K. Rajagopalan. Analytical methods for dealing with periodic systems and the zero placement capability of periodic controllers are also presented. The role of periodic controllers in gain margin improvement of unstable plants with nonminimum phase zeros via the zero placement approach and periodic controllers is treated. Numerous examples demonstrate the effectiveness of the techniques. "Robust Fault Detection and Isolation (FDI) Systems," by R. J. Patton and J. Chen, is an in-depth treatment of the techniques in the field of robustness for fault diagnosis that rest on the model-based residual generation methods (which are defined in this contribution). This approach to fault detection and isolation offers rather significant potential, particularly in view of the role it can and probably will play in future industrial systems, process plants, and other applications of major significance. The next contribution, "Absolute Stability of Discrete Nonlinear Feedback Systems," by Yasuhiko Mutoh, Tielong Shen, and Peter N. Nikiforuk, presents various approaches for the establishment of the stability of discretetime nonlinear feedback systems. Techniques are presented for both singleinput-single-output (SISO) systems and multiple-input-multiple-output (MIMO) systems. Examples illustrate the utility of the methods presented. The final contribution to this volume is "Continuous Time and Discrete Time Lyapunov Equations: Review and New Directions," by Ahmad A. Mohammad and J. A. De Abreu-Garcia. The fundamental reasons for robust design techniques include among other factors the maintenance of satisfactory system performance in the face of model uncertainties and, perhaps more
PREFACE
xi
importantly, the maintenance of system stability. This contribution is an indepth treatment of one of the most powerful means for confirming stability for both linear and nonlinear systems. As such this is a most suitable contribution with which to conclude this volume. The contributors to this volume are all to be highly commended for their contributions to this comprehensive treatment of techniques in discrete and continuous robust systems. They have produced a work which should provide a unique and useful reference on this broad subject internationally for years to come.
This Page Intentionally Left Blank
Optimal and Robust Controllers for Periodic and Multirate Systems Petros G. Voulgaris Coordinated Science Laboratory University of Illinois at Urb~na-Chmmpaign
Munther A. Dahleh L a b o r a t o r y of I n f o r m a t i o n ~nd D e c i s i o n S y s t e m s
Massachusetts Institute of Technology
I. I N T R O D U C T I O N The study of periodically time varying systems is a topic great practical and theoretical importance. In [13] an equivalence between m-input, p-output, linear, N-periodic, causal, discrete-time systems and a class of discrete-time linear, time invariant, causal systems was established. Namely, this class consists of ruN,input, pN-output, linear time invariant (LTI) systems with A- transforms P(A) such that P(0) is a block lower triangular matrix. This equivalence is strong in the sense that it preserves the algebraic structure (isomorphism) and the norm (isometry). This equivalence is termed "lifti n ~ and the LTI system that lifting associates with the N-periodic system is called the ~lifted" system. Hence, we can effectively use the theory of LTI systems to study periodic ones. In fact, the authors in [13] use this equivalence to prove that although the performance is not improved , periodic compensators for LTI plants'offer significant advantages in terms of robustness to parametric uncertainty. In this chapter, we define the problem of optimal disturbance rejection in periodic systems and present solutions to the following three cases: I. Optimal L~~ to t~176disturbance rejection. 2. Optimal t2 to t2 disturbance rejection. 3. Optimal rejection of stochastic disturbances (the L Q G problem). Utilizing the power of lifting and the results in [3] we can easily infere that in all three cases the optimal controller for the N-periodic system can be obtained by solving the equivalent LTI problem. This problem however, includes a constraint on the optimal LTI compensator C(A), namely CONTROL AND DYNAMIC SYSTEMS, VOL. 74 Copyright 9 1996 by Academic Press, Inc. All rights of reproduction in any form reserved.
2
PETROS G. V O U L G A R I S A N D M U N T H E R A. D A H L E H
C(0) should be block lower triangular matrix so that C corresponds to a causal N-periodic controller. This constraint makes the optimization problem a non-standard one and its solution is the theme of sections IV and V. Furthermore in section VI we demonstrate that the problem of optimal disturbance rejection for multirate sampled systems can be treated analogously. In particular, we show how with a simple modification the same approach can be used to obtain the optimal multirate compensator. Also in this chapter, we consider the problem of robust stabilization in periodic and multirate plants. We indicate that this problem can be analyzed without introducing conservatism by considering the same problem for the equivalent LTI system.
II. M A T H E M A T I C A L
PRELIMINARIES
This section presents the notation and definitions to be used throughout the chapter. Also, some important to our development mathematical results are provided. References are given to cover all of the needed mathematical background.
1
Generic
Notation
In this section we give some generic notation that is used throughtout the thesis.
p(A)
The spectral radius of the matrix A.
~[A]
The maximum singular value of the matrix A.
Izlp
The p-norm of the finite dimensional vector z = (zl z~ . . . z , ) T given as
Ixl,, = ()":~ Ix, I) ~/', p < oo i=l
Izlp = m.,~ Ix, I, p = oo. $
Ial~
The 1-norm of the m x n matrix A = (Aij) given as
FJ
IAII= /t(A)
max
i--- 1,...,m
~'~IA,#Ij=0
The A-transform of a m x n real sequence H = {H(k)}~~ defined as: oo
9(~) = ~
k~---oo
#(k)~ ~
CONTROLLERS FOR PERIODIC AND MULTIRATE SYSTEMS
X
I
The dual space of the normed linear space X.
BX
The closed unit ball of X.
•
The left annihilator of S C X*.
S•
The right annihilator of S C X.
IIs
The projection operator onto the subset S of the Hilbert space X.
(Z~
T*
n~
Z*)
The value of the bounded linear functional z" at point z ~ X. The adjoint of the operator T. The kth-truncation operator acting on a m x 1
~e~tor~u~d ~equen~ {~(~)}~~ a~ Hkm 9{u(0), u(1),...}
Am
: {u(0),..., u(k), 0, 0,...}.
The right shift operator acting on a m x 1 vector valued sequence {u(k) } ~~ as
A m : {u(0), u(1),...}
2
3
Some
, {0, u(0), u(1),...}.
Basic Spaces
In this section we define certain important normed linear spaces that we very frequently refer to in the course of our development. These spaces are the following (for details look at [22,16,14,28]):
l~x" 9The Banach space of all m x n matrices H each of whose entries is a right sided, absolutely summable real sequence Hij = {Hij (k) }~~ The norm is defined as:
IIHII~,
mXs
n dell" =
max E i
oo
~ IHij(t)l.
j=l k=0
: The Banach space of real rn x 1 vectors u each of whose components is a magnitude bounded reM sequence { u , ( k ) } r : 0. The norm is defined as:
II~llt: d.j miax(s~pI~,(k)l).
4
PETROS G. VOULGARIS AND MUNTHER A. DAHLEH
~~ 9The extended t~~ space: it is the space of all real right sided m x 1 vector valued sequences.
t ~ 9 The Hilbert space of real rn • 1 vectors u each of whose components is an energy bounded real sequence {u~ (h) }~~ The norm is defined as: m
oo
i=1 k=O
x , " The real Banach space of all m x n matrices H(A) such t h a t H(A) is the A-transform of an I x , x , sequence H. The norm is defined as
This space is isometrically isomorphic to lx.,x, i.e., l ~ x . _-- . A . , x . .
.A~x n 9 The Banach space of all m x n matrices H each of whose entries is a right sided, magnitude bounded real sequence Hij = {Hij(k))~~ The n o r m is defined as: m
IIHII.A:.,,..
def
max(sup IH,i(k)l).
=
dfX
cox. zero.
9The subspace of ~ x .
J
k
consisting of all elements which converge to
O0
s " The Banach space of all m x . m a t r i x valued functions F defined on the unit circle of the complex plain with def
I]F][s = ess sup ~'[F(eJ')]< oo. ,e[0,~.]
oo
7~mx. 9 The Banach space of all rn x n matrix valued functions F analytic in the open unit disk of the complex plain with def
IIFII - =
sup
max ~ [ F ( r e J ' ) ] < oo.
rE[O,1) #El0,2-]
This space can be considered as a closed subspace of s
oo
CONTROLLERS FOR PERIODIC AND MULTIRATE SYSTEMS
5
s 2 1 5 " The Hilbert space of matrix valued functions F defined on the unit circle of the complex plain with
IIFII~ def = [(2x) -1 fo 2" t r a c e ( F T ( e - { ' ) F ( e ~ ' ) ) d e ]
~/~ < oo.
~/~xn " The Hilbert space of all m x n matrix valued functions F analytic in the open unit disk of the complex plane with def
sup [(2r -1 JO02r t r a c e ( F T ( r e - i ' ) F ( r e ~ ' ) ) d O ] ~/2 < oo. re[o,1)
This space can be considered as a closed subspace of s
t 2 = ~2,,1. 3
Input-Output Systems
Characterization
Moreover,
of Linear
In this section we consider the input-output characterization of systems by viewing them as linear operators. 3.1
Causality
We start with the notion of a causal operator D e f i n i t i o n 3.1 L e t T : l~ ,e
n Tu =
be an operator. T is called causal i f
, ~,e
ThOu,
vk = o,
2,...,
T is called s t r i c t l y causal i f II ~ T u = IIk - m -T- InI k - I u,
V k = O , 1,2, . . . .
The class of all causal operators T on/oo,e will be denoted by s215 Such operators can be represented by infite block lower triangular matrices (Toeplitz) of the form
To(O) o "
i
""/
6
PETROS G. V O U L G A R I S AND M U N T H E R A. D A H L E H
where Tk(i) axe m • n matrices for all i, k. This representation is another way to state that these operators are convolution operators; i.e., if y = Tu then k
~(k)
= ~
n(k
-
i)~(~)
i----0
3.2 3.2.1
Stability T h e spaces ~ " ( P ) ,
~"
(t 2)
Next, we consider the notion of/a-stability where ~ = or, 2. D e f i n i t i o n 3.2 Let T be a causa/operator in s if its induced norm over t a is bounded; i.e., i f IITII
=
sup ~,:.~,o Ilulll:.
Then T is ~ - s t a b l e
< oo.
The clase of all/a-stable systems equipped with the induced norm, will be denoted by 15~T~"(t~). This class is s Banach space and in paxticulax it is a Banaeh algebra with multiplication defined as composition. In the cane where a = oo we refer to the space B~T~n(t ~176as the space of Bounded-Input-Bounded-Output (BIBO) stable, or simply, stable systems. Moreover, the following fact can be easily checked. nite rn • n block lower triangular matrices of the form
To(O) 0 ---/ r~(X) T~(O) 9
i
where Tk (i) are rn • n matrices for all i, k such that
,up I(n(k) n(k-
x) . . . n ( 0 ) ) l ~
= As in the proof of Theorem 1.1, let
TsT~ be the operator
on t 1 associated
with the stable system Si-1 and let T~si., be its adjoint on c ~ Then by defining
x = ~ r , (D) we can show that Hence ($I ~Q, X) = 0 if and only if (Q, F) = 0 which completes the proof. The functionals Xy, j = I,...,J-5 obtained by the second method will contain finitelym a n y nonzero components. Namely, Xj(m) = 0 V m > k. This is so because Fj has all its elements but the first equal to zero and hence Dy(m) equals zero for m > k. In the first method however, the obtained functionals will not, in general, have this property. They will decay though, (since they lie in co) and the rate of decay will be dictated by the poles of Ui and ~ . Hence, it seems that the linear programming problem in the dual space [5] will be simpler when the second method is used, provided that a simple way to obtain a Smith decomposition of a transfer matrix exists.
28
1.3
PETROS G. VOULGARISAND MUNTHERA. DAHLEH
Examples
Here we show that in general (OPToo) has different solution than the unconstrained problem. Example 1.1 Consider an optimization problem in a 2-periodic singleinput single-output system with the equivalent LTI problem being as follows; inf IIS - vQII
"-A/
QEll
/ 1 0 / U(A)--/A such that Q(0) is lower triangular, where/t(,~) = 0 2 ' 0 1 We first solve the unconstrained problem following [5]" The basis for the functionals that annihilate S, = {UQ "Q E t 1} consists of the following: F~={
(10) 0 0
F 2 = { l 0o
The resulting optimal solution
0
..
01 ) , 0 ,
..
~. = H-
.}
U Q . is:
(1 0)
where @12,4)22 arbitrary in t I such that II~x211,, + 114,2211,, __< 1
and
Qu= /--~b21 2--~22 / -~21
2 - ~22
Also, I1+.11~, = 1.
Now, in the constrained case, we obtain by using the second approach the following extra functional: F 3 = { I 00 01 1 , I 00 011 ,0,...} The resulting optimal solution is: ~(~)=
I I .5~ 1 0 1.5
"
CONTROLLERS FOR PERIODIC AND MULTIRATE SYSTEMS
with
Q(~)=
Also,
(oo) o .5
ll x then IIAGII -- IIGll > 1 and hence there exists [23] a Ax E 7/~ with II~xll < x such that the pair (AG, AI) in feedback is unstable. But then the pair ( G , A ~ is unstable with A ~ = AxA; mo~eove~ ~in~e I1~~ = II~xll < 1 and also A ~ is strictly proper then by Fact 1.2 we conclude that if IIGpII > 1 there is a destabilizing A~ E/)p with A~ being the inverse lift of A ~ In the discussion above, Gp will depend in general on the particular controller C~ we pick for the nominal unperturbed system. Hence, if we pertain to periodic controllers (i.e., periodic
CONTROLLERS FOR PERIODIC AND MULTIRATE SYSTEMS
1 |
,
|
W ~
! !
, ,,,,
I
! |
:Gp !
.
P,
v
|
!
| I |
| | | | | o I
1
..
| | ! ~
.
,......................................., Figure 4: Robust Stabilization. maps Gp(C~)), then in order to maximize the tolerance to perturbations in the class D the problem we have to solve is inf
C , periodic
[[Gp(C,)[[
which is a performance problem in periodic systems that can be solved with the method indicated in the previous sections. At this point, we do not know whether a general time varying Cv can give better margins than a periodic with respect to this particular class ~Dp of periodic perturbations. However, if we enlarge the class of perturbations to include generally time varying ones as then we have the following necessary and sufficient condition: T h e o r e m 2.1 The following statements are equivalent:
(i) There exists a robustly stabilizing time varying controller with respect to perturbations in l)u. (ii) There ex/sts a robustly stabilizing periodic controller with respect to perturbations in ~)~.
OiO inf
C. periodic
IlGp(C )ll "< 1.
41
42
PETROS G. VOULGARIS AND MUNTHER A. DAHLEH
Proof The (ii)=~(i)is obvious. For (i)::~(ii)consider the lifted feedback system ( G ( C ) , A) where C is the lifted controller; note that G, C, A are not shift invariant, nevertheless, all should posess a block lower triangular feedforward term. To be more specific, consider for example the relation z - Gv; then, the operation of G on v can be represented as the convolution z ( t ) - ~~176 g(t, r)v(r). G has a block lower triangular feedforward term if g(t, 0) is block lower triangular for all t. The same interpretation is valid for A, C. Now following [25] this implies that infk [[GAk[[ _< 1 for, otherwise, a strictly proper A, and hence a legitimate A~ E l)v, exists that destabilizes the system. But then, following the same arguments as in theorem 4.1 of [24] we can check that there is a controller C1 - A-t~CA t~ for some large enough integer kl, such that ]]G(C~)[] _< 1. Note that Cx has a block lower triangular feedforward term since C does and therefore it corresponds to a causal (not necessarily periodic) C1~. But then from Fact 1.4 there is a periodic controller Cp such that [[Gp(Cp)[I < 1 and therefore Cp will robustly stabilize the system for all perturbations in ~ . The (ii)r follows from the analysis of periodic perturbations in ~)p. | Hence, if we wish to maximize the tolerance to perturbations in the class/)~ the problem we have to solve is
inf
C. periodic
[[Gp(C~)] I
which is a performance problem in periodic systems that can be solved with the method indicated in section 4. Finally, we should note that the exactly the same analysis holds when we encounter the robustness problem for multirate systems.
3
Section S u m m a r y
In this section we presented the solutions to the optimal s2 to 12 disturbance rejection problem (7/~176as well as the solution to the L Q G (7/2) problem in periodic systems using the lifting technique. Both problems involved a causality condition on the optimal LTI compensator when viewed in the lifted domain. The 7/~176 problem was solved using the Nehari's theorem whereas in the 7i2 problem the solution was obtained using the Projection theorem. In particular, the 7~~176problem was solved by modifying the standard Nehari's approach in order to account for the additional causality constraint on the compensator. This modification yielded a finite dimensional convex optimization problem over a convex set that needs to be solved before applying the standard solution to the Nehari problem. The solution to the above convex finite dimensional problem can be obtained easily using standard programming techniques. In the 7g2 case the solution was obtained from the optimal (standard) unconstrained problem by projecting only the feedforward term of the standard solution to the allowable
CONTROLLERS FOR PERIODIC AND MULTIRATE SYSTEMS
43
Yl
Ul
K1T
L1T
K,~T
LpT
Um v
Figure 5" General Multirate System. subspace. Finally, we indicated that the small gain condition is necessary as well as sufficient for 12-stability in periodic and multirate systems and that general time varying compensation offers no advantage over periodic as far as robust stability to time varying perturbations is concerned.
VI. M U L T I R A T E SYSTEMS" O P T I M A L D I S T U R B A N C E REJECTION AND ROBUST STABILITY In this section, we utilize the methods of solutions to the problem of optimal disturbance rejection in periodic systems that we presented in sections VI and V in order to solve the same problem in the case of multirate systems. The most general configuration of a multirate system is depicted in Figure 5. In this figure each of the input sequences {uj (k) }~~ enters the system at every Tu~ seconds which might be different from, say, Tuj+l which is the rate of the input sequence {u~ +1 (k) }~~ Also, {yi (k) }~~ leaves the system at a rate of Ty~ seconds which might be different from the rates of the rest of the output and/or input sequences. A typical example of a multirate system is the multirate sampled data system (MRSD) of Figure 6 where not all of the input and output sequences are held and sampled at the same rate. In the case where not all of the rates of the input and output sequences are rational multiples of each other i.e., the sequences are not sychronized the problem of designing the controller is hard. In the case however where the rates are sychronized (i.e., any ratio of rates is rational) then the synthesis problem becomes considerably easier. In particular, lifting techniques can be used to transform a multirate problem to a LTI problem as with the case of periodic systems. Again, this lifting induces a causality constraint on the lifted controller which can be handled exactly with the same manner as in the periodic case. In the subsequent sections we give the details of
44
PETROS G. V O U L G A R | S A N D M U N T H E R A. D A H L E H
Ul . ~ ~
Uc I
Ycl
. . . .
Yl v
L~T K~T Um
G0(,)
o
Uc,~
Yc~
-I!-iii-.......
!
w
LpT
K, n T
Figure 6: General Multirate Sampled Data System. how this is done.
Lifting and Shift Invariant Equivalence Throughout this section we will deal only with the sychronous case. This means that for the system of Figure 5 the rates of all the input and output sequences can be written as integer multiples of some common time period T. In particular we have Tu~ = KiT, j = 1 , . . . , m and Ty, = L~T, i = 1 , . . . , p where we assume that the integers (L1,..., Lp, K1,...K,~) are relatively prime. The period T is called the basic time period (BTP). We will call a multirate system causal if the value of yi(k) Vi = 1 , . . . , m does not depend on any uj(l) with I > kL~/Kj Vj = 1 , . . . , p . For the multirate system of Figure 5 let N denote the least common multiple of the relatively prime integers (Li, L 2 , . . . , Lp, K1, K 2 , . . . , Kin). Also let Pi - ( N / L , ) , i -- 1 , . . . , p , Mj -- ( N / Kj ), j - 1 , . . . , m and P ~ = 1 Pi, M - ~ j = l Mj. In the sequel we introduce s lifting isomorphism [18,21] similar to the one in the periodic case, that transforms them to causal "single rate" systems. m
1.1
Lifting in multirate
systems
Before we proceed we give the following definition: Defin[tion 1.1 Given the set of integers {My )~--I, (Pi}~'=x and a P x M matrix D partitioned as S~--"
I D11D12... D1ml .
.
.
.
.
.
.
CONTROLLERSFOR PERIODICAND MULTIRATESYSTEMS
45
where each Dij is a Pi x Mj matrix we say that D satisfies the (P i, Mj) causality conditions if (D,i)o~ = 0 when (a-l) N
P~
N < O; -(/3- 1)~-j
l 1,0 < / J < 2, with all variables having finite initial conditions.
(6)
DISCRETE-TIME ROBUST ADAPTIVE CONTROL SYSTEMS
63
Regarding relation (6) we introduce the following assumptions" (A1) V(t) 6 ~+ and V(t) < do < co, for all t > 1. (A2) "Exponentially growing rule" - the sequence n,(t) E ~+ can not grow faster than exponentially, i.e., for all t _> 1
n,(t) < Con,(t - 1) + ko + ~l(t), 1 < Co < co, 0 < ko < co
(7)
where n,(t) is defined in terms of e(t), i.e., t
.o(~)- ~ ~'-J~(j + 1) ~
(8)
j=l
for a fixed 0 < A < 1. (A3) nc~(t) E ~+ and for t > 1 (i) n#(t) > An#(t- 1), no(I) r 0, and n#(0) > 0, 0 < A < 1 (ii) no(t) < Cr ne(t)+ C#~ + ~2(t), 0 < C#1, Cr < co
(9) (10)
(A4) Sequence 7(1) E @~and depends on the sequence e(t), i.e., A'-JT(j) 2
< C~
j=l
A'-Je(j + 1) 2
+ k, + ~3(t) (11)
j=l
where 0 _< C.y,k~ < co, 0 < A < I.
(As) On the subsequence at, k = I, 2, 3,..., where S(a~ -4-I) > 0 and S(ak) 0 for all t > 1. Using the definition of S(t) (Eq. (13)) and assumption (A3)(ii), from (6) one obtains
V(t + 1)+ S(t + 1) n,(t)
S(t) < v(t) + n4,(t- 1) -/~Pl
e(t + 1) 2 nr
(16)
where pl is given by assumption (A6). After summation from $ = 1 to N, from the above relation we obtain
~(t+ 1) 2
lim /~p~
S(1)
< V ( 1 ) + ,,~(0) < oo
(17)
~--1
wherefrom it follows that lim e(t + 1) 2
*-~
nr
=0
(~8)
DISCRETE-TIME ROBUST ADAPTIVE CONTROL SYSTEMS
65
Since assumption (A3)(ii) implies
n,(t) -< g'~' A l f2 i = 1,2,3
(65)
where 7 and ,~ are the same as in (61). For frequencies close to f~, conditions (64) and (65) may be difficult to achieve, since we need a filter with short transient band. In practice this is the matter of the designer skills and the specific system that has to be controlled. In our considerations we will assume that (64) and (65) are satisfied. We will also assume that W(z) is a stable polynomial. Specifically, if we design a filter with amplitude characteristics described by Eq. (64) and (65) and some of its zeros appear to be unstable, we will use a filter where the unstable zeros are replaced by their reciprocal values. These two filters have different phase characteristics but the same amplitude characteristics, and that is what we are interested in. Thus, the low-pass filter W(z) has the following form W(z -1) = w0 + wlz -1 + - " +
(66)
w, wz -"W
and we assume that (5'4) the zeros zj of t4Z(z) satisfy Izil _< )~1/2, where )~0 + )ix < )i < 1, and 0 < Ax < < ,~0 and ~0 is defined by assumption (Sx). We will stabilize the system (57) by designing the adaptive controller so that for a given reference signal y'(t), the following functional criterion is minimized J -
+ 1) - y ; ( t +
1)] +
2,
(sT)
where y/(t)-
ITV(q-1)y(t), y } ( t ) - I ~ ( q - t ) y ' ( t ) ,
(68)
while polynomials p(q-1) and Q ( q - t ) are chosen by the designer and are defined as follows
p(q-1)
=
l + ptq -1 + . . . + p,~pq-nP
Q(q-t)
=
qo + qtq - t + " ' ' + qnoq-n~
(69)
In (68), ~'V(q-1) is the low-pass FIR filter designed so that requirements (64) and (65) are satisfied and is given by Eq. (66). Using prior information related to the transfer function Be(z)/Ae(z) at low frequencies f < f2, we can choose polynomials p(q-1) and Q(q-1) so that the following assumption is satisfied:
DISCRETE-TIME ROBUST ADAPTIVE CONTROL SYSTEMS
73
(5'5) All zeros z1 of the polynomial
D(z) - Ae(z)Q(z) + So(z)bV(z)P(z) satisfy [zj[ __ 1. We assume that all initial conditions are finite. Note that the system model (57) can be written in the form
Ao(q-1)yf(t + 1) = Be(q-1)lTVu(t) + 7(t)
(70)
where "r(t)
=
[Be(q-1)A1(q-1)W1(q-1) + Ae(q-1)A2(q-l)w~.(q-l)]~V(q-1)u(t)
+
A3(q- 1) Wa(q-1 ) 1;k(q- 1)~(t)
(71)
It is not difficult to see that from (70) we can obtain e(t + 1)
where
-
[Bo(q-X)l:V(q -1) + Q(q-1)]u(t) - q[Ao(q-:) - 1]y1(t)
+ q[p(q-l)_ l]yl(t)_P(q-i)y;(t+ I)+ 7(t)
(72)
e(t + I) - P(q-1)[yl(t + I ) - y}(t + I)] + Q(q-l)u(t)
(73)
Equation (73) can be written in the form e(t + 1) = oTr
1]y/ (t) -- P(q-1)y~ (t + 1) + 7 ( t ) (74)
where
e0r -
;b0, h i , . . . , b..]
(75)
and
6(t) T = [yl(t),..., y l ( t - nA + 1); u l ( t ) , . . . , u l ( t - nB)]
(76)
u y ( t ) - ~V(q-1)u(t)
(77)
where From Eq. (74) it is obvious that when unmodelled dynamics and external disturbances are absent, control low optimal in the sense of (67) is given by 80Tr
+ Q(q-1)u(t ) + q[p(q-1) _ 1]y/(t) - P(q-1)y}(t + 1)
(78)
Based on the available prior information related to the system model at low frequencies f < f2, we introduce the following assumption:
74
MILOJE S. RADENKOVIC AND ANTHONY N. MICHEL
(ST) The compact convex set O ~ which contains the true parameters 60, the sign of r0 = bowo + qo and a lower bound r0,min, on the magnitude of r0 are known. Without loss of generality we assume that r0 > 0 and r0,min > 0. For the estimation of 00 we propose the following algorithm ~(t + 1) - 7~{8(t)+ n,(t)r
+ 1)}, 0 < v < 2
(79)
where e(t) is given by Eq. (73), while 79{.} projects orthogonally onto (3~ so that :P{0} E O ~ for all/~ E ~,,~+-B+I, and there exists a finite constant do so that [[ 0(t) - a0 [[2_< do < oo and ~0(t) > r0,min > 0 for all t > 0. The algorithm gain sequence n4,(t ) is given by
n,(,) = ,2o + where the vector r satisfying
1)+ II r is given by Eq.
2, 1
< no < o o
(80)
(76), and A is the fixed number
A1 + Ao < A < 1
(81)
for 0 < A1 < < 1 and A0 is defined by the assumption (S1). The role of the number no in Eq. (80) is to prevent division by zero in algorithm (79), and is chosen by the designer. Since 00 is unknown, as the adaptive control law, we will use the "certainty equivalence version" of (78), i.e.,
#(t)To(t) + Q(q-1)u(t ) + q[p(q-1) _ 1]yl(t) = p(q-1)y~( t + 1)
(82)
It is not difficult to see that the estimation scheme, actually represents the algorithm for the identification of the parameters 0o of the nominal system model. Specifically, from Eq. (73) and (82) we conclude that e(t) in the algorithm (79) is given by e(t + 1) = YY (t + 1) - 8(t) r r
(83)
and represents the prediction error, which is usually used in all parameter identification schemes. In the algorithm (79) we are using measurement vector r and e(t) with filtered input and output signals. The purpose of this filtering is to attenuate the influence of the large, high frequency unmodelled dynamics. Since at high frequencies f > f2, unmodelled dynamics are unstructured and complex, they can not be estimated. Our goal is only to estimate the low frequency (f < f2) system dynamics. The obtained
DISCRETE-TIME ROBUST ADAPTIVE CONTROL SYSTEMS
75
estimates for the low frequency system dynamics do not have to be significantly affected by the high frequency unmodelled dynamics. This implies that at high frequencies, updating of the parameter estimates should be suspended (in the ideal case) or significantly slowed down. In other words, we need a frequency selective adaptive controller. In the subsequent analysis it will be shown that the proposed adaptive algorithm provides such properties. Estimation scheme (79) actually represents the normalized gradient algorithm with projection, where we use nr instead of c+ II r c > 0, as is usual in similar situations. The motivation for selecting such nr is to make possible the usefulness of the Robust Ultimate Boundedness Theorem, formulated in the previous section. Specifically, for the construction of the corresponding Lyapunov functions in the proof of the Theorem 3.1, we need property (Aa)(i) introduced in Section 2, and n~(t) given by Eq. (80) satisfies assumption (A3)(i). Let us define the following constants:
7z -
l[ Se(z)Wz(z)I?d(z) Jilt.., 72 -11Ae(z)W2(z)I~(z)[[aH ~ Ae(z) Bo(z)W(z) II Z~(z) II~, C D B - I I D(z) [!~
CD~
--
CDp
-- II P(=) II~|
,
CD~-Ii
Q(=) II~--
c~ap
-
II Ae(z)P(z)~V(~) Be(~)~(~)2P(_-) D(z) [1~**, CDBp =[I D(z) [[~/**
Cw
-
II w.(~)~r
(84)
wh~r~ II" I[~o. is the "compressed" H ~ norm defined by Eq. (5), X is given by (81) and D(z) is defined by assumption (5'5). Since assumptions (Sz) and (5'5) hold, all the above defined norms exist. Regarding the intensity of unmodelled dynamics we introduce the following assumption: (S8)
("/1
"~" ~[2 ) C D p f2, performances of the designed low-pass filter W(z)should provide [Wi(A-Zl2eil)]~r(A-X/~eil)[ < 7, i = 1,2. The following lemma will be useful for future reference. Lemma_ 4.1" Let the assumptions (Sx) - (Sa), (Ss), ($6), (Ss) hold. Then 1) (BeWP + AeQ)u(t) - Aee(t + 1) + AePy~(t + 1 ) - PT(t) (85)
2) (Bor
+ AoQ)yj(t + Z) = Ber
+ Z) + Bor
+ 1)+ Q~(t) (S6)
76
MILOJE S. RADENKOVIC AND ANTHONY N. MICHEL
where the unit delay operator q-1 is omitted in all operator, while 7(t) and e(t) are defined by Eqs. (71) and (73) respectively.
3) II 7(t)I1~_< c~ II ~(t)IIx +k, + ~11($) (87') where II 7(t)Iix and II e(t)I1~ are defined by Eq. (1), when x(t) is equal to 7(t) and e(t + 1), respectively, while
(7, + 72)CDA
C~ = 1 -
(88)
(71 q" 72)CDP
and k~= (
1 [(7t -b 72)CDAPm1 -[" Cw 1 - (71 + 72)CDp
kw]
} ( 1 - 1A)t/2
4) II ~(z)I1~< c~ II ~(t)I1~ +k~ + e,2(t) where !! ~(t)!1~ i~ defined by Eq. (1), when z ( t ) - u(t), while C,, - CDA + CDpC.,,k,, = (1 - mlA)I/2 CDAP -[- ]r
(89) (90)
(91)
where C~ and k7 are defined by Eqs. (88) and (89), respectively. (92) 5) II yj(t)I1~< c~ II ~(t)Ii~ +~ + ~3(t) wh~,~ il y~(t)I1~ is given by Eq. (1), when ,(t) = y(t + 1), whil~, Cy = Cz, B + C~oC-,
(93)
and ml
CDBP q" kTCD Q.
(94)
In relations (87)- (92), constants 71,72, CDA, CDp, Cw, CDAP, CDB, CDQ and DDBp axe defined by Eq. (84) and the constants k~ and ml are defined by assumptions ($2) and ($6), respectively. Proof: The proof of the lemma is given in the Appendix. Important properties of the adaptive algorithm (79)-(82) will be formulated in the following lemma. L e m m a 4.2: Let the assumptions ( $ 1 ) - (Ss) hold. Then 1) ne(t) < C, n e ( t - 1)+ k, + ~14(t)
(95)
where n~(t)is defined by Eq. (2), when z(t) = e(t + 1) and e(~) is given by Eq. (73). The constants Ce and ke do not depend on the gain # of the algorithm (79), and will be specified in the proof of the lemma. At the same time, Co and ke depend on 7i, i - 1, 2, in such a way that their values decrease as 7i, i - 1, 2 decrease, where 7i i = 1, 2 are defined by Eq. (84).
DISCRETE-TIMEROBUSTADAPTIVECONTROLSYSTEMS 2) he(t) < Cr
77
(96)
C~2 or ~15($), 0 no, from (117), we obtain
Ee(t+l) 2 0 we have
84
MILOJE S. RADENKOVIC AND ANTHONY N. MICHEL
fast adaptation and the estimator generates correct parameter estimates. If the algorithms possesses fast adaptation over the interval [vk, ~rk), correct parameter estimates obtained in the interval [at-1, rk) can be significantly deteriorated. Specifically, over the time intervals [rk, crk), the bursting function S(t + 1) < 0 and from the analysis presented in Section III, it is obvious that then the Lyapunov function V(t) may diverge. We now propose two methods of enhancing the slow adaptation over the time intervals Irk, ak). The first method requires prior information related to the constants "11,72, Cw, CDA, COp and CDAp defined by Eq. (84), and the upper bound k~ of the noise w(t). In the next remark it is pointed out that based on the available prior information related to the physical system, the constants 7x, 72, CDA and Cop can be roughly estimated by the designer. The same holds for the constants C~, Cw and ChAp. Thus, if we know k~, we can estimate k~ defined by Eq. (89) and design the following estimation algorithm: over the intervals where ne(t) > 4 k ~ / ( 1 - ~ - Cv) 2 we are using small no in Eq. (80) and in the intervals where n,(t) < 4k.~/(1- a _ Cv)2 we can use estimation scheme (79) with no > > 4~u2k~/(1 - ~ - Cx)2, or completely stop adaptation. The second method for obtaining slow adaptation over the intervals [rk, cry) is to modify the sequence n#(t) in the algorithm (89). Specifically, instead of nr given by Eq. (80), we now define
he(t) = An~(t-- 1)+ II r
=
2
+
1-A
(133)
and we obviously require the upper bound kw of the noise w(t) to be known. Such choice of the sequence n,~(t) does not influence the results derived in Sections III and IV, except that constants Co, k0, C~2 from Lemma 4.2, are different. Since in the intervals [rk, ~rk), the bursting function S(t + 1) < 0, from Eq. (13), we obtain
e(t + 1) 2 _
1
~(t + 1) = ~(t) + m~(t) p.6(f) e(t + 1),
if So(t + 1 ) > 0
/~(t + 1) =/~(t),
if So(t + 1) _< 0
(147)
where 0(I) - 01, IIOx II_ 1 2) So(t + 1) changes its sign during the algorithm operation. In the first case the statements of the theorem follow trivially from the definition of So(t + 1) (Eq. (146)) and Eq. (147). This actually means that the initial value 0(1) is a good estimate of the real parameters. Before proceeding further with the analysis, let us show that the case S0(t + 1) > 0 for all t > 1 is impossible. From (145) we obtain ^
V(t -I- I)+ So(t -I- I) < V(t) + So(t) Ie(t )____+___~~ ~36(t) m#(~) m , ( t - I) -P~ m#(t) ~ m#(t) (151) wherefrom it follows that lim e(t + 1 ) 2 / m # ( ~ ) - O. Similarly, as in Lemma 4.2 from Eqs. assumption ($7)
(152)
(85) and (86), we can obtain by
,n,(t) _ T1, So(t + 1) < 0 by which it is proved that the case S0(t + 1) > 0 for all t _ 1 is impossible. Let us now consider the case when So(t + 1) changes its sign. Similarly, as in the proof of Theorem 3.1, we define the sequences rk and ak as follows -- T1 K0 t
Z
Atm-ie(J+ 1) 2 < k~.rl2
(169)
j--~O'k
for t E [uk, r~+l). Note that from Eqs. (143) and (146) we can obtain
So(, + 1)_< Z
j~ok
+
+
(170)
for t 6 [uk,rt+1). Using the fact that So(ok) K0. Therefore, So(t + 1) _< 0 for all t _> rKo- W e have thus proved that in the estimation algorithm (147), updating of the parameter estimates will be suspended after time t _> 7Ko. By this statement 2), the theorem is proved. Since for t >_ rKo, S0(t + 1) _< 0, statement 1) of the theorem follows simply
DISCRETE-TIME ROBUST ADAPTIVE CONTROL SYSTEMS
91
from Eq. (146). Statement 3) of the theorem is the trivial consequence of statement 1). Thus the theorem is proved. !:3 Theorem 5.1 establishes that the upper bound of the performance index is bounded and that the parameter estimates are constant after some finite time To. Thus for t > To drift of the controller parameters is eliminated, i.e., we have a fixed controller which implies that bursting can not occur for t > To. These performances are achieved without ~r-modification or projection in the estimation algorithm.
VI.
CONCLUSION
Using the concept of the bursting function, new insights into the robust adaptive control problem are obtained. For the estimation of the controller parameters, a normalized gradient algorithm with projection is proposed. Global stability of the adaptive system is established for a class of unmodelled dynamics and external disturbances which are unstructured, complex and large at high frequencies. It is shown that the small algorithm gains/~ may result in unacceptably large input and output signals. The presented results also demonstrate one possible way of incorporating a pr/or/system information into the robust adaptive control design. For the sake of simplicity, the system unit delay case is considered in this chapter. It is not difficult to see that using the proposed methodology, similar results can be established for the general delay" case.
VII.
APPENDIX
P r o o f of t h e L e m m a 4.1- The first two statements of the lemma follow by simple manipulations of Eqs. (70) and (73). Let us prove the rest of the lemma. From the definition of 7(t) (Eq. (71)), by assumption (S~), it is not difficult to see that
II 7(t) IIx< ('r, + 7.~) II u(t) II~,+ c ~ II ~(t) I1~,+69(t)
(A.1)
where [I z(t)I1~, z q. {7, u,w} is given by Eq. (1) in Section II. Constants 7i, i = 1, 2 and Cw are defined by Eq. (84). Similarly, from Eq. (86), we can obtain
II ~(t)i1~< eva [Ie(t)I1~ +CoAp [[u'(t)I[~ +cop II 7(0 I1~ +6o(0 (A.2) where II y'(t)IIx is defined by Eq. (1) when z(t) = y'(t + 1). The H r162 norms CDA, CDP and CDAP are defined by Eq. (84). Statement 3) of the
92
MILOJE S. RADENKOVIC AND ANTHONY N. MICHEL
lemma can easily be obtained by substituting (A.2) into (A.1), where we used the fact that assumption (5"8) holds and II ,,,(t)IIx_
0 .
(2.11) E!
The physical interpretation for the E O L problem is clear if one considers the inequalities in (2.8). Note that in this case
Er,
(2.12)
Ilu,
i=l
Hence, the cost function is an upper bound of the weighted summation of supremums of control signals over all L2 disturbances which belong to ~N" The EOL| problem is similar to the O L problem, but the input disturbances belong to different sets, f2 N and fir, respectively. The next section provides the solutions of the E O L problem in the continuous time case.
III. S O L U T I O N S
OF CONTINUOUS
TIME EOL
PROBLEM
In this section, we shall discuss the first order necessary conditions for static and dynamic controllers, and also present an iterative algorithm. A.
STATIC MEASUREMENT FEEDBACK CASE
For the static measurement feedback case, we seek a constant feedback gain matrix G which is the solution of the E O L problem such that
104
GUOMING G. ZHU AND ROBERTE. SKELTON (3.1)
u =Gz.
Therefore the EOL| problem becomes (3.2)
I min traceNW -~. trace R G M X M r G r P
G.W>O
P
subject to (2.1) and
[o[C,XC,r ].traceNW-I - -
0 such that i)
G = - ' f f - ' B r K X M r ( M , X M pr)-,.,
ii)
0 = KA + A r K + M r G r ' R - t G M p + CrQCp;
iii)
DrKD
iv)
Q{block diag[ C, x c r , C2 x c j ..... C x c r ]. t r a c e N W -' - F} = 0;
v)
A X + XA r + D pWD pr = 0 ,
P
P
= W - ' N W - ' ( t r a c e R G M X M rGr + traceQCpXC r)''2" P
P
(3.4a)
where "R = R. traceNW-' , Q = Q. traceNW-' , F = diag[ 2, I,,~.212 2 ..... 1~I 2 ], and l j is an identity matrix with dimension mj.
The proof of the above theorem is obvious. By using the following augmented cost function
H
trace NW -' { trace RGM pXM r G r } + +trace K[ X ( Ap
i=! + BpGMp ) r + ( Ap +
[ C x c ,rtrace NW -' - E I , (3.5)
BpGM, ) X
+ OpWD p r ],
GUARANTEEING OUTPUT L CONSTRAINTS
105
setting the partial derivatives with respect to X, G, K, W to zero, and adding one equation for inequality constraints in (3.2) (Kuhn-Tucker conditions), one can get the necessary conditions. By setting M = I , one obtains the results for state feedback case which is a special case of measurement feedback control.
Corollary 2.2 Suppose that G is an optimal solution of the EOL problem for the state feedback case, and that the EOL| problem is regular, that is, the EOL problem is not independent of the EOL cost function in (3.2). Then there exists a block diagonal positive semidefinite matrix Q such that i)
G = -'R-'BrKX;
ii)
0 = KA,, + A r K - KB,-R-'B rK + Cr'QCp ;
iii)
D ~, rKD
iv)
Q{block diag[ C, x c r , C~XC[ ..... C x c r ]. traceNW -' - F} = 0;
v)
(Ap + BpG) X + X (Ap + BpG) r + DpWDp r
P
m
p = W-I NW-I (traceRGXG r + traceQC X C ; ) |'~"
(3.6)
"-0,
g
where R , Q and F are the same as those in Theorem 3.1. The solution of the third equation in (3.3) and (3.6) is given in the following Lemma.
Lemma 3.3 Suppose that N and K are positive definite and D has full column rank. Then equation D r KD = W-INW-1 has one symmetric positive definite solution W~ = K~'[K~NK~] ~2K~-',
(3.7)
and one negative definite solution
w, where
r; ',
(3.8)
106
G U O M I N G G. Z H U A N D ROBERT E. S K E L T O N
(3.9)
K,~ = ( D r K D ) ''2 .
The proof was presented in [6]. An algorithm used to solve those necessary conditions is presented in Subsection C. B.
DYNAMIC FEEDBACK CASE Consider the following dynamic controller with order nc xc
=
AcXc + Bcz
u
=
C x +Hz.
(3.10)
It is well-known that by setting
=[APo ;x=
(3.1 la) =
;~=
;G= o
s
Ix:] [':] [5] ;y=
~
;D=
;D
=
x
"
;I)= B,,
; A
[01
(3.1 lb)
'
with compatible dimensions, the closed loop system can be written as (3.12) y
----
[d' , ,~'c" l" X-F H v
In order to obtain a bounded output L
9
norm for an L2 input, system
(3.12) should be strictly proper, i.e.,/4 = 0 or v - 0. Here we consider the case that v = 0. Then the closed loop system is in the form (1.2) with A=A+BGtr
C
[~r
21~/rGr]r
(3.13)
Let (3.14)
GUARANTEEING OUTPUT L CONSTRAINTS
107
where R = diag[r~,r2..... r ] > 0. By solving the following static measurement feedback EOL problem
trace NW -l . trace [CGl~I X~Ir G r
min
I
G.W>O
(3.15)
P
subject to (2.1) and
---
"
^T
[o[CXC ].traceNW
-1
~
2
_ e,, i = 1,2 ..... m.
one can obtain first order necessary conditions for the dynamic controller case. Hence, when v = 0 in (1.1), the EOL problem with a fixed order dynamic controller produces the static measurement feedback EOL controller as a special case. Special Case: Full Order Dynamic Controller For this special case, we consider the following strictly proper full order dynamic controller L
=
ax+
u
=
Cx.
z
(3.16)
Let
X =
IXPx-~Xc]; Y = [ ~ I ; w=IWvl'
(3.17)
then the closed loop system can be put into the form (1.2) with A -BM
I
BM c
D=
The EOL follows:
A +BC
A +BM
p
P 0
-BM
c
C;C= n
c
P 0
-A
(3.18a)
p
P.
(3.18b)
c
problem with full order dynamic controller can be stated as
108
GUOMING G. ZHU AND ROBERT E. SKELTON
min
traceNW-'.traceRC XC r, C = [ 0 , C ]
Ac ~c .Cr .W>O
(3.19)
subject to (2.1) and
"0[ C XC r ]. traceNW-1 0 such that Q{block diag[ C, x c r , C2 XC r ..... C XC r ]. traceNW-' - F} = 0,
(3.21)
and A =A,+B
C-B
M,;
(3.22)
B = ( X t , M pr + D W~2)W2-2" "~-i
C=-R
T
B K22,
where X , , and K22 satisfy X,, A ,r + A , X ,, _ ( X I M,
,r +D Wn)W~-l (X,,
M ,r
+D W~2)r +D W~,D ,r =~,
(3.23)
K=Ap + ArK= - K=B "R-'Br K= + Cr'QC = 0 , and "R = R . traceNW-' and Q = Q . traceNW-l . In addition DrKD = W-'NW-' (traceRC XC r + t r a c e Q C XCry )~2, where K is defined in (3.20) with K~2 = K22 and K~2 satisfying
(3.24)
GUARANTEEING OUTPUT L CONSTRAINTS
109
(3.25)
r ( K , , - K2:) + C~rR'C~ =0,
(K,,-K22)(A,,-BM)+(A,-BM)
and X is defined in (3.20) with X,2 = 0 and X22 satisfying
(3.26)
X22(A +B C ) r + ( A p + B p C ) X 2 2 +BW~2Br =0.
The proof is similar to the static measurement feedback case. Consider the augmented cost function m n
_~.
(3.27)
traceNW-' . traceRC XC r,, + E ' ~ [ C XC r 9traceNW-' - E,] I, : i=l
+traceK .[ AX + XA r + DWD r ].
Then (3.27) can be rewritten into H = traceXCrO.C + traceK .[ AX + XA rDWD r ] _ traceQ. F,
where 0 = block diag['Q,'ff ] and "R = R . traceNW-' , "Q = Q . traceNW-' . By setting the partial derivatives with respect to K, X, W to zero and adding the Kuhn-Tucker condition (3.21),one can complete the proof. _...
.__
Note that (3.22) and (3.23) describe an LQG controller if Q, R, and W are given. Hence, the full order E O L controller is an LQG controller with some special choice of the weighting matrices Q, R and input noise intensity matrix W. (The forthcoming algorithm determines Q, R and W). Remark 3.5 The fact that there exists a W > 0 satisfying the first order necessary conditions requires that D rKD > O. Remark 3.6 The first order necessary conditions for the OL problem only need to satisfy (3.22) and (3.23) for fixed W, which are included in the EOL| problem. Hence the OL| problem is a special case of the EOL problem.
C.
NUMERICAL ALC~RITHM
110
GUOMINGG. ZHU AND ROBERTE. SKELTON
Subsections A and B of this section provide all the necessary conditions for the design of static and dynamic controllers. The following algorithm provides a way to design the controller by iterating on these necessary conditions. This is an extension of the O L algorithm presented in [3].
THE EOL .o ALGORITHM Step I
Given system matrices Ap,Bp,Cp,Dp. Mp, initial weighting matrices Wo , Qo , R , outer product bound N, output L bounds e,, (i = 1,2 ..... m), error tolerance e , and free parameters 0 < [~< 1 and a >0. Define ~ = R. traceNW, -! ~ = Q. trace NW -~ Let i = 0
Step 2
Controller design step: Case 1- static state feedback Compute m-!
G, = - R
7"
(3.28a)
BpK,
by solving (3.28b)
K i A p + A ; K - K B p R---' i Bp"K -t- C;'QiC p = O, and then solve for X, by (Ap+BpG,)X, + X,(Ap+BpG,) r +DpWD r =0.
(3.28c)
Let D, = Op, q = Cp, and C~ = G,. Go to Step 3. Case 2: Static measurement feedback Iterate on the following three necessary conditions to obtain the optimal measurement feedback gain G i for given Qi and W
6,
- R -, ' B prK , Xi M pr ( M P X, M rP ) -''' O= K(Ap+BpG, Mp) r + K ( A p + B p G , Mp)
(3.29a)
=
+ M;
0.
(4.5)
Let w(.) in (4.2) be any e: disturbance which belongs to
a~ -
wO subject to (2.1) and o [ C XC, ]. traceNW -~ < _ E2, . . i. = . . 1. 2,
--
r
m.
Theorem 4.2 Suppose that G is an optimal solution o f the EOL problem defined in (4.17), and that the EOL problem is regular. Then, there exists a matrix Q = block diag[Q~, Q~ ..... Q= ] > 0 such that
116
GUOMING G. ZHU AND ROBERT E. SKELTON
i)
G - - ( R-- + BrpKBp) -' BrpKAFXMFr ( M pX M r ,) -l",
ii)
K = ArKA + MrG r ('R+ B;KBp)GMp +CrQCp;
iii)
D ; K D p = W - I N W -I ( t r a c e R G M p X M p r G r + trace __ QCp X C r )~,2.,
iv)
Q{ block diag[ C, X C r , C: X C r ..... C X C r ] . trace N W -' - F} : 0;
v)
A X A r + DpWD r = X ,
(4.18)
where - - = R . traceNW-', Q = Q. traceNW-', F = diag[e . 2I I ! ,e.212 2 ..... R
e21
],
(4.19)
a n d l j denotes an m j x mj identity matrix.
The proof of Theorem 4.2 is similar to that of Theorem 3.1. By setting M = I, one obtains the necessary conditions for the state feedback case, which is a special case of measurement feedback control. Corollary 4.3 Suppose that G is an optimal solution o f the E O L case, a n d that the E O L
f o r state f e e d b a c k
p r o b l e m is regular. Then there exists a matrix
Q = block diag[Q t , Q2 ..... Q. ] > 0 such that i)
G = -('R + BrKBF) -'BrKAp ;
ii)
0 = ArKAp - ArKBF ('ff+ BrKBp) -'BrKAF +Cr'QCp;
iii)
D pr K D p = W-' N W - ' ( t r a c e R G X G r + t r a c e Q C X C ; )
iv)
e { b l o c k d i a g [ C t X C r , C 2X C r ..... C X C r ]. t r a c e N W - ' - F} = 0;
v)
( Ap + B p G ) X ( A p + B pG ) r + D pW D rp= X ,
(4.20)
u2 .
where R , Q a n d I" are the s a m e as in (4.19).
The solution of the third equations in (4.18) and (4.20) can be solved by Lemma 3.3. The algorithm used to solve those necessary conditions will be given at the end of this section. B.
DYNAMIC FEEDBACK CASE We consider the fixed order dynamic controller
GUARANTEEING OUTPUT L CONSTRAINTS
x(k+l)
=
A x(k)+Bz(k)
u(k)
=
C x ( k ) + H z(k).
117
(4.21)
Let v = 0. Then the closed loop system has the following form x(k+l)
=
_ ( n - r) t h e s y s t e m w o u l d h a v e o n e zero at infinity. C o n s i d e r s u c h a M a n d a s s u m e t h a t t h e
158
SARIT K. DAS AND R K. RAJAGOPALAN
( n - - 1 ) re lo c at able zeros h a v e b e e n p l a c e d at origin, l e a d i n g to t h e e q u i v a l e n t plant configuration n
G1 = z 1n - - 1 / I ~ ( z l - - a ~ )
w h e r e zl "= z M
k-----1
F o l l o w i n g [9] (see A p p e n d i x C), it is n o t difficult to see t h a t t h e m a x i m u m n G M a t t a i n a b l e for t h i s p l a n t is (1 + r/)2/(1 - r/) 2 w h e r e r/ -- 1/1-Ikffil a ~ . Clearly, ~/ will be s m a l l e r for l a r g e r M , m a k i n g t h e G M lower. H e n c e M > ( n - r) n e e d n o t be c o n s i d e r e d . F i n a l l y , since t i m e i n v a r i a n t c o n t r o l l e r s do n o t h a v e t h e zero p l a c e m e n t c a p a b i l i t y , it is o b v i o u s t h a t for t h e case of n - r -- 1, M -- 2 is also to be considered. U] W e are n o w in a p o s i t i o n to o b t a i n t h e b e s t G M c o m p e n s a t i o n for a given p l a n t v i a t h e zero p l a c e m e n t a p p r o a c h . T h e s t e p s to be followed for this p u r p o s e are: 1. F o l l w i n g T h e o r e m 6 i d e n t i f y t h e values of M t h a t a r e to be c o n s i d e r e d . 2. F o r each of t h e s e v a l u e s of M o b t a i n t h e e q u i v a l e n t p l a n t [as given by eqn.(37)] a s s u m i n g t h a t all t h e r e l o c a t a b l e zeros h a v e b e e n p l a c e d at origin. 3. F o r e a c h of t h e s e e q u i v a l e n t p l a n t s o b t a i n G M M ( w h i c h s t a n d s for t h e b e s t G M o b t a i n a b l e u s i n g M - p e r i o d i c c o n t r o l l e r ) u s i n g t h e app r o a c h of [9] or [10]. T h e n o b t a i n t h e m a x i m u m G M as G M m a ~ -m a x [ G M M ] , M v a r y i n g over t h e r a n g e given by s t e p 1 a b o v e . M
4. F o r M = M m a x c o r r e s p o n d i n g to G M , n a x , o b t a i n t h e c o m p e n s a t o r a n d t h e c o m p l e t e C E u s i n g t h e a p p r o a c h of [10]. 5. F i n a l l y , to o b t a i n t h e p a r a m e t e r s of t h e p e r i o d i c g a i n c o n t r o l l e r t h a t a c h i e v e s t h e a b o v e C E , first check if t h e c o n t r o l l e r o r d e r rn as o b t a i n e d in s t e p 4 a b o v e satisfies t h e lower l i m i t c o n d i t i o n given by T h e o r e m 4. If it is satisfied t h e n this rn = rnmin, t h e m i n i m u m o r d e r of t h e c o n t r o l l e r r e q u i r e d . O t h e r w i s e , rnmir, will be t h e s m a l l e s t v a l u e of rn t h a t satisfies e q n . ( 3 1 ) . N e x t e q u a t e t h e coefficients of t h e like p o w e r s of t h e K - t e r m s of e q n . ( 2 8 ) to t h o s e of t h e d e s i r e d C E to g e t a set of n o n - l i n e a r e q u a t i o n s s o l v i n g which t h e c o n t r o l l e r p a r a m e t e r s c a n be obtained.
We n o w p r e s e n t a few e x a m p l e s of o p t i m u m G M c o m p e n s a t i o n of s t r i c t l y proper plants.
TECHNIQUES OF ANALYSIS AND ROBUST CONTROL
Example
4 C o n s i d e r t h e (2, 1 ) - p l a n t ( z -
159
1.5), Ibl > 1.
b)/z(z-
i) V a l u e s of M t o b e c o n s i d e r e d a r e 1 a n d 2. ii) F o r M = 2, t h e e q u i v a l e n t p l a n t w o u l d , o n z e r o r e l o c a t i o n , b e 1 / ( z t 2.25), for w h i c h G M M zt(zt
--- 6.76 a n d t h e c o r r e s p o n d i n g
C E is
-- 2 . 2 5 ) ( z t -- 1 / 2 . 2 5 ) + a K z 2 t = 0
C l e a r l y m = r n m i n = 1. N o w , o n e s e t o f t h e p a r a m e t e r
v a l u e s o f t h e first
o r d e r c o n t r o l l e r t h a t a c h i e v e s t h i s is o b t a i n e d t o b e qo(z)
-- z + 1.2b/(b
+
1.2)
qt ( z ) - - q o ( - - z ) p o ( z ) -- z + 1 / 1 . 5
pl(~) = o and the corresponding
c~ b e c o m e s
= 2(1.5 - b)(b + o . 6 6 7 ) / ( b + 1.2) iii) F o l l o w i n g [9] it is s e e n t h a t , for M
=
1, G M M
> 6.76 if a n d o n l y if
-r < b < 1.083. T h u s for b > 1.083 2 - p e r i o d i c c o n t r o l l e r s y i e l d b e t t e r results than time invariant ones. Example
5 C o n s i d e r t h e (3, 1 ) - p l a n t ( z -
b)/z(z
- 1.5)(z-
2).
i) T h e v a l u e s o f M t o b e c o n s i d e r e d a r e 1 a n d 2. ii) F o r M "- 2, t h e e q u i v a l e n t p l a n t a f t e r t w o z e r o s a r e r e l o c a t e d t o o r i g i n would be zt/(zt - 2.25)(zt -4), for w h i c h G M M -- 1.56 a n d t h e c o r r e s p o n d i n g C E is z t ( z t -- 2 . 2 5 ) ( Z l -- 4 ) ( z t -- 1 / 2 . 2 5 ) ( Z l -- 1 / 4 ) + o ~ K z ~ ( z t H e r e rn -- rn,,+in = 2. C E for b - 1.5 is
- 1 . 6 ) ( Z l - 1 / 1 . 6 ) -- 0
Now a second order controller which achieves this
q o ( z ) = 0 . 3 3 6 z 2 + 0 . 1 6 9 z + 0.946 qt ( z ) -- q o ( z ) p o ( z ) -- z 2 + 3 . 6 8 2 z + 1.109 P l ( z ) -- - - 3 . 2 6 8 z + 1.05 and the corresponding
c~ is a = 0.799.
iii) F o r M = 1, G M M
> 1.56 if a n d o n l y i f - 1 . 2 6 5
< b < 1.25.
T h u s for
values of b outside this range the 2-periodic controller yields better results.
160
SARIT K. DAS AND E K. RAJAGOPALAN
Example
6
C o n s i d e r t h e (4, 1 ) - p l a n t ( z -
b)/z2(z-
1.5)(z-
2).
i) M = 1, 2 a n d 3 n e e d o n l y be c o n s i d e r e d . ii) F o r M ---- 2, o n l y two z e r o s c a n be r e l o c a t e d to origin. T h e n t h e e q u i v a lent p l a n t w o u l d be 1 / ( Z l - 2 . 2 5 ) ( z 1 - 4 ) , for w h i c h G M M -~ 1.08, a n d t h e c o r r e s p o n d i n g C E is z2(zl
2.25)(Zl - 4 ) ( z l - 1 / 2 . 2 5 ) ( z l - 1 / 4 ) ( z l
-
+ o, K z ~ ( z l
-
+ 0 . 1 7 4 ) ( z l + 5.734) 1 . 9 2 ) ( z l - 1 / 1 . 9 2 ) -- 0
O b v i o u s l y a f o u r t h o r d e r c o n t r o l l e r will be r e q u i r e d t o a c h i e v e t h i s c o m pensation. iii) F o r M --- 3, t h e e q u i v a l e n t p l a n t a f t e r z e r o r e l o c a t i o n w o u l d b e z l / ( z , 3 . 3 7 5 ) ( z l - 8 ) , for w h i c h G M M "- 1.16 a n d t h e c o r r e s p o n d i n g C E is
-
Z2(Zl -- 3 . 3 7 5 ) ( Z l -- 8)(z I 1 / 3 . 3 7 5 ) ( Z l -- 1 / 8 ) + o l K z ~ ( z l -- 2.46)(Zl -- 1 / 2 . 4 6 ) = 0 -
-
H e r e t h e c o n t r o l l e r o r d e r n e c e s s a r y is a p p a r e n t l y 2. T h i s value, h o w e v e r , d o e s n o t s a t i s f y e q n . ( 3 1 ) w h i c h r e q u i r e s t h e m i n i m u m c o n t r o l l e r o r d e r to be 3. So, a t h i r d o r d e r c o n t r o l l e r is to b e e m p l o y e d . B u t t h e t h i r d c o n t r o l l e r zero a n d p o l e m a y b o t h b e p l a c e d at o r i g i n so t h a t t h e a b o v e C E r e m a i n s essentially the same. iv) For M = 1, t h e b e s t G M of 1.22 is o b t a i n e d w h e n Ib[---. 1. F o r b ~ 1.5 or 2, h o w e v e r , t h e a t t a i n a b l e G M is o n l y 1. T h e n 3 - p e r i o d i c c o n t r o l l e r s will be t h e b e s t . Example
7 C o n s i d e r a g a i n a (4, 1 ) - p l a n t ( z - b ) / z ( z
+ x/-~)(z
2 - 2 z + 2).
i) M -- 1, 2 a n d 3 n e e d o n l y be c o n s i d e r e d . ii) F o r M -- 2, t h e p l a n t a f t e r zero r e l o c a t i o n w o u l d b e Z l / ( Z l for w h i c h G M M -- 1.56, t h e c o r r e s p o n d i n g C E b e i n g zl(zl
-
+ r
1.5)(Zl2 4 4 ) ,
1.5)(z~ + 4)(Zl - 1 / 1 . 5 ) ( z ~ + 1 / 4 ) ( z l + 1 . 5 ) ( z l + 1 / 1 . 5 ) - 2 . 1 4 ) ( z 1 - 1 / 2 . 1 4 ) ( z 1 + 2 . 1 3 ) ( z l + 1 / 2 . 1 3 ) -- 0
C l e a r l y a 5 t h o r d e r c o n t r o l l e r is r e q u i r e d to a c h i e v e t h i s c o m p e n s a t i o n . s u c h e v e n - i n p u t c o n t r o l l e r is
One
q o ( z ) = z ( 0 . 0 0 3 4 z 4 - 4 . 4 2 3 2 z 3 - 8 . 3 3 9 5 z 2 - 6 . 0 1 2 4 z - 1.3049) ql ( z ) "- q o ( z ) po(z)
-- z s + 1.8851z 4 + 2.4966z 3 + 2.506z 2 + 1.5455z + 0.4083
pl(~)
= 0
iii) F o r M - 3, t h e z e r o r e l o c a t e d p l a n t w o u l d be z ~ / ( z l + 1.837)(z~ + 4 z l + 8), for w h i c h G M M -- 1.313, a n d t h e c o r r e s p o n d i n g C E is
Z2(Zl + 1.837)(Zl2 + 4zl + 8 ) ( z I + 1 / 1 . 8 3 7 ) ( z ~ + 0 . 5 z l + 1 / 8 ) + c ~ K z ~ ( z 2 + 3 . 1 1 8 z l + 3 . 0 5 3 ) ( z ~ + 1.02zl + 0 . 3 2 7 ) - - 0
TECHNIQUES OF ANALYSIS AND ROBUST CONTROL
161
A f o u r t h o r d e r controller would be required to achieve this. iv) For M = 1, t h e best G M of 2.53 is o b t a i n e d when Ibl ---* 1. For b ---. - x / 1 . 5 , however, the GM will be 1, a n d t h e n t h e 2-periodic controller will be t h e best.
VII
Conclusions
and Discussions
T h e c o n t e n t s of this c h a p t e r can be s u m m a r i s e d as follows: 1. A linear, discrete, M - p e r i o d i c controller s t r u c t u r e t h a t has t h e m a x -
i m u m possible degrees of freedom for its o r d e r has been presented. 2. T w o a n a l y t i c a l m e t h o d s - the lifting technique and t h e Floquet theory approach ~ have b e e n p r e s e n t e d for dealing with periodic systems. It is seen t h a t while the f o r m e r is m o r e general in t h a t it can, besides analysing stability, yield t h e i n p u t - o u t p u t relation as well, the l a t t e r yields t h e C E in a m o r e s t r a i g h t f o r w a r d fashion. 3. Based on a s t u d y of t h e features of the closed loop C E of a L D T I p l a n t c o m p e n s a t e d by a M - p e r i o d i c controller, t h e zero p l a c e m e n t c a p a b i l i t y of periodic controllers has been investigated. It is seen t h a t while periodic controllers can relocate all the zeros of a n t h o r d e r bicausal plant, they can relocate only u p t o ( n - 1) zeros ( s o m e of which might originally have been at infinity) for n t h o r d e r strictly p r o p e r plants. 4. Finally the aspect of GM i m p r o v e m e n t of u n s t a b l e p l a n t s with N M P zeros via the zero p l a c e m e n t a p p r o a c h has been i n v e s t i g a t e d leading to t h e conclusions t h a t (a) 2-periodic controllers can provide infinite G M c o m p e n s a t i o n to bicausal plants, a n d (b) for o p t i m u m G M comp e n s a t i o n of a n-pole, r-zero strictly p r o p e r plant the p e r i o d i c i t y M of t h e controller should, in general, be a p r i m e n u m b e r not g r e a t e r t h a n m a x { 2 , (n - r)}. It m a y be n o t e d t h a t zero p l a c e m e n t is not the only way t h r o u g h which G M c o m p e n s a t i o n can be provided to a plant, in fact, in [2] p e r i o d i c controllers have been used to do t h e s a m e job via t h e factorization a p p r o a c h which, as such, is m o r e general t h a n t h e zero p l a c e m e n t a p p r o a c h . T h e p r o b l e m of o p t i m u m G M c o m p e n s a t i o n using the f a c t o r i z a t i o n a p p r o a c h , however, has not so far been solved in literature. (Some results r e g a r d i n g this are p r e s e n t e d in [11].) Besides GM i m p r o v e m e n t , t h e possibilities of using periodic controllers for d i s t u r b a n c e rejection and for s i m u l t a n e o u s stabilization have also been
162
SARIT K. DAS AND P. K. RAJAGOPALAN
investigated in literature. Regarding the former, a l t h o u g h it has been shown [2][3] t h a t periodic controllers offer no a d v a n t a g e over L D T I ones so far as u n i f o r m d i s t u r b a n c e rejection is concerned, the possibility t h a t such controllers may provide superior d i s t u r b a n c e rejection at certain r e g u l a r intervals of t i m e (say, the even instants) can not be ruled out. Further, regarding the latter, although it has been claimed in [2] t h a t generically a M - p e r i o d i c controller can not stabilize a set of M plants s i m u l t a n e o u s l y for M > 2, it has been shown in [12] t h a t M - p e r i o d i c controllers can not only stabilize but can as well place the poles of M plants s i m u l t a n e o u s l y for any M.
VIII
References
1. T. Kailath, L i n e a r S y s t e m s , Prentice Hall Inc., Englewood Cliffs, N.J. (1980). 2. P. P. K h a r g o n e k a r , K. Poolla, and A. T a n n e n b a u m , " R o b u s t control of linear time invariant plants using periodic c o m p e n s a t i o n " , I E E E T r a n s a c t i o n s on A u t o m a t i c C o n t r o l A C - 3 0 , pp.1088-1096 (1985). 3. J. S. S h a m m a , and M. A. Dahleh, " T i m e varying versus t i m e invariant c o m p e n s a t i o n for rejection of persistent b o u n d e d d i s t u r b a n c e s and robust stabilization", I E E E T r a n s a c t i o n s on A u t o m a t i c C o n t r o l A C 36, pp.838-847 (1991). 4. M. A. Dahleh, P. G. Voulgaris, and L. S. Valvani, " O p t i m a l and robust controllers for periodic and m u l t i r a t e systems", I E E E T r a n s a c t i o n s on A u t o m a t i c C o n t r o l AC-37', pp.1734-1745 (1992). 5. S. K. Das, and P. K. R a j a g o p a l a n , "Periodic d i s c r e t e - t i m e systems: stability analysis and robust control using zero p l a c e m e n t " , I E E E T r a n s a c t i o n s on A u t o m a t i c C o n t r o l A C - 3 7 , pp.374-378 (1992). 6. E. T. W h i t t a k e r , A c o u r s e in m o d e r n a n a l y s i s , C a m b r i d g e University Press, N.Y. (1962). 7. B. A. Francis, and T. T. Georgieu, "Stability theory for linear timeinvariant plants with periodic digital controllers", I E E E T r a n s a c t i o n s on A u t o m a t i c C o n t r o l A C - 3 3 , pp.820-832 (1988). 8. S. K. Das, and P. K. R a j a g o p a l a n , "Infinite d e t e r m i n a n t m e t h o d s for stability analysis of periodic s y s t e m s " , P r o c e e d i n g s I E E P a r t - D 131, pp.189-201 (1984).
TECHNIQUES OF ANALYSIS AND ROBUST CONTROL
163
9. A. T a n n e n b a u m , "Feedback s t a b i h z a t i o n of linear d y n a m i c a l plants with u n c e r t a i n t y in the gain factor", International Journal of Control 32, pp.1-16 (1980). 10. I. Horowitz, "Design of feedback s y s t e m s with n o n - m i n i m u m - p h a s e u n s t a b l e plants", International Journal of System Science 10, pp. 1025-1040 (1979). 11. S. K. Das, and P. K. R a j a g o p a l a n , " M - p e r i o d i c controller for maxim u m gain m a r g i n for N M P s y s t e m s " , P r e s e n t e d at S I A M Conference on Linear Algebra, San Francisco, Nov. 5-8, (1990). 12. A. K. Singh, "Periodic c o m p e n s a t i o n of discrete linear time invariant plants", Ph.D. Thesis, Indian I n s t i t u t e of Technology, K h a r a g p u r , India (1994). 13. C. A. Lin, and C. W. King, "Minimal periodic realizations of transfer m a t r i c e s " , I E E E Transactions on Automatic Control A C - 3 8 , pp.462466 (1993).
IX A
Appendices SISO
Periodic
Realization
of Transfer
Matrices
In Section III-A we have seen how the equivalent M x M transfer m a t r i x ,
G(z M), c o r r e s p o n d i n g to a M - p e r i o d i c . s y s t e m can be o b t a i n e d . Here we a t t e m p t the reverse p r o b l e m of obtaining the minimal, SISO, M - p e r i o d i c realization of a given M x M C~ satisfying the causality condition (~(oo) lower triangular. It will, however, be seen t h a t such a m i n i m a l realization does not utilise all its possible degrees of freedom. Noting t h a t for a 2 x 2 transfer m a t r i x C~(z2) the (1, 1)-element, G l l , s t a n d s for the even instant input to the even instant o u t p u t t r a n s f o r m a t i o n , and, similarly, G12, G21 and G22 stand, respectively, for the odd to even, even to odd and odd to odd t r a n s f o r m a t i o n s , G may as such be realized by realizing each GU, i , j = 1, 2, individually with the a p p r o p r i a t e input signal applied to it and its o u t p u t taken at the correct instants. Fig.4 shows this realization. (Note t h a t for causality the o u t p u t s of the blocks G12 and G21 must be delayed by one sampling period.) It is clear t h a t for a M x M G(z M) to be realized in this fashion one would require (in the worst case) M2rn M - d e l a y elements (i.e., 1/z M blocks) where rn is the order of t h e c o m m o n d e n o m i n a t o r d(z M) of the elements of (~. However, it a p p e a r s logical t h a t the s a m e should be realizable using only as m a n y delay e l e m e n t s as is the highest degree of z occurring in d (i.e., M r n ) provided, of
164
SARIT K. DAS AND R K. RAJAGOPALAN
y,(.,~)
U~(~ 2) "-~ (z~)
,
!
U
J_l] 7 z
J .
I
--1
7 ~
z - 1 r.,ro(z 2 )
7 ,G"(z
)
_/
"x.
z - ~ Y o ( z 2)
F i g u r e 4: D i r e c t r e a l i z a t i o n of a 2 • 2 t r a n s f e r m a t r i x
course, t h e g a i n s are now allowed to b e c o m e p e r i o d i c . W e s h o w n e x t t h a t such i n d e e d is t h e case. F i r s t we b r i n g o u t an i m p o r t a n t p r o p e r t y of t h e G c o r r e s p o n d i n g to a c a u s a l p e r i o d i c m a p ( b e s i d e s t h e one t h a t G ( o o ) is lower t r i a n g u l a r ) [12]. 3 Given a causal, SISO, M-periodic system, if one constructs the corresponding M x M L D T I transfer matriz G ( z M) without performing any (possible) pole-zero cancellation at origin in its elements and ezpresses the 9~m~ ~ ~ ( ~ ) = ~(zU)/d(~), ~ h ~ d i~ th~ t.~.m, olth~ a r of G and 1V is a polynomial matriz, then, apart from G(oo) bein 9 lower triangular, fit(O) would be upper triangular. Conversely, if.for a given G with
Lemma
G(o~) tow~,, t,.ia,~g,,u,,, th~ .,~(o) i, . o f ,,pp~,. t,.i,,ng,,h,,., th~n it sig,,,i/ie~ that the system has a hidden mode at origin not reflected in d. Proof: For c o n v e n i e n c e we p r o v e t h e r e s u l t for M = 2. T h e p r o o f can easily be e x t e n d e d for M > 2. It was s h o w n in S e c t i o n I I I - A t h a t t h e even a n d o d d o u t p u t s of a c a u s a l 2 - p e r i o d i c m a p are r e l a t e d to t h e even a n d o d d i n p u t s by t h e t r a n s f e r m a t r i x G as given by e q n . ( 7 ) or, a l t e r n a t i v e l y , by t h e m a t r i x r as given by e q n . ( 8 ) . N o t e , however, t h a t of t h e two m a t r i c e s r a n d r G is m o r e f u n d a m e n t a l b e c a u s e it r e l a t e s t h e even a n d o d d i n p u t a n d o u t p u t s i g n a l s in t h e c o r r e c t fashion. In o t h e r words, given a 2 - p e r i o d i c m a p , its 2 • 2 L D T I r e p r e s e n t a t i o n can p h y s i c a l l y be o b t a i n e d o n l y in t h e f o r m G, a n d , conversely, a given 2 • 2 m a t r i x of t h e form of r m u s t first be t r a n s f o r m e d to t h e form r if it is to b e p h y s i c a l l y realized as a 2 - p e r i o d i c s y s t e m . Let d'(z 2) be t h e l.c.m, of t h e d e n o m i n a t o r s of Gij(z2), i , j = 1,2. Now, d e p e n d i n g on w h e t h e r G i j , i ~: j , h a v e n u m e r a t o r f a c t o r s of z 2 or n o t , four cases m a y arise:
TECHNIQUES OF ANALYSIS AND ROBUST CONTROL
165
1. Neither G12 nor G21 has a n u m e r a t o r .factor of z 2" T h e n f r o m eqns. ~,
(7) a n d (8) t h e c o m m o n d e n o m i n a t o r forms for G a n d G b e c o m e
[ z2Nll G -z N21
(~ --
z Nil z 2 N2i
zN12 ] z 2 N~2
1
N12 ] 1 z 2 N22 Z 2d'
1V d
w h e r e Nij --- G q d ' a n d d e g ( N i j ) o
Vk ~ [kL, ku]
(33)
STABILITY OF DISCRETE NONLINEAR FEEDBACK SYSTEMS
239
where
d2
-
- b T(P T PT PdlP2)b
dl
-
2{aT(p + P f P d l P 2 ) b - bTpTPdlpn-1} Pn,n - aT (p + pT Pdl P2)a - p nT_ l P d l p n _ l
do -
-t-2aT pT Pdlpn_ 1.
(34)
Since detPd > 0, kL and ku can be obtained as two real roots of the quadratic equation q(k) =
+
+ do = 0.
(35)
Since A is assumed to be stable, q(0) > 0, which implies that do > 0. From this and the fact that d2 < 0, eq. (35) always has two real roots. This means that there always exists a set of simultaneous Lyapunov feedback gains and it is a connected closed interval [kn, ku] determined by the two real roots of q(k), kn and ku. The simultaneous Lyapunov sector obtained here is the maximum sector for a certain choice of the Lyapunov matrix equation, or more specifically for an arbitrary positive definite matrix Q in eq. (27), which means that the thus obtained sector is not necessarily the maximum simultaneous Lyapunov sector for the given nonlinear feedback system. However, it should be emphasized that the simultaneous Lyapunov sector [kL, ku] and corresponding positive definite matrix P for the Lyapunov function xT(t)Px(t) for a given nonlinear feedback system can be obtained at the same time. The following example illustrates this. Example 2 Consider the same plant of Example 1 described by
G(z)-
1
2
9
z + 0.3z - 0.4
(36)
A corresponding realization in the observability canonical form is
A_[0 04] 1
-0.3
1
cT
=
[0,1].
(37)
Note that the Hurwitz sector for this case is [-0.3, 1.4]. In the following, three sets of the simultaneously stabilizing feedback gains are evaluated according to the choice of a symmetric positive definite matrix Q.
240
YASUHIKO MUTOH ET AL.
1) Q-
[10 o] ~_[ -0.921 284, 1
-0.921 ] 1.841
(38)
a simultaneously stabilizing gain sector :
2)
[kL, kv] = [-0.263, 1.029]
(39)
] Q_[21 31] ' ~_[ -0.651 6.635 -0.651 4.635
(40)
a simultaneously stabilizing gain sector : (41)
[kL, kU] = [-0.297, 1.225]
l tp
qg(y)= 1.4y Hurwitz Sector Simultaneous Lyapunov Gain Sector qgfy)=1.225y ~y) = -0.297y =
Fig.8 Hurwitz Sector and Simultaneous Lyapunov Sector for Example 1
-0.3y
STABILITY OF D I S C R E T E N O N L I N E A R F E E D B A C K S Y S T E M S
241
3) Q-
[5 2] 2 1 '
[6.587 2.540
P-
2.540] 1.587
(42)
a simultaneously stabilizing gain sector :
[kL, ku] = [-0.084, 0.301]
(43)
Among the above cases, since the second sector is the largest, it should be taken as a set of the simultaneously stabilizing feedback gains. Fig. 8 shows the Hurwitz sector and the simultaneously stabilizing gain sector for this plant. The corresponding positive definite matrix P is given by eq. (40) for the Lyapunov function of this nonlinear feedback system. Fig. 9 shows the contour of this Lyapunov function in the state space. It must be noted that another simultaneous Lyapunov sector may exist which contains those obtained here.
/
:
\i
eq
i i
!
O
!
//, X1
Fig.9 Contour of xT Px (Case 2)
242
YASUHIKO MUTOH ET AL.
V. ALGEBRAIC RICCATI INEQUALITY APPROACH
In this section, the absolute stability of multivariable systenls is discussed based on algebraic Riccati inequality approach. It will be shown that for the absolute stability of multivariable systems, a certain transfer matrix must satisfy some H ~ -norm condition, which can be regarded as the general circle criterion for the multivariable case. The relation between the absolute stability and quadratic stability will be shown also as will the fact that if the absolute stability is guaranteed by the Hor norm condition, the stability implies the exponential stability.
A. ABSOLUTE STABILITY OF MULTIVARIABLE SYSTEM Consider a multi-input and multi-output discrete time nonlinear feedback system ~ M described by the following equations. Linear Part: z(t+l) y(t)
= =
Ax(t)+Bu(t) Cx(t) + Du(t)
(44)
Nonlinear Part: u(t) = - ~ ( y ( t ) , t )
(45)
where u(t), y(t) E R m, A E R nxn, B E R '~xm,C E R mxn and D E R mxm. It is assumed that A is stable and the linear part is controllable and observable. The nonlinear function ~ : R m x R --~ R m is assumed to satisfy p(O, t) = O, and p(g(t), t) is said to lay in sector [kL, ku], if p(y(t), t) satisfies the following quadratic constraint {~(y(t), t) - kLg(t)} T {p(y(t), t) - f u y ( t ) } 0 and the Riccati inequality - p+ATpA+CTC+(ATpB+CTD)N-I(BTpA+DTC)
< O
(47)
holds.
!
The following result gives a sufficient condition for the absolute stability of the system NM with respect to a particular sector I-k, k](k > 0). T h e o r e m 4 System NM is absolutely stable with respect to the sector 1
[-k, k] (k > O)if [[a(z)llo, < ~.
!
(Proof) Note that since kv - --kL -- k the quadratic constraint eq. (46) becomes ~T(y(t), t)~(y(t), t) 0 satisfying the Riccati inequality eq. (47) with 7 - ~. Consider the quadratic function V ( x ( t ) ) " - xT(t)Px(t) as a candidate of Lyapunov function for system EM. Then,
AV(x(t))
= :
V(x(t + 1 ) ) - V(x(t)) {Ax(t) - Bp(y(t), t)} TP{Ax(t) - BT(y(t), t)} --xT(x)Px(t) X T (t)(A T PA - P)x(t) - 2~ T (y(t), t)B T PAx(t) +pT (y(t), t)B T PB~(y(t), t). (49)
By substituting eq. (47) into eq. (49) and completing the squares gives
AV(x(t))
0, system eq. (57) with the perturbation satisfying eq. (64) is quadratically stable if there exist a scalar ~ > 0 and a positive definite matrix P > 0 such that k 2 I - A2ETpE > 0 and the Riccati inequality 1 T G + A2AT P E { k 2 I - ~2ET RE} -1 E TPA < 0 (66) - P + A TPA + ~-~G
holds.
!
From Lemma 3, it is easy to show that the condition in Corollary 2 is equivalent to the H ~ norm constraint IIC(zI - A) - Etl
_ 1) function in z with f(t, 0) - 0, vt ~ R. D e f i n i t i o n 9 System eq. (70) is said to be exponentially stable if and only if for any to and xo - x(to), there exist constants 0 < fl < 1 and p(.~0) > 0 such that the solution of eq. (70) with z(to) - xo satisfies
IIx(t)l[ < p(~0)~ (~-'~
(71)
for all t >__to.
l
D e f i n i t i o n 10 A continuous function ff 9R + ~ function, if it is strictly increasing with ~(0) - 0.
R + is called a class-K; l
The following Lemma is a slight modification of Lemma 1 in [13]. L e m m a 4 For system eq. (70), if there exists a function Y ( t , x ( t ) ) with V(t, 0) - 0 such that
1) y(t,~(t))>_ r
vt _ to.
2) A V ( t , x ( t ) ) - V(t + 1,x(t + 1 ) ) - V ( t , x ( t ) ) < M - c r V ( t , x ( t ) ) ,
vt >_ to.
for some ff E/C, and constants M > 0 and 0 < a < 1, then t-to-1
r
< w(t, ~(t)) < (1 - ~)('-to)W(to, ~(t0)) + M Z i=0
(1 - a) i. (72)
STABILITY OF DISCRETE NONLINEAR FEEDBACK SYSTEMS
249
(Proof) From condition 2 ,
V(t,x(t)) 0
Q
:~
-{-P
(75)
1 GT G + A2AT P E N - 1E T PA} > 0 (76) + A T P A + --~
Then, for any admissible AA(t) system eq. (57) is exponentially stable with
C(xo)
"-
/ d
Xl
Pxo
(77)
v "-
x/'l - a
(78)
where a is any constant scalar satisfying (79)
(Proof) Suppose that there exist a scalar ,k > 0 and a positive definite matrix P > 0 satisfying eqs. (75) and (76). Let
r
m;
(P)llx(t)lt 2
(80)
where I1" II denotes the Euclidean norm. It is clear that ~ E/C. Define a function Y(x(t)) xT(t)Px(t). (81) -
Then, V(x) >_ ((]Ix[]) and from the proof of Theorem 6, AV(z(t)) __ -~mi~ (Q)x T (t)x(t)
(82)
250
YASUHIKOMUTOHET AL.
From eq. (82)
AV(x(t)) It has b e e n and o n l y (8)
=
if the
the
(~8)
x0,
a control states
input u with m i n i m u m
to the o r i g i n
at a g i v e n
to. shown
that
the
solution
controllability
is n o n - s i n g u l a r
[26] .
The
is p o s s i b l e
Grammian optimal
given
control
if
by Eq. law
is
given by U. where
-- B T e A T (t-t~
=
star d e n o t e s
demonstrates Grammian The
the optimal
the n e c e s s i t y
to be invertible,
condition
(19)
(to, tf) x0, control.
(19)
for the c o n t r o l l a b i l i t y i.e of full rank as a
for c o n t r o l l a b i l i t y .
observability
considering
the
Grammian
same a u t o n o m o u s
can
be
= Cx(t).
interpreted
s y s t e m with
taken as y(t)
Equation
(20)
by
the output
262
A H M A D A. M O H A M M A D AND J. A. DE A B R E U - G A R C I A
Now,
with
knowledge
interval state input Eq.
[t0, tf],
x 0.
of it
Without
is loss
of
both
[to,tf],
into Eq.
the
initial
the
control
for x(t)
from
(20) gives
of
Eq.
(21)
by
eAT(t-t~
T
__ e A T (t-to)c T C e A (t-to) X0,
Eq.
both
sides
re-
eAT(z-t~ CTy (~) dZ
x0 =
solution
to
can
given by
be
the
(22)
over
the
itf 0
interval
eAT(z-to)cTce A(~-t~ d~xo,
(~) d~,
necessity
for
the
order
to
invertible also
be
represents be
Eq.
eAT(~-t~
to the p r o b l e m
should
Grammian
=
Wo I (to, tf)
indicates
Grammian
of
(22)
(22) becomes
itf 0
This
Solving
the
in
integrating
It
find
over
(21)
sides
e AT(t-to) CTy (t)
which
to
output
C e A(t-t0) x0,
multiplying sults
and
generality,
zero.
(18) and s u b s t i t u t i n g =
input
desired
u can be c o n s i d e r e d
y(t)
or
the
seen
in
being noted
a measure by
(23)
(24) observability have
a unique
considered. that of
considering
the the the
observability output output
energy. energy
CONTINUOUS AND DISCRETE TIME LYAPUNOV EQUATIONS
i
h
263
(25)
tf yT (~) y (I;) d~, 0
e A T (~-t0)c T c e A (~-t0) dZx0,
=
(26)
d to
=
x~W0 (to, tf) x0.
E.
THE
DISCRETE
Consider x(k+l)
The
=
x(0)
Lyapunov
time
EQUATIONS system
[15] (27)
= x 0, function
x (k) T p x (k) .
change =
V(k+l)
AV
=
x(k)T(ATpA
means
stable,
ATpA can
as (29)
-V(k),
that
the
or if one
(28)
in V is d e f i n e d
AV
This
one
the
LYAPUNOV
discrete
= Ax(k),
and define V
the
(27)
_p)x(k)
for
change
.
system
(27)
in V m u s t
be
(30) to
be
asymptotically
non-positive
definite,
writes -
p
=
derive
tinuous
time
meaning
of
-
(31)
Q,
similar
case.
stability
The
arguments only
in b o t h
to
those
difference cases
[24].
of
the
is t h a t
con-
of the
264
A H M A D A. M O H A M M A D AND J. A. DE A B R E U - G A R C I A
Except and
for
theorems
equations case.
are
similar
_
Wc
ATWo A
-
W o =_
time
-- -
of the
to
equations
AWcAT
the
meaning
concerning
These
where
stability, discrete
those
of
definitions
time
the
Lyapunov
continuous
time
are
BBT,
(32)
cTc,
(33)
matrices
[A,B,C]
are
those
of
the
discrete
system.
This
concludes
retical III.
aspects
the
Lyapunov
the
OF
THE
early
equation
LYAPUNOV
1960 's,
This
importance
and
review
of
the
theo-
functions/equations.
has b e e n
investigation.
extreme
historical
of L y a p u n o v
SOLUTIONS
Since of
the
the
under
is
EQUATIONS
solution
a considerable
particularly
vast
to
due
the
amount
to
their
number
of
applications
took
on
three
in
the area of controls. The
research
in
this
area
directions.
Namely,
tion
implications,
and
tions,
its
and n u m e r i c a l
In the solution, [29],
one
finds
Barnett
Barnett [16].
in this
Section
II.
et
aspects
closed
the
solu-
form e x p l i c i t
solu-
the et
key
al
al
theoretical papers
[30] . [15],
direction the
have
by
aspects
Gantmacker that most
already
interested
of
Taussky
Other m a i n
It s h o u l d be n o t e d However
of
solutions.
direction,
and O s t r o w s k i
include sults
first
theoretical
different
been reader
et
the al
references [31],
and
of the
re-
addressed
in
is r e f e r r e d
CONTINUOUS AND DISCRETE TIME LYAPUNOV EQUATIONS
to
references
above
and
references
therein
265
for
more
details. In the the
second
research
when
the
Jordan gives
is
focused
system
form
in this
direction, is
or any
direction
a summary
continuous Other [34] .
It
require
in
other
be
special
form
Power
time
are
that
transformation
of
This
is
a
solution such
[32,33].
to
as
A key p a p e r Power
for both
Lyapunov
due
noted
form.
form.
form s o l u t i o n s
discrete
solutions,
the
to H.M.
contributions
the
a
form
finding
canonical
is due
should
canonical
given
and
closed
around
of s p e c i a l
time
major
the
equations.
Peter
Lancaster
explicit the
the
solutions
system
numerically
to
a
demanding
procedure. In
the
third
finds
two
main
where
the
error
direction, approachesin
imized.
For
referred
to
al
Bartels
[35],
based This
on
details the
widely A.
work et
reducing
approach
Bartels
and
the
is
methods
this
of
K.
HIGHLIGHTS
this for
presented.
first
Zietak
A matrix most
is
one
iterative
successively
min-
the
reader
is
[11-14],
Peters
et
approach
is
The
second
to
a lower
This
here
THE
one
reliable
[36].
OF
is
solutions,
approach,
[36] .
u s e d and is g i v e n
LYAPUNOV
In
of
the
Stewart
The
solution
al
the
numerical
Schur
and
approach
is is
form.
due the
to
most
in detail.
SOLUTION
OF
THE
EQUATIONS
section
a
detailed
the
solution
The
first
of the
method,
due
discussion Lyapunov
of
three
equations
to B a r n e t t
is
and Story
266
AHMAD A. MOHAMMAD AND J. A. DE ABREU-GARCIA
[15],
is
method
based
is due
on
the
to B a r t e l s
m e t h o d is due to S.J.
1.
As
Solution via
was
Barnett
This
using
the L y a p u n o v
+ ATp
=
The
[36].
second
The
third
[3] .
Equations
Products
earlier,
to s t u d y the t h e o r e t i c a l
PA
Stewart
Lyapunov
Kronecker
Story.
Consider
and
product.
Hammarling
of
mentioned
and
Kronecker
this
method
is
aspects
method
is
normally
due
utilized
of the solution.
equation
-Q,
Kronecker
to
(34)
products,
this
equation
can be w r i t t e n
as [(AT| where
+
vec(A)
transpose
(I|
T) ] v e c ( P )
defines
of the
A|
=
the v e c t o r
rows of A,
a11B a12B
--"
-
vec(Q),
f o r m e d by
(35)
stacking
alnB ,
......
the
and
. amlB
=
(36)
amnB
a s s u m i n g A is mxn. Equation A has
no
solution vec(P)
(35)
has
eigenvalues
a unique on
the
solution imaginary
if and axis
only
[15],
if the
is g i v e n by =
-
[ (AT|
+
(I|
T) ] - i v e c ( Q ) .
(37)
CONTINUOUS AND DISCRETE TIME LYAPUNOV EQUATIONS
It
should
inversion
be
noted
that
of an n2xn 2 matrix.
is not u s e d for actual 2.
This
Bartels
is
one
and
of
popular MATLAB
AP
Stewart
the
most
is why this m e t h o d
Algorithm
numerically
For e x a m p l e
the
more
=
reliable
it is u s e d
and
in the
general
case
of
the
Lyapunov
-Q,
(38)
Q is symmetric.
lution
the
lets c o n s i d e r the e q u a t i o n
+ PB
where
This
involves
s o f t w a r e package.
consider
equation,
solution
calculations 9
widely used algorithms. To
the
267
if and
only
Equation
(38) has
if any e i g e n v a l u e s
a unique
[~i of
A
so-
and
~j
of B s a t i s f y (Xi + To
~j ~
solve
0 Eq.
for (38)
g e s t e d the f o l l o w i n g i)
Reduce
the
all
=
uTAu
=
and
for
P,
steps
9
matrix
unitary transformation
A
i
to
a
j.
(39)
Bartels
lower
and
Schur
Stewart
form
sug-
using
a
U as follows
Azl
0
.--
0
A21
A22
".
.
_ Apl
Ap2
---
App
where each m a t r i x Aii is at most
,
_
2x2.
(40)
268
2)
A H M A D A. M O H A M M A D AND J. A. DE A B R E U - G A R C I A
In
Schur
B
same
form
as
=
where 3)
the
V
The
vTBv
is
fashion,
B
B11
BI2
.--
Blq
0
B22
.--
B2q
0
-..
0
Bqq
=
also
reduce
a unitary
transformed
Q and
to
upper
triangular
(41)
_
matrix 9 P are
given
by
N
Q11 Q
=
uTQv
......
Q1q (42)
=
_
QpI
......
Qpq
P11
......
Plq
_
and
P
=
uTpv
(43)
=
Ppl 4)
Recursively,
(k=l,2,...,p; It
should
be
solve
...... for
the
Ppq _ blocks
of P
as
k-i
L-I
j=l
i=l
follows
1= 1,2,...,q). noted
that
the
(44) solution
of
Eq.
(44)
is
CONTINUOUS AND DISCRETE TIME LYAPUNOV EQUATIONS
easy to carry mension
out
of 2x2.
since One
ber of c o m p u t a t i o n s B=A T.
This
Schur
should
a m a x i m u m di-
also notice
that
is r e d u c e d by almost
the num-
one half when
is needed.
of
the
computational
decomposition
(2+4g) n 3 flops,
and
where
tions
for
The actual
the
is one
due to
addition number
form
reduction
of Eq.
to
be
average
Schur
solution
is
estimated
a flop
and g is the
required
burden
is
multiplication, verge.
have
is due to the fact that only one Schur de-
composition Most
all blocks
269
the
about
and
of
one
iterato
(44) requires
conabout
7n 3 flops.
3. S.J.
Hammarling Hammarling
Algorithm
1982,
modified
the
Bartels/Stewart
a l g o r i t h m such that the solution to the L y a p u n o v equations
is
Hammarling directly
more was
numerically
able
for the
to
upper
solve
well the
conditioned.
Lyapunov
triangular
Cholesky
equations factors
of
the Grammians.
B.
SOLUTION
OF
THE
DISCRETE
LYAPUNOV
EQUATIONS As for
in the
the
possible system. reader
continuous
discrete
time
for specific For
details
time
Lyapunov
canonical of
is r e f e r r e d to N.J.
and references
case,
this
equations
solutions are
representations type
Young
t h e r e i n such as
explicit
of
of the
solutions,
[9], K e q q i a n WU
[38].
only the [37],
270
AHMAD A. MOHAMMAD AND J. A. DE A B R E U - G A R C I A
It s h o u l d be n o t e d that discrete ment
time
The
discrete
common
time
continuous rithms
procedure
efficient
practice
Lyapunov
time
can
nature
Lyapunov equation prevented
of n u m e r i c a l l y
tion.
the n o n l i n e a r
be
to
the d e v e l o p for the solu-
in n u m e r i c a l l y
equation
one w h e r e used
algorithms
solving
is to convert
efficient obtain
of the
the
it to a
and r e l i a b l e
algo-
solution.
This
the
is o u t l i n e d next.
Consider
the
discrete
Lyapunov
equation
given
by
[32,33] ATLA
-
L
=
- Q,
(45)
u s i n g the t r a n s f o r m a t i o n A
=
(B+I) (B-I)
converts
the
continuous BLb
+
discrete
LbB
=
(A-I)
it is
conditioned. for
the
tance.
=
(B_I)T
time
Lb 2 (B-I) ,
Lyapunov
(46)
equation
to
the
(47)
(47)
- Q. can
be
or
S.J.
Bartels/Stewart of
L
time L y a p u n o v e q u a t i o n
Equation However,
-I ,
should
be
solution
of
for
out
demanding
the d e v e l o p m e n t this
Lb
using
H a m m a r l i n g 's
pointed
numerically Thus,
solved
problem
that and
either
algorithm.
the
inversion
might
be
ill
of a new t e c h n i q u e is
of
great
impor-
CONTINUOUS AND DISCRETE TIME LYAPUNOV EQUATIONS
C.
A
NOTE
ON
THE
DESCRIPTOR
SOLUTION
OF
Ex
= Ax
y =
EQUATIONS
system
+ Bu
(48a)
Cx,
where
(48b) -I
(sE-A)
condition The
THE
LYAPUNOV-LIKE
Consider the regular d e s c r i p t o r
exist
as
a
sufficient
and
necessary
for regularity.
expected
Grammians
controllability
for this
and
system satisfy
observability
[39]
A W c ET
+ E W c AT =
- BB T
(49)
ATWo E
+ ETWo A
- C TC.
(50)
Under
the
=
assumption
consistent
initial
eliminate
any
that
the
conditions
system
is
(initial
impulsive
behavior),
given theorems
concerning
reachability,
and
similar
stability
However,
the
unique.
In addition,
ness
with
of the
Grammians to
the
corresponding
divided to
these 1987
about
part the
of
with
conditions
that
Lewis
regular
systems.
equations [40],
the
is
not
has shown that
in general.
existence he has
two parts, the
noncausal
system part
has
observability,
He also
for these Grammians
Briefly, into
regular
Frank
for
satisfied
solution. causal
of
definition
theorems
are
those
Bender
are not
gave a more general gether
to
solution
these equations
ing
271
and
shown one
uniquethat
the
correspond-
and of
to-
the
the
other
system.
272
A H M A D A. M O H A M M A D AND J. A. DE A B R E U - G A R C I A
According
to
Bender,
the
teachability
Grammians
are
given by
r (AWc~ E T + ~.wo~A ~) r r
= - CoBB Tr T
(~.wo2~. ~ - Aw~2A =) r
while the o b s e r v a b i l i t y
= r162
are e x t r e m e l y applications tions.
izations, technique solution
and
the
one
hence,
to obtain to this
(54)
the coefficients
series expansion solutions This
require
For e x a m p l e
(53)
= CT-~CTCr
difficult. that
Tcr
respectively,
s o ,s -I in the Laurent
that
are given by
= - r
CT_~(~.TWo2~. - ;JWo2A) r
It is clear
(52)
Grammians
r (;JWo~. + ~.TWo~A)r
where ~-I and ~0 are,
(51)
the
can
specific
for these
solution
not
reduced
of H(s).
is a m a j o r
can not
of
obtain utilize
equations
drawback
in
of these
equa-
balanced
real-
the
order models.
balancing However,
p r o b l e m has been p r o p o s e d
a in
[2].
This
concludes
the
review
of
the
solutions
to
the
Lyapunov equations. IV.
IMPORTANT
APPLICATIONS
LYAPUNOV
Different have
already
OF
THE
the
Lyapunov
EQUATIONS
applications been
of
introduced
However,
it is
felt
that
of these
applications
equations
in the p r e v i o u s
a more
is n e e d e d
comprehensive for the
sake
section. summary of clar-
CONTINUOUS AND DISCRETE TIME LYAPUNOV EQUATIONS
ity and completeness
of this
of these applications A.
LYAPUNOV
These
Consider =
theory
represent
and
a summary
here.
+ Bu,
the
in control stable
x(0)
=
first
application
system design
[41].
system given by (55 .a)
x0,
= Cx.
(55.b)
It is desired to design uTu
Thus,
CONTROLLERS
the minimal
Ax
y(t)
is p r e s e n t e d
controllers
of Lyapunov's
chapter.
273
a control
input u such that
___ 1
such
(56)
that
the
initial
state
returns
problem
starts
to the
origin
as rapidly as possible. The
solution
Lyapunov V
xTpx
=
x T[PA that
symmetric unique PA
+ ATp]x A
is
symmetric
Substituting
The
by
assuming
a
(57)
=
control
2uTBTpx. which
(58)
implies
semi-definite
positive
definite
Q,
that
there
for
any
exists
(59)
(59) into Eq. +
input
a
P such that
-Q.
Eq.
- xTQx
+
stable
positive
+ ATp
=
this
function
=
Notice
to
(58) yields
2uTBTpx. u
should
(60) be
chosen
.such
that
V
is
274
A H M A D A. M O H A M M A D AND J. A. DE A B R E U - G A R C I A
negative
with
the
largest
magnitude
clear that u must be parallel two
vectors
must
have
ond term negative.
to BTpx.
opposite
signs
It
However, to make
It
is
the
sec-
Thus u should be taken as
known
(61)
important
optimal LQR
is
these
BTpx
u
the
possible.
to
notice
controller
design
that
among
gives
only
this
all the
controller
controllers. linear
is The
optimal
con-
troller. B.
LYAPUNOV
EQUATIONS
ANALYS
This
has already
beginning
of
time
descriptor
earlier. suffer
It
especially responding transfer
chapter.
was
some
the
that
to
at the
discrete
also
addressed
these
extensions
problems.
descriptor
case
into two
This
is
where
the
subsystems
and p o l y n o m i a l
parts
cor-
of the
function.
LYAPUNOV
EQUATIONS OF
Controllability, ity analysis and
the
proper
MINIMALITY
have
out
to be s e p a r a t e d
to
addressed
were
serious in
been
Extensions
systems
pointed
evident
system needs
C.
this
from
STABILITY
IS
application
and
IN
already
implications
THE
systems
addressed of
these
THE
SYSTEM
observability,
of dynamic been
DEFINE
and
hence
via Lyapunov
earlier.
concepts
The in
minimalequations
importance
control
system
CONTINUOUS AND DISCRETE TIME LYAPUNOV EQUATIONS
design the
were also
considered
applications
sign,
observer
lems
have
noted
design,
been
that
of these
concepts
and
discussed
the
in details
of
controller
However,
these
systems
suffers
problems.
mainly
arise
tion
and
latter
the
two
problems
solution
cases.
of
Lyapunov
These
problems
de-
probit
concepts
crete time and d e s c r i p t o r These
Among
realization
in detail.
application
therein.
the
minimal
275
was
in dis-
from serious
in the
defini-
equations have
in
been
the
briefly
introduced earlier. D .
LYAPUNOV
EQUATIONS
IN
MODEL
ORDER
REDUCTION
B.C. the
Moore
[21]
demonstrated
controllability
'balanced was
used
der m o d e l s Lyapunov
constitute
and o b s e r v a b i l i t y in the
same paper
of d y n a m i c a l
equations
and is essential tion
observability
coordinates,
controllability fact
and
algorithm
how the eigenvalues a measure
to obtain This
introduced
later;
the This
reduced
application
the most
in the d e v e l o p m e n t
in
of
of each state.
systems.
is p r o b a b l y
Grammians,
of
important
orof one
of the continuizahence
it
is
intro-
duced here in detail. 1.
Since
The
Technique
its i n t r o d u c t i o n
technique
has
[42-53].
By
niques
Balancing
triggered now,
(BT)
by B.C. Moore, an
there
to obtain b a l a n c e d
intensive
are
several
realizations
the b a l a n c i n g
wave
of
research
numerical
tech-
[see references
276
AHMAD A. MOHAMMAD AND J. A. DE ABREU-GARCIA
above] . rather
However,
than the
they
only
concept.
differ
An o u t l i n e
in
the
of this
details technique
is g i v e n next. Consider
y
the
system
[50,52,53]
=
Ax
+
Bu
(62a)
=
Cx
+
Du,
(62b)
and the c o r r e s p o n d i n g AWc
+ Wc AT
ATw o + Next,
consider
Wc = where
WoA
Lyapunov
equations
=
-BBT
(63)
=
-cTc.
(64)
the s i n g u l a r
value decomposition
of Wc
U c ~ c U T,
Uc
is
(65)
unitary,
~c
is
diagonal
with
entries
(;i~(Yi+ 1 - Let TI and
=
_
~i/2
UcLc
apply
a
,
(66)
similarity
transformation
to
system
(62)
to get
-1/2
A1
=
Zc
CI
=
CUcLc
Wcl
=
The next
Wol
I,
,.,1/2
.T..
Uc~UcZ~c
Wol
=
T 2 as
=
-1/2
Zc
1/2 T..... .., 1/2
~c
UcWoUc~c
step is to p e r f o r m
= UoiZoiUoT1-
Choosing
, B1
uTB, (67)
.
(68)
an SVD on Wol as (69)
277
CONTINUOUS AND DISCRETE TIME LYAPUNOV EQUATIONS
= Uo Zo /' and
applying
system
(67)
a
(70) similarity
gives
transformation
the b a l a n c e d
using
T 2 to
system
_i14 T
Ab
-
/~oi U o l A I U o l ~ o I14 1 ,
Bb
=
~I/4UolB~ ir 2~oi
Cb
(71a)
_-I14
CiUol2~ol
=
,
(71b)
with the new G r a m m i a n s Wcb
=
_I14 T i14 Lol U o l l U o l ~ c
Wob
=
Lol
It mal
_-I14
should
and
-iI~
uTIWoIUoIEol
be
output
tained
from
_I12 ~oI =
=
noted normal
Eq.
i/2 Eol =
=
that,
the
by
E,
well
realizations
(71)
(72)
~, (~i>_(~i+l .
known
can be
applying
the
(73)
input
nor-
readily
ob-
similarity
transformations Ti
T 2 T IE ~/2
=
for input To =
normal,
nique
and
T 2 T I ~ -I/2
for output The
(74)
normal.
final is
(75)
to
step
in this
look
for
model
a break
order in
the
reduction singular
techvalues
such that (~r>>(~r + I ' and t r u n c a t e
the b a l a n c e d
system
after the
r th state.
278
A H M A D A. M O H A M M A D AND J. A. DE A B R E U - G A R C I A
To
illustrate
system
the
last
step,
:[A11 A12][xl
x2
A21
A22
bl
the
balanced
u (t)
b2
x2
~: [c,c.][x.]
consider
(76a) (76b)
X2
with
the G r a m m i a n s
Wc
then
=
the
'
~2
reduced
Xr
=
AllXr
Yr
=
CllXr,
with
I (Z~)>>k (Z2), min max
r th o r d e r m o d e l
(77)
can be t a k e n
as
+ bllU
(78a) (78b)
Granunians
Wcr For
o]
Wo =
-
Wor
=
simplicity,
renamed
as
2.
-- ~'r
the
(79)
reduced
order
model
matrices
are
[A r,B r,C r] .
Properties
of
the
Technique
a)
The
entries
of
the
system.
between
~i
them
t r i x H of the
of ~
are
There
a n d the system-
Balancing
[44, 4 5 , 5 2 ]
called is
an
singular
the
second
interesting values
Specifically,
order
modes
relationship
of the
Hankel
ma-
CONTINUOUS AND DISCRETE TIME LYAPUNOV EQUATIONS
279
2 k i ( H H T) = ~ib)
In general,
and the
the
eigenvalues
controllability
of the
level
of
of
the
Grammians
controllability
are
and
observability a quantization
observability
of
each state. c) In balanced coordinates, as it is observable d) The subsystems
a state is as controllable
(Wo=Wc=~).
[Aii,Bi,C i] in Eq.
(76) are balanced
with Woi = Wci = ~i-
e) T r a c e ( W o o ) = ]~ D C 2
gain
f)
bound
Wco = C r o s s There
is
an
of
the
system,
Grammian. upper
on
the
frequency
error
given by ]E ( s ) ~ C g)
The
(s I-A) -IB-c I (s I-All )-IB iI~-(~j i f
i>j.
and b a l a n c e d the
input
The
system.
and output
286
AHMAD A. MOHAMMAD AND J. A. DE ABREU-GARCIA
matrices
of the CTS Bcb,
Bcb =
Bdbl~,
This
choice
Euler's
method,
trices CTS
except
Ccb can be taken as
Ccb = is
(83)
Cdb/~.
justified
for example,
if
one
does
not c h a n g e
for the m u l t i p l i c a t i o n
dynamics
matrix
Acb
is then
notices
these ma-
factor
chosen
to
that
~.
The
satisfy
the
CTS L y a p u n o v e q u a t i o n s
so
Acb%
+
~ATb
=
-- P / T - -
ATb ~
+ ~Acb
=
-
that
the
DTS.
method
and the
and
output
while
in
bilinear
the
noticing
or Eq.
transform one
same
of
the
combination
of
t r a n s f o r m method-
the
is
are
Grammians a
taken
as
values
method.
It
is
Acb
separately,
the
solution
if
one
uniqueness
of
there are two s o l u t i o n s Then
solves the
from
for
the s o l u t i o n
Euler's taken
as
worthwhile either will
Acb
Eq.
not
from
be
Eqs.
is unique.
solution,
Acb I and Acb 2 that
it follows
in are
for
simultaneously,
(84)-(85).
the
solves
(84)-(85)
Eqs.
(85)
singular
However,
show
Qc,
bilinear
unique. To
(84)
strategy
Hankel
if
(85)
-
matrices
the
that
=
have
this
method
(84)
will
Clearly,
Euler's input
CTS
Q/T
Pc
suppose
that
satisfy both
that
Acbl%
+
ZAcTbl =
-- Pc
(86)
Acb2~
+
T ~Acb2
- Pc-
(87)
Subtracting
Eq.
=
(87) f r o m Eq.
(86) and l e t t i n g
CONTINUOUS AND DISCRETE TIME LYAPUNOV EQUATIONS
Acb I - Acb 2 =
287
N
yields N]~
+
]~N T =
This e q u a t i o n
reduces
(~jnij + (Yinji Similarly, must
(88)
O.
-
f r o m the
which
+
NT~
=
observability
Lyapunov
equation,
reduces
0,
N
(90)
to
(~inij + nji(~j = Comparing
Eqs.
(~i~(~j i f
i~j,
nij
=
nji
and
is
seen
it
that
(91) , that
and
the
provided
only
solution
to
the
Acb 2.
solution
solution
of
of Eqs. these
two
(84)-(85),
[61]
9
Provided
i~j,
Acb can be w r i t t e n
(~i~(~j i f
- Pcii 2(Yi
it
equations
simple p a r a m e t e r i z a t i o n
acbii =
to
is
that Acb I =
the
that
that
-- 0,
implies
Returning
(91)
0.
(89)
these two e q u a t i o n s
seen
(89)
0.
satisfy
~N
which
to
is
is
a
as (92)
288
A H M A D A. M O H A M M A D AND J. A. DE A B R E U - G A R C I A
for the d i a g o n a l
elements
choice
that
the
of b a l a n c e d
Clearly, (Acb,
elements
system
Ccb,
as
This
is due
and e x p l a i n s
to the
coordinates.
Dc=Dd)
it
of Acb.
is b a l a n c e d
from L y a p u n o v
Bcb,
balanced
(93)
2 2 ((~i -- (~j)
for the off d i a g o n a l fact
and
Pcijf~j -- qcij(~i
acbij =
the
of Acb,
theory,
is
satisfies
both both
the
resulting
stable, Lyapunov
system
minimal Eqs.
and
(84)
and
(86). 3.
This
Theoretical
is best
Theorem-
Aspects
summarized
For
any
given
(Adb, Bdb, Cdb, Dd, ~, T) , modes,
there
balanced
exists
CTS model
Bob = B and
Acb
/fT,
is
a
of
the
LBCT
in the following stable,
with
minimal
distinct
unique,
theorem-
balanced second
minimal,
order
stable,
(Acb, Bcb, Ccb, Dc=Dd, ~),
DTS and
where
Ccb = C b/f{,
the
unique
solution
of
the
CT
Lyapunov
equations
AcbZ
+
~ATb
=
T - BcbBcb
=
- TBdbBTb
T Acb~
+ ~Acb
=
-
T CcbCcb
=
- TC~Cdb.
the
CTS
Moreover,
the f o l l o w i n g
approximation
properties-
of
the
DTS
model
has
CONTINUOUS AND DISCRETE TIME LYAPUNOV EQUATIONS
i)
The
model 2)
CTS
model
is m i n i m a l
There
minimal
a
solution
if
the
DTS
for
Eqs.
(84)-(85),
for
a
of 2) is unique.
4) The I~2 n o r m of the DTS CTS
stable
of T, B, and C.
3) The s o l u t i o n the
and
and stable.
exists
fixed choice
is
289
(notice
that
we
error
in
is equal only
to the ~HI2 n o r m
deal
with
the
of
strictly
p r o p e r part) . 5)
The
initial
the
step
response
is
identi-
the
Hankel
cally zero.
Proof
:
(i) G u a r a n t e e d by L y a p u n o v equations. (2,3) (4)
Proved A
in s e c t i o n
direct
consequence
S i n g u l a r values (5)
First,
same value
it
p r o p e r part
4.
is
noted term
the
that
D.
is i d e n t i c a l l y
Comparison
with
matching
Euler's
both
systems
Moreover,
initial
of
share the
the
the
initial strictly
zero.
Other
Techniques
the p r o p o s e d
and
from
response
with
In order to v a l i d a t e parison
of
of the DT and CT systems.
feedthrough theorem,
2.
the
technique,
Bilinear
a com-
transform meth-
ods is p r e s e n t e d next. To model
set
the
stage,
suppose
and the c o n t i n u o u s
discretization In this
case,
technique the
first
model using
that
the
are r e l a t e d a
relation between
sampling the DTS
discrete
via Euler's period
T.
and the CTS
290
A H M A D A. M O H A M M A D AND J. A. DE AB R E U - G A R C I A
state
space m a t r i c e s
Adb =
I
+ TAcb,
Substituting (I
+
Eq.
Bdb =
(94)
TAcb)~(I
where
Wcb
will
be
anced' Thus, to
is
(95a)
BobBc b
(95b)
method,
must
satisfy
T Bcb Bcb,
-
that
Hankel
t e r m in Eqw.