Radio Monitoring
A volume in the Nanostructure Science and Technology series. Further titles in the series can be found at: http://www.springer.com/series/7818
Lecture Notes in Electrical Engineering Volume 43 Radio Monitoring: Problems, Methods, and Equipment Anatoly Rembovsky, Alexander Ashikhmin, Vladimir Kozmin, and Sergey Smolskiy 978-0-387-98099-7 Incorporating Knowledge Sources into Statistical Speech Recognition Sakti, Sakriani, Markov, Konstantin, Nakamura, Satoshi, and Minker, Wolfgang 978-0-387-85829-6 Intelligent Technical Systems Martínez Madrid, Natividad; Seepold, Ralf E.D. (Eds.) 978-1-4020-9822-2 Languages for Embedded Systems and their Applications Radetzki, Martin (Ed.) 978-1-4020-9713-3 Multisensor Fusion and Integration for Intelligent Systems Lee, Sukhan; Ko, Hanseok; Hahn, Hernsoo (Eds.) 978-3-540-89858-0 Designing Reliable and Efficient Networks on Chips Murali, Srinivasan 978-1-4020-9756-0 Trends in Communication Technologies and Engineering Science Ao, Sio-Iong; Huang, Xu; Wai, Ping-kong Alexander (Eds.) 978-1-4020-9492-7 Functional Design Errors in Digital Circuits: Diagnosis Correction and Repair Chang, Kai-hui, Markov, Igor, Bertacco, Valeria 978-1-4020-9364-7 Traffic and QoS Management in Wireless Multimedia Networks: COST 290 Final Report Koucheryavy, Y., Giambene, G., Staehle, D., Barcelo-Arroyo, F., Braun, T., Siris,V. (Eds.) 978-0-387-85572-1 Proceedings of the 3rd European Conference on Computer Network Defense Siris, V.; Ioannidis, S.; Anagnostakis, K.; Trimintzios, P. (Eds.) 978-0-387-85554-7 Data Mining and Applications in Genomics Ao, Sio-Iong 978-1-4020-8974-9, Vol. 25 Informatics in Control, Automation and Robotics: Selected Papers from the International Conference on Informatics in Control, Automation and Robotics 2007 Filipe, J.B.; Ferrier, Jean-Louis; Andrade-Cetto, Juan (Eds.) 978-3-540-85639-9, Vol. 24 Continued after index
Anatoly Rembovsky · Alexander Ashikhmin · Vladimir Kozmin · Sergey Smolskiy
Radio Monitoring Problems, Methods, and Equipment
123
Anatoly Rembovsky JSC IRCOS Staroalexeevskaya str. 14 Moskva Bldg. 2, Apt. 133 Russia 129626
[email protected] Alexander Ashikhmin JSC IRCOS Staroalexeevskaya str. 14 Moskva Bldg. 2, Apt. 133 Russia 129626
[email protected] Vladimir Kozmin JSC IRCOS Staroalexeevskaya str. 14 Moskva Bldg. 2, Apt. 133 Russia 129626
[email protected] Sergey Smolskiy Department of Radio Receivers Technical University Moscow Power Engineering Institute Lefortovskly Val ul, 7, Apt. 66 Moskva E-116 Russia 111116
[email protected] ISBN 978-0-387-98099-7 e-ISBN 978-0-387-98100-0 DOI 10.1007/978-0-387-98100-0 Springer Dordrecht Heidelberg London New York Library of Congress Control Number: 2008943693 © Springer Science+Business Media, LLC 2009 All rights reserved. This work may not be translated or copied in whole or in part without the written permission of the publisher (Springer Science+Business Media, LLC, 233 Spring Street, New York, NY 10013, USA), except for brief excerpts in connection with reviews or scholarly analysis. Use in connection with any form of information storage and retrieval, electronic adaptation, computer software, or by similar or dissimilar methodology now known or hereafter developed is forbidden. The use in this publication of trade names, trademarks, service marks, and similar terms, even if they are not identified as such, is not to be taken as an expression of opinion as to whether or not they are subject to proprietary rights. Printed on acid-free paper Springer is part of Springer Science+Business Media (www.springer.com)
Preface
Automated radio monitoring (ARM) technology obtained wide distribution as a tool for problem-solving in various areas, beginning from radio frequency spectrum usage control to the use of radio environment checks to search for illegal radio transmitters. Radio monitoring equipment serves as the basis of technical measures for counteracting unapproved information pick-up, including the all-important investigation of compromising emanations. The list of problems solved with the help of ARM equipment includes: – Revelation and analysis of radio emissions, for the identification of signal and interference sources, – Measurement of radio emission parameters, and the estimation of their danger or value for the user, – Electromagnetic field strength, or the power flow density measurement, – Radio signals and interference direction-finding in the terrain. In particular, ARM equipment allows radio engineering facilities and computer hardware to be checked for the presence and level of incidental emanations. As such, the main functions of ARM equipment are the permanent or periodic observations of airwaves in the wide frequency range, the effective detection, analysis and localization of potential or specially-organized channels of information drain. Based on the authors’ development experience, fundamental information concerning the described ARM systems, reference data, and recommendations on the best methods and approaches for obtaining solutions to the above-mentioned problems are included in the book, together with the classification and detailed description of modern high-efficient hardware-software ARM equipment, including equipment for detection, radio direction-finding, parameters measurement and their analysis, and the identification and localization of electromagnetic field sources. Examples of ARM equipment structure and application, within the complicated interference environments found in industrial centers, inside of buildings, and in the open terrain, are included, together with the software required for such applications.
v
vi
Preface
The book is prepared on the basis of Russian and foreign publications and as a result of various research and implementation activities of IRCOS1 company experts, under the supervision and direct participation of the authors. The book contains 12 chapters. In Chapter 2, the list of problems solved by ARM systems is discussed in detail. An analysis of the nomenclature, structure, functions and parameters of ARM equipment is performed, and the system hierarchy of the facilities is developed. The composition, the functions, and the main technical characteristics for each class of equipment are determined. Chapter 3 is devoted to the basic parameters of up-to-date radio receivers affecting ARM problem fulfillment. The peculiarities of the digital receiver structure for the 9 kHz – 18 GHz frequency range are shown. Design examples and the characteristics of single-channel and double-channel digital receivers are discussed. Chapter 4 is dedicated to the mathematical aspects of narrow-band signal detection, as well as the signals with dynamic frequency-time distribution (with frequency hopping) for single- and double-channel radio equipment. ARM problem-solving via multi-channel panoramic digital receivers is analyzed in Chapter 5, together with the hardware and software structure peculiarities of these receivers and their main technical data. Chapters 6 and 7 are devoted to the radio signals used in communication, broadcasting, TV and data transmission systems, and to the technical analysis and parameter measurement of modulated and non-modulated signals. Examples of radio signal parameter measurement are discussed and recommendations for software applications are given. A review of and the theoretical bases for direction-finding methods are presented in Chapter 8, and the main parameters of radio direction finders are explained. Examples of multifunctional radio monitoring and direction-finding equipment in VHF, UHF, and microwave ranges are described. The affect of used digital receivers on direction-finding effectiveness is shown. Chapter 9 is devoted to the development of geographically-distributed radio monitoring systems and to direction-finding systems for radio emission sources. The application of stationary, mobile, portable and hand-held ARM equipment is considered. Moreover, the problems related to ARM station system equipment, organization of data transmission through the communication, navigation and power supply channels, are considered in this chapter as well. The possible uses of software for signal detection, their parameter measurement, and direction-finding of radio emission sources – with positions indicated on an electronic map – are discussed. Chapter 10 includes information on determining the position of radio emission sources by mobile radio monitoring stations, and estimation of field strength distribution, taking into account terrain relief and area reclamation, to obtain covering zones of broadcasting and communication. Solutions to the problems of
1 IRCOS
means: Investigations on Radio Control and System design
Preface
vii
electromagnetic compatibility and the parameters testing of radio electronic equipment are discussed also. Chapter 11 describes the structural peculiarities of radio monitoring equipment inside the premises and the revelation of technical channels of information leakage and unapproved radio emission sources. Revelation methods are discussed, together with these source localization methods on checked objects. Implementation examples for hardware-software facilities for technical channel leakage revelation, used both inside the premises and on the boundary of the checked zone, are presented. In Chapter 12, the problems of radio system structure in performing compromising emanations investigation are considered. The theoretical aspects and the practical approaches for the revelation of the informative components are discussed, with calculation of the checked area and object immunity radii. The equipment and the software examples for these investigations are given. The authors are confident that the materials offered in the book will be useful to experts in the area of radio monitoring, to operators and leaders of civil and military radio-checking services, and to security service employees of both state and commercial structures. The book can be recommended to the students of technical universities and colleges, studying in the appropriate fields.
“This page left intentionally blank.”
Contents
1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
1
2 Problems, Classification and Structure of ARM Equipment Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . Classification of Radio Monitoring Equipment . . . . . . . . . Operation Zone Sizes . . . . . . . . . . . . . . . . . . . . . Application . . . . . . . . . . . . . . . . . . . . . . . . . . Equipment Performance . . . . . . . . . . . . . . . . . . . . Design Constraints . . . . . . . . . . . . . . . . . . . . . . Radio Monitoring Equipment Design Philosophy . . . . . . . . Requirements for RM Equipment Technical Parameters . . . . Quality Criterion Selection . . . . . . . . . . . . . . . . . . Main Technical Parameters of RM Equipment . . . . . . . . Characteristics of RM Equipment Families . . . . . . . . . . . Radio Monitoring and RES Location Detection Systems . . . Stationary and Mobile RM Stations . . . . . . . . . . . . . . Portable RM Equipment . . . . . . . . . . . . . . . . . . . Manpack ARM Equipment . . . . . . . . . . . . . . . . . . Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . References . . . . . . . . . . . . . . . . . . . . . . . . . . . .
. . . . . . . . . . . . . . . . . .
. . . . . . . . . . . . . . . . . .
. . . . . . . . . . . . . . . . . .
. . . . . . . . . . . . . . . . . .
5 5 6 7 7 8 8 9 12 12 13 16 16 16 18 19 21 22
3 Radio Receiver Applications for Radio Monitoring System Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . Tuned Radio Receiver . . . . . . . . . . . . . . . . . . . . . Main Radio Receiver Parameters . . . . . . . . . . . . . . . Operating Frequency Range . . . . . . . . . . . . . . . . Amplitude-Frequency Response of the Linear Receive Path Voltage Standing Wave Ratio . . . . . . . . . . . . . . . . Main Channel and Spurious Channels . . . . . . . . . . . RR Selectivity . . . . . . . . . . . . . . . . . . . . . . . . Inherent Noise and Receiver Sensitivity . . . . . . . . . . Sensitivity Increase with the Help of Pre-amplifiers . . . . Pre-amplifier Gain Factor Selection . . . . . . . . . . . . . Receiver Multi-Signal Selectivity . . . . . . . . . . . . . .
. . . . . . . . . . . . .
. . . . . . . . . . . . .
. . . . . . . . . . . . .
. . . . . . . . . . . . .
23 23 23 27 28 28 29 30 34 36 39 42 43
. . . . . . . . . . . . .
ix
x
Contents
Intermodulation Noise . . . . . . . . . . . . . . . . . . . . Intercept Points on IP2 and IP3 Intermodulation . . . . . . . Intermodulation-Free Dynamic Range Determination . . . . Attenuator Influence on the Intermodulation Value . . . . . . Determining the Intercept Points . . . . . . . . . . . . . . . Blockage Effect . . . . . . . . . . . . . . . . . . . . . . . . Crosstalk Distortions . . . . . . . . . . . . . . . . . . . . . Phase Noise and Retuning Rate of the Panoramic RR . . . . Digital Radio Receivers . . . . . . . . . . . . . . . . . . . . . General Principles of Digital Radio Receiver Implementation Types of ARM Receivers . . . . . . . . . . . . . . . . . . . Development of Russian Arm Systems . . . . . . . . . . . . . First- and Second-Generation Systems . . . . . . . . . . . . Radio Receivers of the Third and Fourth Generation . . . . . Fifth-Generation Radio Receivers . . . . . . . . . . . . . . ARK-CT1 Digital Radio Receiver . . . . . . . . . . . . . . . . ARK-D1TP Digital Panoramic Measuring Receiver . . . . . . ARK-CT3 Digital Receiver . . . . . . . . . . . . . . . . . . . ARK-KNV4 External Remote-Controlled Converter . . . . . . ARK-PR5 “Argamak” Digital Radio Receiver . . . . . . . . . ARGAMAK-I Panoramic Measuring Receiver . . . . . . . . . Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . References . . . . . . . . . . . . . . . . . . . . . . . . . . . .
. . . . . . . . . . . . . . . . . . . . . . .
. . . . . . . . . . . . . . . . . . . . . . .
. . . . . . . . . . . . . . . . . . . . . . .
. . . . . . . . . . . . . . . . . . . . . . .
43 47 50 51 52 52 53 54 56 56 58 60 60 62 66 67 73 75 79 81 93 93 94
. . . . . . . . . . . .
95 95 97
4 Single-Channel and Multi-Channel Radio Signal Detection Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . Single-Channel Signal Detection . . . . . . . . . . . . . . . . Characteristics of Single-Channel Detection of Narrow-Band Signals . . . . . . . . . . . . . . . . . . . . . Single-Channel Detection of Radio Signals With POFT . . . . Probabilistic Features of the Frequency Observation Time . . Probability of Separate Frequency Registration . . . . . . . Estimate of the Total Number of Registered Frequencies . . Optimization of ARM System Parameters . . . . . . . . . . Detection Characteristics . . . . . . . . . . . . . . . . . . . Double-Channel Detection of Narrow-Band Signals . . . . . . Comparison of Single-Channel and Double-Channel Processing Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . References . . . . . . . . . . . . . . . . . . . . . . . . . . . .
. . . . . . . . . . .
. . . . . . . . . . .
. . . . . . . . . . .
. . . . . . . . . . .
105 108 109 111 112 113 115 117 119 121 121
5 Multi-Channel Digital Receivers . . . . . Introduction . . . . . . . . . . . . . . . . . Panoramic Multi-Channel Receivers . . . . ARK-D11 Double-Channel Complex . . . . ARK-RD8M Multi-Channel Complex . . . SMO-MCRM Customized Software Package
. . . . . .
. . . . . .
. . . . . .
. . . . . .
123 123 123 125 126 130
. . . . . .
. . . . . .
. . . . . .
. . . . . .
. . . . . .
. . . . . .
. . . . . .
. . . . . .
. . . . . .
. . . . . .
Contents
Software Purpose and Performance Capabilities Software Operation Modes . . . . . . . . . . . Conclusion . . . . . . . . . . . . . . . . . . . . . References . . . . . . . . . . . . . . . . . . . . .
xi
. . . .
. . . .
. . . .
. . . .
. . . .
. . . .
. . . .
. . . .
. . . .
. . . .
. . . .
130 131 133 133
6 Modulation and Signal Types in Modern Radioelectronic Means Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Administrative Division of the Frequency Spectrum . . . . . . . . . Modulation in Communication and Broadcast Systems . . . . . . . . General Information . . . . . . . . . . . . . . . . . . . . . . . . . Types of Analog Modulation . . . . . . . . . . . . . . . . . . . . Types of Discrete (Digital) Modulation . . . . . . . . . . . . . . . Signals of Modern Radio Electronic Means . . . . . . . . . . . . . . SW Range Signals (Less Than 30 MHz) . . . . . . . . . . . . . . VHF Range Signals (More Than 30 MHz) . . . . . . . . . . . . . International System for Signal Designation . . . . . . . . . . . . International Frequency Range Distribution . . . . . . . . . . . . . . Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
. . . . . . . . . . . . . .
135 135 135 138 138 140 147 158 158 165 182 188 191 192
7 Measurement of Radio Signal Parameters . . . . . . . . . . . . . Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Frequency Measurement . . . . . . . . . . . . . . . . . . . . . . . . Instantaneous Frequency Measurement Method . . . . . . . . . . FFT Method . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Measurement of Spectrum Width . . . . . . . . . . . . . . . . . . Determination of Modulation Type and Its Parameter Measurement . Determination of Modulation Type . . . . . . . . . . . . . . . . . Modulation and the Determination of Shift-Keying Characteristics SMO-STA Software for the Analysis of Automated Radio Signals . . STA Software Possibilities and Its Functional Diagram . . . . . . Examples of Radio Signal Modulation Type and Parameters’ Determination . . . . . . . . . . . . . . . . . . . . . . . . . . . . Automated Technical Analysis of Radio Signals . . . . . . . . . . . Unit for Automated Radio Signal Analysis . . . . . . . . . . . . . Peculiarities of SMO-PA Application . . . . . . . . . . . . . . . . Application of Automatic Signal Analysis in SMO-RD2 . . . . . . . Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
. . . . . . . . . . .
195 195 195 196 197 199 201 201 204 206 206
. . . . . . .
208 219 219 221 233 235 236
8 Direction Finding of Radio Emission Sources . . . . . . . . . . Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . History of Radio Direction-Finding Technique . . . . . . . . . . . Structural Diagram and Characteristics of Radio Direction Finders Main Technical Parameters of Radio Direction Finders . . . . . . . Accuracy of Direction Finding . . . . . . . . . . . . . . . . . .
. . . . . .
237 237 238 241 242 243
. . . . . .
xii
Contents
Sensitivity . . . . . . . . . . . . . . . . . . . . . . . . . Noise Immunity . . . . . . . . . . . . . . . . . . . . . . Operating Rate . . . . . . . . . . . . . . . . . . . . . . Resolution . . . . . . . . . . . . . . . . . . . . . . . . . Operating Frequency Range . . . . . . . . . . . . . . . Types of Being-Found Signals . . . . . . . . . . . . . . Deployment Time . . . . . . . . . . . . . . . . . . . . . Weight and Size . . . . . . . . . . . . . . . . . . . . . . Complexity in Manufacture and Operation . . . . . . . . Cost . . . . . . . . . . . . . . . . . . . . . . . . . . . . Classification of Direction-Finding Methods . . . . . . . . Systems Based on a Rotating Directional Antenna . . . . . ARK-RP3 Handheld Radio Direction Finder . . . . . . . . ARK-RP4 Handheld Radio Direction Finder . . . . . . . . Automatic Radio Compass . . . . . . . . . . . . . . . . . Automatic Radio Direction Finder with Low Antenna Base Doppler and Quasi-Doppler Direction Finders . . . . . . . Phase and Correlation Interferometers . . . . . . . . . . . . Peculiarities of Correlation Interferometer . . . . . . . . . Algorithm of Correlation Interferometer Measuring System Single-Channel Measuring System on the Basis of a Correlation Interferometer . . . . . . . . . . . . . . . ARTIKUL-M4 Foldable Correlation Interferometer . . . . ARTIKUL-M1 Mobile Direction Finder . . . . . . . . . . ARTIKUL-P Portable Foldable Direction Finder . . . . . . ARTIKUL-P11 Portable Foldable Direction Finder . . . . . Direction Finding Error Correction in Mobile Systems . . . Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . References . . . . . . . . . . . . . . . . . . . . . . . . . .
. . . . . . . . . . . . . . . . . . . .
. . . . . . . . . . . . . . . . . . . .
. . . . . . . . . . . . . . . . . . . .
. . . . . . . . . . . . . . . . . . . .
. . . . . . . . . . . . . . . . . . . .
. . . . . . . . . . . . . . . . . . . .
246 247 248 248 249 249 249 249 249 250 250 252 256 262 263 266 270 273 278 281
. . . . . . . .
. . . . . . . .
. . . . . . . .
. . . . . . . .
. . . . . . . .
. . . . . . . .
289 295 299 301 306 307 315 316
. . . . . . . . . .
317 317
. . . . .
318
. . . . . . . . .
320 321 322 322 323 326 326 328 328
9 Radio Monitoring Systems and Determination of Radio Emission Sources Location . . . . . . . . . . . . . . . . . . Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . Requirements for Radio Monitoring and Location Determination Systems . . . . . . . . . . . . . . . . . . . . Structure of the Radio Monitoring System and Determination of RES Location . . . . . . . . . . . . . . . . . . . . . . . . ARK-POM1 System . . . . . . . . . . . . . . . . . . . . ARK-POM2 System . . . . . . . . . . . . . . . . . . . . ARK-POM3 Geographically-Distributed System . . . . . Combined ARK-POM System . . . . . . . . . . . . . . . Control Arrangement in the System . . . . . . . . . . . . . . Data Exchange between Stationary Posts . . . . . . . . . . Data Exchange with the Mobile and Deployed Posts . . . . Peculiarity of the Low-Speed Radio Channel Application .
. . . . . . . . .
. . . . . . . . .
. . . . . . . . .
. . . . . . . . .
Contents
xiii
Usage of Radio Modems of the Cellular Communication Systems Data Exchange Implementation in Combined ARK-POM System . “Archa” Stationary Station . . . . . . . . . . . . . . . . . . . . . . . “Argument” Mobile Station . . . . . . . . . . . . . . . . . . . . . . System-Wide Car Equipment . . . . . . . . . . . . . . . . . . . . “Arena” Portable Station . . . . . . . . . . . . . . . . . . . . . . . . “Arena” Station Structure . . . . . . . . . . . . . . . . . . . . . . Mast Devices for Radio Monitoring Stations . . . . . . . . . . . . . Navigation Systems for Radio Monitoring Stations . . . . . . . . . . Features of Modern Navigation Systems . . . . . . . . . . . . . . Navigation Systems for Mobile Stations . . . . . . . . . . . . . . Electric Power Supply Systems . . . . . . . . . . . . . . . . . . . . Requirements for Electric Power Sources . . . . . . . . . . . . . Electric Power Sources for Radio Equipment . . . . . . . . . . . . Secondary Electric Supply Sources . . . . . . . . . . . . . . . . . Example of Pulse Power Supply of Low Power . . . . . . . . . . Multi-Channel Pulse Power Source . . . . . . . . . . . . . . . . . ARK-UPS12 Universal Power Supply Unit . . . . . . . . . . . . Autonomous Electric Station Usage . . . . . . . . . . . . . . . . Special Software Support and Operation Modes of Stations . . . . . Software Support Structure . . . . . . . . . . . . . . . . . . . . . “Spectrum” Mode . . . . . . . . . . . . . . . . . . . . . . . . . . “Search” Mode . . . . . . . . . . . . . . . . . . . . . . . . . . . “Bearing” Mode . . . . . . . . . . . . . . . . . . . . . . . . . . . “Measurement” and “Technical Analysis” Modes . . . . . . . . . “Review” Mode . . . . . . . . . . . . . . . . . . . . . . . . . . . “Multi-Channel Direction Finding” Mode . . . . . . . . . . . . . Peculiarities of the Direction Finding of POFT Stations . . . . . . “Electronic Map” Mode . . . . . . . . . . . . . . . . . . . . . . . Post-processing Mode . . . . . . . . . . . . . . . . . . . . . . . . Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10
Radio Emission Source Localization Using Mobile Stations and Field Strength Measurement . . . . . . . . . . . . . . . Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . Methods of RES Localization Using the Mobile Station . . . Drive Method . . . . . . . . . . . . . . . . . . . . . . . . . Quasi-Stationary Method . . . . . . . . . . . . . . . . . . . Method of Automatic Calculation of RES Coordinates During Movement . . . . . . . . . . . . . . . . . . . . . . . Peculiarities of Multi-Channel Direction Finding . . . . . . Simultaneous Direction Finding . . . . . . . . . . . . . . . Electromagnetic Field Strength Measurement . . . . . . . . . . Main Mathematical Relations . . . . . . . . . . . . . . . . .
. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
331 332 334 338 343 346 346 347 349 350 355 356 356 357 360 362 364 365 369 372 372 373 374 377 378 381 382 383 385 388 390 391
. . . . .
. . . . .
. . . . .
. . . . .
393 393 395 395 396
. . . . .
. . . . .
. . . . .
. . . . .
398 401 402 403 404
xiv
Contents
Peculiarities of the Field Strength Distribution Estimation . Field Strength Measurement . . . . . . . . . . . . . . . . On-Site Calculation of Field Strength Distribution . . . . . District Topography . . . . . . . . . . . . . . . . . . . . . Urban Build-Up . . . . . . . . . . . . . . . . . . . . . . . Vegetation Influence . . . . . . . . . . . . . . . . . . . . . Calculation of Field Strength in the SMO-KN Application Processing of Field Strength Measurements . . . . . . . . Determination of RES Location . . . . . . . . . . . . . . . Checking Transmitters for Announced Parameters . . . . . Calculation of Electromagnetic Compatibility . . . . . . . Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . References . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
. . . . . . . . . . . . .
. . . . . . . . . . . . .
. . . . . . . . . . . . .
. . . . . . . . . . . . .
Detection and Localization of Technical Channels of Information Leakage . . . . . . . . . . . . . . . . . . . . . . . . . Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Main Search Stages for Electromagnetic Channels of Information Leakage . . . . . . . . . . . . . . . . . . . . . . . . Detection of Radio Signals Emitted in Monitored Premise . . . . . . Radio Signal Intensity in Near-Field and Far-Field Regions . . . . Generalized Structure of Equipment for TCIL Detection . . . . . . Comparison Technique for Signal Intensities . . . . . . . . . . . . Detection Algorithm for Radio Signal Sources in Monitored Area Detection Effectiveness Dependence on the Equipment and the Ways of “Standard” Panorama Obtaining . . . . . . . . . Identification and Localization of Radio Microphones . . . . . . . . Distant Radio Monitoring Systems of Remote Premises . . . . . . . Construction Principles of Remote Radio Monitoring Systems . . Examples of Remote Radio Monitoring Systems . . . . . . . . . . Peculiarities of ARK-D3T Remote Radio Monitoring System . . . Peculiarities of the ARK-D9 Remote Radio Monitoring System . . Peculiarities of the ARK-D13 Remote Radio Monitoring System . Software for Remote Radio Monitoring Systems . . . . . . . . . . . Purpose and Possibilities of SMO-DX Application . . . . . . . . . Peculiarities of Radio Microphone Detection . . . . . . . . . . . . Joint Usage of the Various Detection Algorithms . . . . . . . . . . Radio microphone Localization Inside of Monitored Premises . . Equipment Operation in the Remote Radio Monitoring System . . Detection of TCIL Sources by the Mobile Station . . . . . . . . . . Antenna System Selection . . . . . . . . . . . . . . . . . . . . . Methods of RES Detection . . . . . . . . . . . . . . . . . . . . . Equipment Structure of ARTIKUL-M6 Mobile Direction Finder . Software Structure and Search Procedure Implementation . . . . . Aggregation of the Initial Data Frames . . . . . . . . . . . . . . .
. . . . . . . . . . . . .
406 407 408 409 411 412 414 415 419 421 423 424 424
. .
427 427
. . . . . .
428 430 430 432 433 435
. . . . . . . . . . . . . . . . . . . .
436 437 438 438 441 443 448 450 451 452 453 454 455 455 457 458 459 461 462 464
Contents
xv
Frame Processing and Generation of “Suspicious” Frequency List Checking the Frequencies from the List and More Precise RES Detection . . . . . . . . . . . . . . . . . . . . . . . . . . . . Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
.
465
. . .
467 468 469
. . . . . . . . . . . .
471 471 471 473 475 475 476 478 479 482 484 485
. . . . . . . .
486 488
. . . .
. . . .
490 492 496 496
Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
497
Subject Index . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
499
12
Methods and Equipment for Protection Against Information Leakage Via CEE Channels . . . . . . . . . . . General Information . . . . . . . . . . . . . . . . . . . . . . . Special Investigation Types and Information Security Index . . Calculation of Information Security Index . . . . . . . . . . . Estimation of the Testing Mode Parameters for a LCD Monitor Estimation of the Testing Mode Parameters for a CRT Monitor Methods of Detection of CEE Informative Components . . . . Probabilistic Features of Periodogram Samples . . . . . . . . . TDM Algorithm . . . . . . . . . . . . . . . . . . . . . . . . . Application of ARK-D1TI Measuring Complex . . . . . . . . Search of CEE Informative Components . . . . . . . . . . . Measurement of CEE Informative Component Intensity . . . Calculation of the Monitored Zone Radius by SMO-PRIZ Application . . . . . . . . . . . . . . . . . . . . . . . . . . Information Security Monitoring . . . . . . . . . . . . . . . . SMO-PRIZ Application Operation for Information Security Monitoring . . . . . . . . . . . . . . . . . . . . . . . . . . Purposes and Functions of SMO-THESIS Application . . . . . Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . References . . . . . . . . . . . . . . . . . . . . . . . . . . . .
. . . . . . . . . . . .
. . . . . . . . . . . .
. . . .
. . . . . . . . . . . .
. . . .
“This page left intentionally blank.”
Acronyms
AA ACS ADC AE AFH AFR AGC ALE AM AnM APM APSK ARI ARM ARME ARQ ARU AS ASK BP BPT BRPSK BRPT BS BWLL CB CDCS CDMA CE CEE CEEP CEPT CF CIM
- antenna array - amplifying-converting section - analog-digital converter - antenna element - automatic frequency hopping - amplitude-frequency response - automated gain control - automatic link establishment - amplitude modulation - angle modulation - amplitude-pulse modulation - amplitude-phase shift-keying - Autofahrer Rundfunk Information - automated radio monitoring - automated radio monitoring equipment - Auto ReQuest - antenna-receiver unit - antenna system - amplitude shift-keying - bearing pair - binary phase telegraphy - binary relative phase shift-keying - binary relative phase telegraphy - basing station - Broadband Wireless Local Loop - Citizen s Band - Continuous Dynamic Channel Selection - Code Division Multiple Access - consumer equipment - compromising electromagnetic emanation - compromising electromagnetic emanation and pick-up - Conference European for post and telecommunication - computing facility - Correlation interferometric meter (correlative interferometer) xvii
xviii
COFDM CP CPFSK CRC CRT CTF CW DAC DAM DARC DB DC DDM DECT DF DFT DPRS DPSK DRA DRM DRMS DRR DSBAM DSBSC DSP DSSS DVBT EBU EDGE EHF EMA EMC EMF EMW ETSI FCU FDMA FEC FFSK FFT FH FM FP FS FTD
Acronyms
- Coded Orthgonal Frequency Division Multiplexing - central post - continuous phase frequency shift-keying - Cyclic Redundance Check - cathode-ray tube - complex transfer factor - continuous wave - digital-analog converter - DAM modulation - Data Radio Channel - database - distant control panel - difference-distance measuring - Digital Enhanced Cordless Telecommunications - direction finding, direction finder - discrete Fourier transform - DECT Packet Radio Services - differential phase shift-keying - distributed random antenna - Digital Radio Mondiale - distant radio monitoring system - digital radio receiver - double sideband amplitude modulation - double sideband suppressed carrier - digital signal processing - Direct Sequence Spread Spectrum - Digital Video Broadcasting - European Broadcasting Union - Enhanced Data rates for Global Evolution - extremely high frequency - electromagnetic availability - electromagnetic compatibility - electromagnetic field - electromagnetic wave - European Telecommunication Standards Institute - frequency conversion unit - Frequency Division Multiple Access - Forward Error Correction - fast frequency shift-keying - fast Fourier transform - frequency hopping - frequency modulation - frequency position - frequency synthesizer - frequency-time diagram
Acronyms
FV GEG GFSK GIS GPRS GPS GSM GTC HFF HiperLAN ICAO IEEE IF IFM IMC INS IP ISSB ITA2 ITU LCD LF LMSK LNA LO LRA LSB LW MASK MF (UHF) MFSK MMDS MP MPC MSK MUSIC MW NB NFM NICAM OBW OFDM OOK OQPSK PBF
xix
- flying vehicle - gasoline electric generator - Gaussian frequency shift-keying - geo-information system - General Packet Radio Service - Global Positioning System - Global System for Mobile communications - gain-transfer characteristic - high-frequency filter - High Performance Local Area Network - International Civil Aviation Organization - Institute of Electrical and Electronic Engineers - intermediate frequency - instantaneous frequency measurement - intermodulation component - inertial navigation system - interception point - Independent Single Sideband - International Teleprinter Alphabet - International Telecommunication Union - liquid-crystal display - low frequency - frequency shift-keying with minimal shift and with level regulation - low-noise amplifier - local oscillator - lumped random antenna - lower sideband - long waves - multiple amplitude shift-keying - microwave frequency - Multiple frequency shift-keying - Multichannel Multipoint Distribution System - monitored premise - microprocessor control - Minimum Shift Keying - Multiple Signal Classification - medium waves - Normal Burst - Narrow Frequency Modulation - Near Instantaneous Companded Audio Multiplex - occupied bandwidth - Orthogonal Frequency Division Multiplexing - On/Off Keying - offset quadrature phase shift-keying - pass-band filter
xx
PC PM POFT PPM PS PSA PSF PSK PTA PWM QAM QASK QM QPSK RDS REE REM RES RF RFA RFS RMD RMS RO RPSK RPU RR RRMS RSS RTTY SA SAN SBD SCA SFH SG SGU SHF SMPS SMS SNR SPS SQPSK SR SRNS
Acronyms
- personal computer - phase modulation - programmable operating frequency tuning - phase-pulse modulation - phase-shifter - panoramic spectral analysis - power source filter - phase shift-keying - panoramic-technical analysis - pulse-width modulation - quadrature amplitude modulation - quadrature amplitude shift-keying - quadrature modulation - quadrature phase shift-keying - Radio Data System - radio electronic environment - radio electronic means - radio emission source - radio frequency - radio frequency amplifier - radio frequency spectrum - reference-methodical documentation - root-mean-square value (deviation) - reference oscillator - relative phase shift-keying - reception and processing unit - radio receiver - Remote Radio Monitoring System - reference spatial signal - Radio Tele Type - spectrum analyzer, space apparatus - system of active noisiness - spectral and bearing data - Sub-carrier Communication Allocation - Slow Frequency Hopping - signal generator - signal generation unit - super high frequency - switch-mode power supply - special mathematical software - signal/noise ratio - secondary power source - staggered quadrature phase shift-keying - Selective Repeat - satellite radio navigation system
Acronyms
SS SSBh SSBl SSBSC SV SW SWRV TCIL TCP/IP T-DAB TDM TDMA TDS TOI TTF UE UHF UMTS UPS URES US USB VHF VLF VSB VSWR WARC WCDMA WFM WLAN WMAN WPAN WTSC
xxi
- software support - single-side band (higher) - single-side band (lower) - Single Sideband Suppressed Carrier - space vehicle - short waves - standing-wave factor on voltage - technical channel of information leakage - Transport Control Protocol/Internet Protocol - Terrestrial Digital Audio Broadcasting - Testing and Detection Mutual - Time Division Multiple Access - Testing and Detection Separate - Third Order Intercept - tactical-technical features - user equipment - Ultra High Frequency - Universal Mobile - uninterrupted power supply - unwanted radio emission source - user station - Upper Sideband - very high frequency - very low frequency - Vestigal Side Band - Voltage Standing Wave Ratio - World Administration Radio Conference - Wideband-Code Division Multiple Access - Wide Frequency Modulation - Wireless Local Area Network - Wireless Metropolitan Area Network - Wireless Personal Area Network - World Telecommunication Standards Conference
Chapter 1
Introduction
Equipment for automated radio monitoring (ARM) can be considered an information extraction system. ARM equipment is widely used in various areas and was developed at the same rate as information transmission systems through radio channels. There are many problems which ARM can address: planned checking of regular equipment parameters, unpremeditated interference level measurement, detection and determination of non-licensed transmitter locations, measurement of energy covering zones during the estimation of radio communication quality, and determination of radio frequency resource usage intensity. ARM equipment also solves the problem of informational security. The increase in ARM equipment at present is caused by several reasons, the first of which relates to the continued technical progress of radio communication equipment, but the second relates to changes of an economical and political nature, which have occurred in the world. It should be noted that, in Russia, prior to 1992, radio frequency band loading, new frequency allocation, and radio frequency usage regulations were effectively and strictly controlled by the appropriate state agencies, including the security services of the various levels, and that strong restrictions on new radio communications equipment import and usage were simultaneously enacted. Under these conditions, ARM problems were effectively solved by existing and newly-developed native equipment. Standard modernization and replacement of equipment were fulfilled in planned order. An evident and rapid increase of ARM problems, in solving the tasks of radio monitoring and technical informational security, became apparent in Russia after 1992, due to the political and economical changes that occurred there. A strong reduction in the large number of radio electronic enterprises, which had earlier occupied leading positions in the development and manufacture of ARM equipment, caused leading-expert outflow from this field and, hence, a large reduction of modern ARM equipment delivery by these companies. This circumstance caused the slowing down of high-quality native ARM equipment delivery, both in product assortment and in the parameters and performance of the equipment. At the same time, in highly-developed, foreign countries, the radio monitoring equipment evolution advanced, as earlier, by an increasing rate, since the high effectiveness of radio electronic means (REM) for various types of information transmission – at A. Rembovsky et al., Radio Monitoring, Lecture Notes in Electrical Engineering 43, C Springer Science+Business Media, LLC 2009 DOI 10.1007/978-0-387-98100-0_1,
1
2
1
Introduction
constant cost price reduction – stimulated REM distribution greatly to all corners of the world. The fundamental complications of the radio electronic environment (REE) observed now in Russia (and typical, apparently, for the other countries), can be related to the following factors: – Increase in the number of regular TV and radio transmitters, introduction and further modernization of cellular communication systems and an increase in its usage intensity, a process which is far from complete; – Overloading of several regions of the radio range (e.g., sub-ranges 40, 100, 400, 800, and 2450 MHz), caused by a series of objective circumstances, such as the best conditions for radio wave propagation, no need to grant a license, etc.; – Permanent increase of REM operating range upper limit (at present to 18– 60 GHz), corresponding to the rapid development of modern technologies and instruments; – Application of various types of new waveforms: narrow-band with fixed frequency distribution, or with the dynamic frequency-time distribution of emission, and wide-band with code user division; – General tendency of REM transmitter power increase, dictated by the attempt to extend their action range, which is equivalent to the REM number increase acting in the receiver site point of the ARM equipment and leading to the unpremeditated interference level rise both at the main frequency and the harmonics; – Successful research on various receiver sensitivity giving rise to the necessity of an appropriate increase in ARM receiver sensitivity, required for reliable REE revelation and analysis. Additionally, the number of non-licensed radio emission sources (RES) with various power levels, and the large (over level and spectrum) number of spurious emissions not corresponding to the permissible norm and the international standards, increases permanently in the cities and industrial centers of many countries, which requires the responsible agencies to keep a closer watch for its number, parameters, and territorial allocation. One of the results of the last 10–15 years in Russia has been the definite liberalization of the radio frequency spectrum usage, which has become apparent, in particular, in the distribution of a huge number of uncontrolled devices capable of intercepting private information and the non-licensed equipment for its transmission . . .. Nevertheless, after some time passed, the appropriate legislative documents were issued and confirmed in Russia, but these efforts, unfortunately, have had limited effectiveness. As a result, not only has legalized equipment for information interception been placed in the market now – the manufacture and delivery of which can be controlled by responsible authorities – but uncontrolled equipment with very “exotic” modulation types – and very dangerous from the standpoint of economic security – is also available. A consequence of this period of time is the drastic increase in the volume of used office equipment and electronic equipment for household and industrial
1
Introduction
3
applications. These devices have the compromising electromagnetic emanations that are, in many cases, information leakage channels, for example, due to the microphone effect of the HF i UHF oscillators, the correlation between the monitor emission parameters and the computers with processed data. Moreover, it is necessary to note the following several factors related to REE complications on protected (controlled) objects: – The first relates to the large number of used REM, located in limited and often in enough low space, which can lead to the great complexity of unwanted RES (URES) revelation; – The second is the essential increase of the information transmission rate and the redundancy application to increase the secrecy and noise immunity of several REM, to which the equipment used in measuring and information radio systems of state and commercial enterprises, the wide-band systems with dynamic frequencytime structure, etc. can be, primarily, concerned; – Non-uniform (in time) REM usage leads to additional REE complications, at instances of radio system maximal intensity operation. The increasing problem of ARM effectiveness is redoubled by the fact that, with the growth in the number of international contacts and due to the liberalization of the REM market in Russia, the threats from foreign states – which collect data concerning the industrial and economic secrets of Russian enterprises and watch the scientific and technological developments in the field of perspective technologies – also grows as well. Technical facilities, in particular, the radio electronic ones, are the most important for this activity, because they are very suitable for secretive information transmission. A similar technique for veiled information theft is oriented toward obtaining and transmitting through radio channels any and all messages: from acoustic signals and speech, phone and fax signals, to emissions from computers and monitors, and other information signals, modulating the radio waveforms by many various methods. It is quite evident that the information security services of private and state organizations, and state agencies as a whole, cannot ignore the problem of possible veiled information theft and should take reciprocal measures to use radio systems as an effective counteraction to these threats. On a new level, the appearance of information security problems on controlled objects clearly shows the definite scientific, and particularly the technical, lag in ARM techniques capable of adequately resisting this threat during REE checking, of revealing and localizing potentially dangerous RES, and of detecting the electromagnetic emission and cross-talks capable of leading to important information leakage. As a matter of fact, the technical and procedural level of ARM equipment must be equal to any future achievements within the field of “information transmission,” otherwise the information struggle will be lost. All these factors stimulate both the development of ARM technologies and equipment, and the creation, by the experts, of ARM technical systems, integrated
4
1
Introduction
by the generality of the problem, by the unity approach to its structures, and by the universality and multifunctionality of its solutions. The main goal of the present book consists in the description of the structure and function of digital radio receivers and radio systems intended for radio monitoring and technical information security tasks, beginning from the characteristics and structural diagrams of radio electronic sets and systems, including ARM systems to the description and explanation of the functioning of complicated systems. To this end, we consider the methods, algorithms and peculiarities of the appropriate software. The theoretical discussions are explained via specific equipment examples of REM, created at present in Russia.
Chapter 2
Problems, Classification and Structure of ARM Equipment
Introduction The fundamental purposes of radio monitoring (RM) equipment are: • • • • •
Permanent or intermittent monitoring of airwaves in the wide frequency range Detection and analysis of new emissions Determination of the emission sources location Evaluation its danger or value Detection of unintentional or specially-organized radio channels, for information leakage.
Each of these tasks is a complex, multistage one. Each can be solved under the conditions of the complex electromagnetic environment, and each requires the application of a wide range of radio electronic means (REM), which execute definite functions [1, 2]. These functions can be divided into the following main groups: 1. Universal functions, which, as a rule, are executed by modern, automated RM (ARM) systems 2. Additional functions for specific RM task solutions in the field 3. Additional functions for RM task solutions at one, separately-controlled location, or at a group of the most important premises of the controlled object 4. Additional functions for detecting compromising electromagnetic emanations (CEE). Regarding the first RM functions group (universal functions) one can consider the following: • Real-time panoramic spectral analysis with the maximum high rate, resolving capacity, and adaptation to the complex electromagnetic environment • Fast search for new emissions, including wide-band, and emissions with the dynamic time-frequency structure, its parameters measurement, and comparison to the database, to determine its danger (value) for the user A. Rembovsky et al., Radio Monitoring, Lecture Notes in Electrical Engineering 43, C Springer Science+Business Media, LLC 2009 DOI 10.1007/978-0-387-98100-0_2,
5
6
2
Problems, Classification and Structure of ARM Equipment
• Creation of a signals database, its replenishment, and the registered data comparison with the references stored in the database • Control of the radio emission sources (RES) with emission parameters estimation • Radio signals recording, including digital signals, simultaneously with the service parameters (frequency, time, signal level, spectrum data, etc.) and its further play-back • Real-time and post-processing technical analysis of radio signals. The following functions can be classified to the second functional group: • Field strength measurement • RES direction finding with arbitrary types of modulation on azimuth and elevation angles • Stationary and mobile RES location determination in the field and on extensive objects, and its representation in cartographic diagrams (digital object image). The third group of tasks includes: • Search and detection of the technical channels of information leakage, at the separate or combined premises • RES identification as a radio microphone • RES site location. The compromising electromagnetic emanation (CEE) detection (the fourth group function) provides the following: • Technical means emission parameters and electromagnetic field strength measurement in the receiving antenna near-field zone • Confidential information immunity examination during the course of its processing and storing by the intended technical facilities • Survey of confidential information immunity against leakage, due to the pick-up from the auxiliary technical facilities, systems, and its communication lines • Allocated premises immunity analysis against speech information leakage through the acoustic and electric transformation channels • Measures effectiveness control concerning information security against the CEE leakage.
Classification of Radio Monitoring Equipment It is expedient to classify RM equipment based on specific signs, with further determination of the RM facilities’ efficient structure within each group. These signs are: • Size of the RM operation zone (territory) • RM means application
Classification of Radio Monitoring Equipment
7
• Executed functions • Performance of RM means • Design constraints. Let us consider, in detail, the RM means categories, according to these signs.
Operation Zone Sizes Based on operation zone size, all radio monitoring means can be grouped as follows [3]: • Means for RM task solutions in the field and RES direction finding • Means for information protection measures on the external boundaries of the controlled objects • Means for RM task solutions within separate or several controlled premises of the object; these facilities will be referred to as eavesdropping detection means • Facilities for CEE special investigations. The first and second group means should be able to cover substantial territories with the possibility of RES detection at the exits, and at the external boundaries of the controlled objects. The third group means should provide RM task solutions with maximum operating rate, to detect the RES location and to identify it as a radio microphone. These tasks should be solved at both the separate premise and the premises group, under control from one post. Control facilities are located inside the premises. Special investigations of the technical means for CEE presence can be executed, as a rule, in specially-allocated premises, but investigations are possible directly at the place of the means location, as well.
Application Based on the applications, RM means can be classified into three groups: • For open operation at stationary or temporary posts, as well as when moving on different transport carriers • For concealed operation with RM means carried in an attaché case, handbag, or on the operator’s body. In this instance, appropriate measures for camouflaging the antenna system should be provided, as well as measures to conceal the technical means design, and, in several cases, in combination with fully-autonomous functioning during operator movement • For combined (open or concealed) RM means application, with the possibility of carrier control, and the necessary measures for camouflaging the antenna systems and the appropriate, RM means design.
8
2
Problems, Classification and Structure of ARM Equipment
Equipment Performance RM means performance can be characterized by the signal panoramic spectral analysis operating rate at the given resolution and the dynamic range. Typically, the following classifications are used: • • • •
Low performance (10–100 MHz/s) Medium performance (100–1,000 MHz/s) High performance (1,000–10,000 MHz/s) Ultra-high performance (more than 10 GHz/s).
Design Constraints Modern RM equipment has been created based on a system approach: hardwaresoftware means united by overall design. This approach provides the ability to link each separate means with its weight and size parameters, its electromagnetic compatibility, the decoupling of its power supply, and the development of its design implementation, all of which correspond to the used-carrier parameters. Such problems can be solved by effective classification of the means into groups. Each of the groups shall fulfill each – or a number of – stated conditions. It is often suggested to divide all means into families: stationary, mobile, portable and handheld means. When developing the means of each family, those technical solutions are preferable, which first of all comply with the set of main parameters, secondly, which comply with the minimal weight and size parameters, and, lastly, which cost the least. For stationary RM means, weight and size constraints are practically absent, and therefore the best technical parameters can be achieved via RM means. Thus, to ensure a large operating area for stationary posts, an antenna system located on remote masts can be applied, which can then be mounted on high buildings or in elevated areas. For mobile RM means, which are located on a vehicle or air transport carrier that is able to execute the main function while moving, it is important to take into consideration any constraints on weight, dimensions, and power consumption. This relates to the dimensions and carrying capacity of the carriers themselves, as well as the power capacity and the power of the sources located on the transport carriers. Since the above-mentioned constraints are not very strict, in the mobile RM means family, similar to the stationary means family, one can use multi-channel digital panoramic receivers to obtain high values on the dynamic range, the rate of panoramic analysis, and on received-information processing. Portable RM means are intended for transportation by one or a number of operators and are destined for further operation at stationary or temporary posts equipped or not equipped with power sources, and in the field. There is no requirement for these means to function during transport. Thus, serious constraints are formulated as to the weight, power consumption, and dimensions of the detection and
Radio Monitoring Equipment Design Philosophy
9
direction-finding antenna system. Moreover, for portable RM means, it is necessary to have an autonomous power source to provide for its function (e.g., accumulator charge, unfolding solar batteries, fuel supply for gasoline-electric generators, etc.). Hand-held RM means are intended, first of all, for operation during operator movement when placed on the operator’s body (or in his arms). Additionally, these means can be used to solve RM tasks at temporary or stationary posts. From the point of view their application, these means are universal and their usage is appropriate to detect RES locations in out-of-the-way places or where concealed operation is needed. Due to the serious constraints of energy consumption and weight and size parameters, such means parameters should be selected taking into account the unit’s operating life with a single power source set. Measuring radio-receiving devices and antenna systems are required for the measurement of regular radio electronic means (REM) parameters at the emission control of the officially-registered communications equipment, and also to estimate the effectiveness of information-leakage prevention measures at the boundaries of the controlled objects, and for SEE investigations. Usage of RM means for measurements can occur in the stationary, mobile, portable or hand-held versions. In Russia, the possibility of such usage for measurements must be approved by the respective certificates of Gosstandard and the Federal Service for Technical and Export Supervision of the Russian Federation. Therefore, we formalize the classification of all RM means into the following groups: • Stationary RM means family • Mobile RM means family, mounted on vehicles, and on air and sea transport carriers • Portable RM means family, operation of which is provided only after its deployment at the temporal location posts • Hand-held RM means family, for concealed and open operation, intended for operation while the operator is moving (without operator participation in the mean control, or with partial or complete participation) • Measuring means, to ensure effective control of the attempts made at information-leakage prevention, and also to measure the emission parameters for regular radio facilities. In order to decrease the number of means, it is expedient to combine the first and second families (stationary and mobile) into one family, provided that the execution by mobile means equals all the functions of the stationary means, taking into account the constraints to the antenna systems and the electric power systems of mobile means.
Radio Monitoring Equipment Design Philosophy The main purpose of RM equipment development is the creation of universal hardware-software systems, using the limited range of devices to carry out the
10
2
Problems, Classification and Structure of ARM Equipment
maximum possible RM task scope [4, 5]. The main requirements for RM equipment, aimed at the minimization and unification of the equipment and software, are the following: • Universality and multifunctionality of basic RM equipment for each family • Universality and multifunctionality of the additional means • Provision for the combined operation of the family’s basic equipment with the additional equipment, common for all families of RM equipment • Unification of the different families’ equipment • Unification of software, using similar modules, data formats, and interface formats for the different families • Unification of power supply • Effective distribution of processing tasks between the hardware signal processors and the controlling computer • Creation of code libraries for the basic set of each family’s equipment • End-to-end solutions for electromagnetic compatibility problems. A partial decrease the amount of necessary RM equipment can be achieved at the development stage of each family’s equipment, based on the functionally-modular principle of combining each family’s basic equipment with the additional means common for all RM families’ equipment. Investigations of the various types of digital receiver structures with wide operating frequency ranges show that minimizing the number of means can be achieved by restricting the operating range of the family’s basic equipment combined with the additional means common for all RM families. Implementation of this principle allows the selection of a fixed, basic equipment structure for each family. Another argument in favor of this principle is the consideration that, at present, technologically, the implementation of all or most of the functions mentioned in section “Introduction” into one constructively-completed mean would lead to an unreasonable increase of weight, dimensions, power consumption, and cost. Realization of this multi-functionality principle assumes that it is possible to reduce the structure of RM equipment, based on hardware digital-unit usage, with the possibility of quick reprogramming to execute the various signal processing algorithms, to combine the functions of separately-manufactured devices, and to effectively distribute problems between two software layers, namely, those used in the hardware digital unit and in the controlling computer. An end-to-end solution to the power supply problem assumes the unification of voltages supplied, including, in the equipment structure, any units that provide power supply from the AC network, from the onboard network of the mobile vehicle (car, helicopter, etc.), as well as from the battery with the charging device for autonomous operation, and for fail prevention at any supply interruption. The following principles defining the RM equipment structure are: • Unification of different families’ equipment; possibility of combining the equipment of various families, for example, combining the radio signals analog-digital
Radio Monitoring Equipment Design Philosophy
11
converter (ADC) of the mobile unit with the double-channel or single-channel unit of the analog-digital processor of the portable family • Unification of software packages, application of a similar data structure and format to achieve the possibility of using the same software package (with various drivers) within the different families • End-to-end solution of the electromagnetic compatibility problem, accounting for the carrier’s electric equipment. Minimization of the total expenses spent on RM equipment development relates directly to the possibility of its modernization during duplication. The open command library for each equipment type allows the possibility for the user himself to program and solve individual specific tasks, using the available RM equipment hardware. RM equipment development and usage experience shows that the equipment structure should include: • Single-channel or multi-channel (with coherently-related local oscillator) radio signal converter • Single or multi-channel analog-digital processing unit • Equipment for digital radio signals recording, at the intermediate frequency (IF), to magnetic or other storage devices • Equipment for real-time signal technical analysis and post-processing • Digital demodulation unit • Equipment for recording the demodulated signal simultaneously with the service signals (current time in the record moment, current frequency, etc.) • Power supply unit with reduced interference level • Universal control equipment allowing for the possibility of fast replacement and changing of modes, based on special mathematical software (SMS) program selection • Uniform SMS packages. Measuring radio-monitoring devices should be certified by the authorized, statestandard agencies. The additional equipment includes: • Wide-range unidirectional antennas of various applications • Antenna system sets for automated direction finding when moving, at stops, and for the stationary posts • Antenna modules sets with directional properties for hand-held direction finders of open and concealed application • Radio signals tuners, to widen the operating frequency ranges • Digital signal recorders • Equipment for positioning the RM means at the geographical coordinates.
12
2
Problems, Classification and Structure of ARM Equipment
Additionally, the following argument should be taken into account in favor of separate implementation of the basic equipment and the additional means. At the stationary and mobile RM posts, one can try, if possible, to move the receiving antenna to a very high place (on the roof, on the mast, etc.) in order to widen the post’s area of operation. In that case, the RF cable from the antenna to the basic equipment may be rather long. The losses and noise generated in the cable increase with the frequency. If all units, for example, in up to the 18 GHz range are concentrated in one place (say, on the operator’s desk), then this situation – even in the case of very good cable application – would lead to unjustifiable significant signal damping in the RF cable with receiving signal frequency increase and, hence, to a reduction in system sensitivity. Moreover, in this situation, the so-called antenna effect will reveal radio signal crosstalk to the RF cable and will distort the directional pattern of the antenna. Using an additional frequency converter, for example, in the 1–18 GHz range, provided that it is located near the antenna, essentially decreases the requirements for the upper boundary frequency of the RF cable, increases the sensitivity, and reduces the antenna effect.
Requirements for RM Equipment Technical Parameters Quality Criterion Selection It is nearly impossible to execute all the parameters necessary for complete optimization of all RM tasks, due to large number of parameters. Nevertheless, it is evident that, for most executed tasks, there is a common approach. This approach consists of estimating the necessary RM equipment using an “effectiveness-cost” criterion. Under this method, the area of possible decisions is restricted as follows: • Minimal number of important parameters is defined for each task or group of tasks • Permissible (or acceptable, in the absence of clear recommendations) limit for each parameter is fixed. In a number of cases, the probability P(t ≤ Ts ) of appropriate RM task execution during the operating time interval, not exceeding the given value of signal time Ts , can be successfully used as the main index of ARM equipment effectiveness. In this instance, the important technical parameters of the equipment should not be worse than required. RM equipment can be considered optimal when it provides the greatest probability of task execution during the same time at the same cost. At the same time, at the selection of the specific equipment by the user, other indexes can be the most important, for example, the accuracy of the current frequency measurement or the accuracy of the direction finding, as well as the equipment cost.
Requirements for RM Equipment Technical Parameters
13
Let us use the probability criterion to estimate the performance of a variety of equipment, for the task of signal detection. We consider detection probability functions at the panoramic spectral analysis, under the assumption that the radio signal has the duration of, say, 3 s. This time interval is the typical average value at the radio interchange. The probability of single-frequency signal detection with Ts duration, under assumption that the signal/noise ratio (SNR) is high, is defined in Chapter 4. Let us assume that the scan range is equal to 1,800 MHz. The calculation results for the case of single-frequency REM detection are shown in Fig. 2.1. In Chapter 4, the suggested probability criterion is used for more complex cases of signal detection, for example, signals with programmable operating frequency tuning (POFT). Pdet 0.8
0.6
0.4
0.2
0
400
800
1200
1600 g, MHz/s
Fig. 2.1 Detection probability of single-frequency signal vs. a function of system performance during time interval T=3 s. Search range is 1,800 MHz at the analysis bandwidth 2 MHz
Fig. 2.1 shows that, at search range 1,800 MHz, the continuous wave (CW) signal with duration 3 s can be detected with the probability P = 0.5 at the system performance 300 MHz/s, and with unit probability beginning with the panoramic analysis speed 600 MHz/s. Plots of the “new” signal detection for the panoramic analysis rate of 1,500 MHz for several values of the search range are shown in Fig. 2.2. At this given rate, the CW signal for the maximum search range of 3,000 MHz is positively detected in only 2 s.
Main Technical Parameters of RM Equipment The basis of any RM equipment is the panoramic radio receiver, executing the functions of panoramic analysis and signal detection during its search.
14 Fig. 2.2 Detection probability of single-frequency signal vs. time at performance of 1,500 MHz/s, bandwidth 2 MHz at the various search range (1 – search range is 300 MHz; 2 – search range is 900 MHz; 3 – search range is 2000 MHz; 4 – search range is 3000 MHz)
2
Problems, Classification and Structure of ARM Equipment
Pdet
2
0.9
1
3 4
0.7
0.5
0.3
0.1 0
0,5
1,5
2,5
t, s
As mentioned above, the probability P(t ≤ Ts ) of the appropriate RM task during the definite signal time interval Ts can be used as the RM equipment performance index. At that time, the main parameter values of this equipment are fixed and recorded. At detection-problem solution, this probability depends mainly on panoramic spectral analysis speed, which is ensured by the radio receiver (RR). However, the analysis speed cannot be examined separately from the other receiver parameters: the dynamic range on the intermodulation of 2nd and 3rd order, the frequency resolution, the sensitivity, the operating frequency range, the simultaneous bandwidth, the frequency stability, and the spurious rejection. Thus, the needless increase of the resolution can essentially reduce the detection possibilities of a fast radio signal with the dynamic time-frequency distribution. At present, the panoramic digital RR’s (DRR) have the widest application in the area of RM tasks. DRR is the combination of radio signal frequency converters with the fixed IF and the analog-digital processing unit, which provide the parallel signal processing within the simultaneous bandwidth with the necessary frequency resolution [6]. This implementation provides the maximum operating rate, however, it is necessary to take into account that the simultaneous bandwidth growth at high range load leads to ADC overloading, and the application of the attenuator leads to weak signal suppression, i.e., to reducing its electromagnetic accessibility zone. The solution for this situation is the usage of frequency selection sections (so-called “comb” sections) adjoined to each other. For example, to obtain the simultaneous bandwidth equal to 80 MHz one can use 8 sections of 10 MHz; however it essentially complicates signal processing and increases the cost. The methods for creating RM multi-channel panoramic DRR are discussed in Chapter 5. The lowest frequency range boundary in RM applications is usually equal to 9 kHz for both Russian and foreign RM equipment. The upper frequency range boundary for the RR equipment base is equal to 3 GHz; it can be extended to 6, 8 or 18 GHz, with additional devices, and, at that point, the trend of upper range boundary growth is steady, as mentioned above. In any case, the implementation of the principle formulated above ensures minimum expenses on the existing equipment modernization. According to this principle, the basis DRR and the additional devices are provided in order to increase the upper boundary of the operating range.
Requirements for RM Equipment Technical Parameters
15
Nowadays, the dynamic range of 70–80 dB and resolution of 6–25 kHz are considered sufficient for RM equipment. This corresponds to 3–12 kHz spectrum discretization. The spurious rejection should be not less than 70 dB, the relative frequency stability of the reference oscillator should be not worse than 10–6 –10–7 . When necessary to obtain the better frequency stability, for example, for measuring equipment, it is possible to use the highly stable external or internal reference frequency oscillator in DRR. Table 2.1 presents the typical tactical technical characteristics of stationary and mobile RM equipment, manufactured by one Russian company.1
Table 2.1 Typical tactical and technical characteristics of stationary and mobile RM equipment
Characteristic
Stationary RM station Mobile station with antenna system with mast-mounted Vehicle-mounted Mast-mounted antenna system Panoramic spectral analysis (PSA)
Operating range, MHz Basic equipment With optional equipment PSA rate in the operating range, MHz/s For medium performance For high and ultra-high performance, MHz/s Frequency sampling, kHz For medium performance For high performance, MHz/s Dynamic range, dB Sensitivity, μV
25–3,000
25–3,000 0.009–18,000
100–1,000 More than 1,000
3 6–12 75 Not worse than 3 Direction finding
Rate in the range, MHz/s For low radio range load For high load Signal bandwidth, MHz Sensitivity, μV/m Instrumental accuracy (RMS), degrees Technical analysis Analysis bandwidth, kHz/resolution, Hz
1–12 1
50–100 More than 300 Arbitrary 2–15 1.5
2,000 (5,000)/15; 250/500, 120/240; 50/100; 9/20; 6/12 Multi-channel radio monitoring
No. of monitored channels: For low range load For high load
1 IRCOS
(www.ircos.ru)
3–15 1.5
2–4 6–8
16
2
Problems, Classification and Structure of ARM Equipment
Characteristics of RM Equipment Families Radio Monitoring and RES Location Detection Systems The required terrain coverage and RES location detection can be achieved using a system of distant RM and direction-finding (DF) stations (the central and several peripheral stations), which provide detection and signal-receiving by the central post and also the simultaneous (synchronous) direction finding by the central station command, as well as RES location calculation with representation on the map [7, 8]. The number of stations for the stationary system is defined by the relief, the possibility of using high-rise buildings for antenna mast-mounting, by the controlled RES power and the detection and DF equipment’s sensitivity For locations where it is difficult to receive signals via stationary stations, mobile RM stations shall be used as additional support for stationary RES detection and direction-finding systems. These mobile stations are intended for more accurate localization of the detected RES. Handheld direction-finding and manpack RM equipment shall be provided for RES localization inside of buildings and in places that are beyond the reach of mobile equipment. The same regularities are true for the mobile system of detection-finding and RES location detection, with the only peculiarity being that the antenna system is mounted on the remote mast with less length, in the flat country, which leads to a reduction of the REM-monitoring operating zone. The antenna system dimensions (diameter) for the mobile equipment, evidently, will be less than the appropriate dimensions of the stationary system, due to application conditions, which leads to less direction-finding accuracy at the low section of the operating range (less than 100 MHz). The portable system can be characterized by the more strict limitations on weight, dimensions, and power consumption, which inevitably adversely affect the performance and the functions. The necessity of operation under field conditions requires autonomous power-supply means in the system structure. Similar system creation tasks are discussed in detail in Chapter 9.
Stationary and Mobile RM Stations Possible organization of the stationary and mobile RM stations includes the following posts, along with the handheld direction finder or the manpack RM unit. • Post No.1. The direction finder with the stationary (Fig. 2.3a) antenna system or with the mast-mounted and vehicle-mounted antenna system (Fig. 2.3b). • Post No.2. Panoramic radio receiver. • Post No.3. Multi-channel panoramic radio receiver. • Post No.4. The cartographic and RES location-calculation equipment.
Characteristics of RM Equipment Families
a)
17
b)
Fig. 2.3 Stationary (a) and mobile (b) RM stations
a)
b)
c)
d)
Fig. 2.4 Single-channel (a), double-channel (b) and multi-channel (c, d) DRR
The single-channel DRR structure used in RM stations (Fig. 2.4a,b) is: • • • • • •
Wide-band RM antenna Tuner Analog-digital processing unit Control system with customized software package Power supply from the vehicle’s on-board power or from the AC net Additionally, the measuring antennas set and the frequency converter for the frequency range extension. The multi-channel DRR structure (Fig. 2.4c,d) is:
• • • • •
Wide-band RM antenna Multi-channel panoramic DRR with remote control Control system with customized software package Power supply from the vehicle’s on-board power or from the AC net Additionally, the frequency converter for the frequency range extension.
The parameters, the structural diagrams, and DRR examples are considered in Chapter 3, and the examples of multi-channel DRR are given in Chapter 5.
18
2
Problems, Classification and Structure of ARM Equipment
Portable RM Equipment As previously mentioned, portable equipment is mainly purposed for radio monitoring at temporary and stationary posts, as well as in open terrain and in out-ofthe-way places, where mobile and stationary equipment usage is impossible. The portable equipment functions for executing RM tasks in the field should correspond, if possible, to the functions of the stationary or mobile RM stations. It is clear that the direction-finding antenna system of the portable equipment will never be completely equivalent to the stationary or mobile antenna system. Taking into account the restrictions on power consumption and the strict weight requirements, the portable, automated, direction finders are developed, which are described in Chapter 8. RM system structure for RES location detection, consisting of portable stations, (Fig. 2.5) is: • Three or more portable stations (central and several peripheral) • Handheld direction-finding equipment of open or concealed application • Manpack RM unit. In order to achieve multi-functionality, additional tasks related to CEE, may be entrusted to portable RM equipment. Possible versions of such units are described in Chapter 11. The family of CEE detection equipment (Fig. 2.6) should include the following: control equipment for one or several premises and mobile RM, direction-finding and emission parameters measurement stations for the controlled zone boundaries. The problems of CEE detection are discussed in Chapter 11.
Fig. 2.5 System comprising the portable stations
Characteristics of RM Equipment Families
a)
b)
19
c)
Fig. 2.6 Portable ARM units, with: single-channel panoramic DRR (a); double-channel DRR (b); and multi-channel panoramic DRR (c)
Certified RM equipment with measuring antennas and additional facilities can be used together with the appropriate customized mathematical software for special CEE investigations. Chapter 12 is devoted to these applications.
Manpack ARM Equipment Implementation options for open and concealed RM applications relate mainly to manpack equipment. On the basis of our experience with communications surveillance services in Russia, we can affirm that the following manpack equipmentimplementation options are desirable [9]: in the document-case, in the handbag (rucksack) or in the multi-pocket vest (Fig. 2.7). If possible, manpack equipment should have the functions of stationary or mobile RM equipment. Taking into consideration the need to minimize equipment weight and dimensions, as well as the variety of application conditions and the absence of strict requirements for direction-finding accuracy, it is reasonable to use the amplitude
a)
b)
c)
d)
e)
Fig. 2.7 The implementation options of RM manpack equipment: RM equipment in the document-case (a); in the multi-pocket vest (b); RM handheld equipment for open applications (c); concealed direction finder (d); automatic direction finder (e)
20
2
Problems, Classification and Structure of ARM Equipment
method of direction finding, based on the directional antennas in the manpack equipment. The manpack direction finders may have both open and concealed application options. The RES search process using manpack automatic direction finders differs from the similar process using mobile facilities, by convenience mostly, since it is possible to use manpack automatic direction finders in places that are inaccessible for portable and mobile facilities. In such instances, the fundamental method for RES position determination – the “homing” method – is based on operator motion, with the manpack direction finder in the RES position area, along the bearing direction. When the distance to the RES decreases, the direction-finding signal amplitude increases, an additional sign that the direction finder is moving in the right direction. The structure of the manpack automatic direction finder is close to the portable facility structure. With the help of handheld direction finders, the RES search process is provided by means of some basic stages, which are as follows [10]: • Fast panoramic spectral analysis in the given (operating) range and the detection of “new” signals • Qualitative or quantitative estimation of the detected emission parameters • Obtained parameters comparison with the database and the determination of RES value (danger) • RES location detection is an iterative process of operator-executed RES direction-finding stages, where the estimate of each stage’s level, its comparison with the previous iteration level, and the operator’s choice of movement direction, with the equipment, for the next iteration, are fulfilled. The handheld direction finder consists of: • Exchangeable directional antennas for open application and the indicator • Exchangeable directional antennas for concealed application and the control panel • Panoramic digital radio receiver • Additional facilities providing the signal level indication, level variation, signal demodulation and audition, and (when necessary) the operating frequency range extension • Power source from the autonomous accumulators, car power net and AC net, as well as the accumulator re-charge. The problem of field strength measurement using manpack equipment can be solved by using the measuring antennas only, and including required masts or tripods for their mounting, as part of the equipment structure, if necessary. To reduce the equipment range, and to provide the unification of high-quality radio monitoring, the main technical requirements for all types of equipment, including the portable and manpack ones, should not differ greatly from the appropriate requirements for the stationary and mobile equipment. The requirements for weight, dimensions and
Conclusion
21
power consumption may be the exception to this rule. Therefore, it is expedient to compile the single-channel and multi-channel DRR from the unified modules, to put them in cases, in keeping with the main requirements for the tactical technical characteristics – and for multi-functionality – and to provide a power supply from various power sources (AC net, car on-board supply, and accumulators). Moreover, it is desirable to provide full-scale radio monitoring, with the manpack equipment in operation at the temporary or stationary posts. Such an approach to RM equipment development allows for a lightening of the workload, with regard to the interaction between the technical facilities for the various families, and provides for unified database formation, as well as for electrical and informational compatibility. Sensitivity and direction-finding accuracy are also the main parameters of the manpack equipment. Moreover, its weight and operation duration from the single power supply set are also important. For the manpack equipment, expert estimates concerning the needed accuracy of RES direction finding show that, for practical purposes, it is enough to have the angle error 100 –150 . Equipment sensitivity (across the field) defines the action zone size, which can sometimes influence the ability to safely execute the operation. With this aim in mind, it can be recognized as necessary to have the option of concealed equipment application. The sensitivity (across the field) of modern handheld direction finders, for open application, is equal to 5–25 μV/m in the frequency range of 25–3,000 MHz. In the opinion of the professional experts, this solves most of the problems. The weight of the equipment set should not exceed 5–10 kg, and the operating duration from the single power source set should be not less than 3–5 h.
Conclusion In the present chapter, ARM technical means are classified by territorial coverage zone, by application, by the character of the function, by the equipment performance, and by the construction constraints. It is expedient to divide the equipment range into the following families: • • • •
Stationary equipment Mobile equipment Portable equipment Handheld equipment for open and concealed application.
ARM equipment can be used for CEE measurements and investigations, in the presence of the authorized organization certificates and the additional facilities. This chapter proves the rationality of the approach that, in each equipment family, there is basic ARM equipment, the possibilities of which can be improved by the additional facilities that are mutual for all families. To reduce the ARM equipment set, it is necessary to use the programmable units of digital-signals processing, to share efficiently the problems that exists among the hardware and software means. It is desirable to have unified customized mathematical software packages and the
22
2
Problems, Classification and Structure of ARM Equipment
usage of the similar data structure and format to be able to use the same package (with various drivers) in all ARM equipment families. A unified set of hardware-software facilities is offered. It includes: • Single-channel or multi-channel frequency converter • Single-channel and double-channel unit of analog-digital processing • Multi-channel equipment for the radio signal digital record in the bandwidth of simultaneous analysis • Equipment for the real-time and post-processing of signal technical analysis. Digital demodulators unit • Equipment for recording the demodulated signal simultaneously with the service signals • Means for RM equipment localization, as per geographical coordinates • Power supply • Customized mathematical software for the solution of RM problems, suitable for all families. As the general estimation index for ARM equipment, the criterion “effectivenesscost” can be chosen as the most convenient criterion. At this time, equipment effectiveness is the best of all criteria at characterizing the probability of executing the appropriate RM problem solution during the fixed-time interval, under the condition of the presence of the essential additional parameters for the given equipment.
References 1. Poisel, R.A., Target Acquisition in Communication Electronic Warfare Systems. Artech House, 2004, 370 pp. ISBN:1580539130 2. Poisel, R.A., Modern Communications Jamming Principles and Techniques. Artech House, 2003, 502 pp. ISBN:158053743X 3. Rembovsky, A.M., Automated Radio Emission Monitoring – Problems and Facilities (in Russian). Special technologies. 2002. Special Edition, pp. 2–6. 4. Rembovsky, A.M., Combined Solutions of the Automated Radio Monitoring Problems by the Restricted Set of Facilities (in Russian). INFORMOST – Communication facilities No. 5 (29), Sept. 2003. p. 23–29. 5. Rembovsky, A.M., Search Facility Effectiveness Increase for Automated Radio Monitoring (in Russian). Special technologies. No. 4, 2003, pp. 40–47. 6. Ashikhmin, A.V., Sergeev, V.B., and Sergienko, A.R., Radio Receiver Front-Ends for Automated Radio Monitoring Complexes: Peculiarities, Solutions and Prospects (in Russian). Special technologies. 2002. Special Edition, pp. 57–64. 7. Rembovsky, A.M., Automated Radio Monitoring and Emission Bearing – Problems and Facilities (in Russian). Uspekhi Sovremennoi Radioelektroniki, No. 6, 2003, pp. 3–21. 8. Ashikhmin, A.V., Kozmin, V.A., and Rembovsky, A.M., Ground-based Mobile Complexes of Radio Monitoring and Direction Finding (in Russian). Special technologies. 2003. Special Edition, pp. 30–41. 9. Rembovsky, A.M., Problems and Facility Structure of Automated Radio Monitoring Facilities (in Russian). Special technologies. 2003. Special Edition, pp. 2–7. 10. Ashikhmin, A.V., and Rembovsky, A.M., Carried Direction-Finders for Emission Sources (in Russian). Special technologies. 2003. Special Edition, pp. 34–40.
Chapter 3
Radio Receiver Applications for Radio Monitoring System
Introduction Radio receivers are a system of interconnected units used for the extraction of energy from electromagnetic fields, as well as for the selection, amplification and conversion necessary to recover information from radio signals. The structural diagram of the radio receiver (RR) is shown in Fig. 3.1. Fig. 3.1 Structural diagram of the radio receiver
Radio receiver
Terminal unit
The receiving antenna executes the first main RR function: it extracts the electromagnetic field energy and converts it into the electric signal. The radio receiver fulfills the second main function: selection and conversion of the electric signal generated by the antenna, as well as its amplification. This conversion is executed in such a way as to ensure the normal operation of the terminal unit, which fulfills the third main RR function: the extraction of useful information from the received signal. Radio receivers, in turn, are sub-systems of the more complicated systems of communication, radio broadcasting, TV, radio navigation, radar, radio direction finding, radio monitoring, radio control, etc. At present, the direct conversion receiver and superheterodyne receiver (receiver with frequency conversion) are the most well known RR types. Structural diagrams of these receivers differ according to the structure of the radio frequency (RF) section.
Tuned Radio Receiver In tuned radio receivers (Fig. 3.2), the RF path contains the input circuit and the RF amplifier (RFA). In this case, all resonant circuits are tuned to the received signal frequency fs , on which the main pre-detection amplification is fulfilled. Signal and noise from the receiving antenna enter the input circuit. The input circuit is designed to match the antenna output and RFA, which ensures the main frequency A. Rembovsky et al., Radio Monitoring, Lecture Notes in Electrical Engineering 43, C Springer Science+Business Media, LLC 2009 DOI 10.1007/978-0-387-98100-0_3,
23
24
3
Input circuit
Radio Receiver Applications for Radio Monitoring System
RF amplifier
Detector (demodulator)
Base-band amplifier
Terminal unit
Fig. 3.2 Structural diagram of the tuned radio receiver
selection and pre-detection signal amplification. RFA resonant circuits are tuned within operating frequency range. Since both high selectivity and amplification are usually required (RFA gain may have the order of 106 –107 ), several amplification cascades and resonant circuits may be required. Synchronous frequency tuning of all these units is not a simple task. In microwave range, it is difficult to match the RR bandwidth with the useful signal spectrum width for filtering a noise, which does not coincide with the signal frequency. The number of resonant circuits is rarely more than three or four, due to the design complexity of the tuning. Therefore, at the operating frequency fs , the amplifier can be non-stable, and its selectivity will be insufficient because the selective circuit bandwidth B with the quality-factor Q is related to its resonant frequency f0 = fs by the formula B = f0 /Q.
(3.1)
As the tuning varies, the selectivity and gain coefficient vary: at fs growth, the bandwidth B widens and thus the selectivity decreases. The detector (or the demodulator) extracts the message from the RF signal. A low frequency amplifier intensifies the message signal to the value necessary for normal operation of the terminal unit. The advantage of the tuned radio receiver is its simplicity and relatively low level of inherent noise. The crystal receiver is the simplest direct conversion RR, having the minimum number of functional units required for signal receiving: the antenna, the selective circuit, detector, and the terminal unit. Since there are no amplifying elements in this receiver, the noise immunity and receiving quality are small. These receivers are used restrictedly in microwave, millimeter and optical ranges. The reflex receiver is a type of tuned radio receiver in which the same amplifier is used simultaneously for both pre-detection and post-detection amplification. The main principle of the reflex receiver is as follows: firstly, the active element amplifies the RF signal, which is detected, and then the audio frequency signal enters the input of the same amplifier. To reduce the amplifying cascade number and to simplify the design, in the past, regenerative and super-regenerative amplifiers were widely used in tuned radio receivers. In regenerative receivers, the negative resistance, which partially compensates for the losses, is introduced into the resonant circuit, due to the positive feedback, and it increases the equivalent Q−factor and the gain coefficient. However, these receivers have low stability because they operate in a mode close to self-excitation. Thus, it is possible to expect generated oscillations penetration
Tuned Radio Receiver
25
into the antenna, and its emission leads to the interference amplification for another receiver, which is extremely undesirable from the point of view of electromagnetic compatibility (EMC). The super-regenerative receiver is a tuned radio receiver that contains the amplifying cascade with smoothly-controlled self-excitation. In the super-regenerative receiver, the positive feedback from the RFA varies periodically with some auxiliary frequency, which greatly exceeds the signal modulation frequency. At that, during part of the cycle, the introduced resistance becomes negative and the oscillations are excited in the resonant circuit. During the next part of the cycle, the oscillation break occurs. These oscillation amplitudes exceed the received signal amplitude for 104 times, and more. The intensity is proportional to the receiving signals acting on the resonant circuit, i.e., the generated signals are, in essence, the amplified signals. The super-regenerative receivers have somewhat better stability than the regenerative receivers. Their advantage is high sensitivity in the presence of a simple electric circuit. Their shortcomings are the signal distortions and intensive spurious emissions that do not meet EMC requirements. In the superheterodyne receiver, radio signal frequency conversion occurs, which is the linear spectrum transition into the range suitable for the received-signal processing. Such receivers found the most circulation. The feature that most distinguishes the superheterodyne receiver from the direct conversion receiver is the presence of a special cascade for the frequency conversion. The structural diagram of the superheterodyne receiver is shown in Fig. 3.3. The linear receive path contains the relatively wide-band tuning pre-selector operating at the signal frequency fs , and the intermediate frequency (IF) path,
Linear receive section Signal frequency section Preselector Input circuit
RF÷àñòîò amplifier û
Tuning
Terminal unit
Base-band amplifier
Detector (demodulator)
Fig. 3.3 Structural diagram of the superheterodyne receiver
Intermediate frequency section Frequency converter Mixer
Local oscillator
Intermediate frequency amplifier
26
3
Radio Receiver Applications for Radio Monitoring System
which operates at the fixed frequency fIF with the bandwidth corresponding the signal spectrum. The pre-selector, which consists of the input circuit and RFA, provides the preliminary amplification, necessary for the signal extraction, and the receiver selectivity over the spurious channels, mainly, over the image channel. The frequency converter is the unit executing the frequency conversion (down or up) and which contains the local oscillator (so-called, heterodyne) and a mixer. In this cascade, the high frequency oscillations are converted into oscillations of another frequency, so-called, intermediate frequency (high enough), which is constant for any frequency of the received signal. As a result of the conversion, the received signal spectrum undergoes the linear transition from one frequency range to another, usually to the lower frequency range. Compared to the tuned radio receiver, the superheterodyne receiver has the following advantages: high selectivity and sensitivity, selectivity and sensitivity persistence over the frequency range, and increased stability. The high selectivity of the superheterodyne receiver is provided by the filtering on the reduced intermediate frequency. As is well known, the selectivity depends on the relative detuning f /f0 , which, at non-variable absolute detuning f , increases with the frequency fall. Therefore, the selective properties of the oscillating systems are improved. The high sensitivity of the superheterodyne receiver is also the result of processing frequency decrease, because the IFA can have a rather large stable gain. The sensitivity and selectivity persistence over the frequency range can be explained by the IF nonvariability, which allows the amplifying and selective properties to be maintained almost without variations, for any frequency of the received signal. The increased stability of the superheterodyne receiver is provided as a result of amplification distribution between the various frequency paths: radio frequency and intermediate frequency. The reduction of the number of cascades, operating at the same frequency, decreases the amplifier self-excitation danger due to the feedback. The tuner operation considerably defines superheterodyne RR performance. First of all, the tuner affects such RR parameters as the frequency range. The given range coverage depends on the local oscillator (LO) operation, which should provide stable oscillation throughout the whole receiving frequency range. LO voltage amplitude within the range should stay relatively permanent because it defines the tuning parameters and, hence, the tuner transfer factor constancy over the frequency range. The IF value strongly affects the superheterodyne RR operation and its sensitivity, selectivity and bandwidth. High selectivity over the adjacent channel is provided at low IF. Good selectivity over the image channel is ensured at high IF. It is difficult to meet simultaneously the high selectivity requirements over both channels; therefore, IF selection represents the engineering compromise. Thus, several frequency tunings are applied in professional receivers. The first IF has the high value, to provide the selectivity over the image channel. The last IF usually has the low value, to provide the selectivity over the adjacent channel. When selecting IF values, one must take into account that the IF should
Main Radio Receiver Parameters
27
not fall in the receiving signal bandwidth. This means that the first IF value must vary, depending on the received signal frequency. Moreover, it is important that the LO harmonics do not fall into the IF bandwidth. In a number of cases, it is sufficient to have double frequency conversion (Fig. 3.4). In such superheterodyne RR, there are two tuners and two IFA. To ensure high selectivity over the image channel, the first IF is selected large enough; the high selectivity over the adjacent channel, as well as the narrow bandwidth, is ensured at the low enough second IF. Tuning
Input circuit
RF amplifier
Mixer 1
Local oscillator 1
IFamplifier 2
Mixer 2
IFamplifier 1
Local oscillator 2
Fig. 3.4 Structural diagram of superheterodyne linear path with double conversion
The serious shortcoming of the superheterodyne receiver is the appearance of spurious channels: some signals can enter the receiver through the antenna circuit and may cause the output signal to appear, even at useful signal absence for the tuned frequency. The spurious channel is the frequency band outside the main receiving channel, in which radio interference creates the response appearance, caused by its passage to the demodulator or detector input. The channels including intermediate frequencies, combination frequencies, and the frequencies which are less in integer times with respect to the RR frequency tuning, IF and image frequencies, can be attributed to the spurious channels.
Main Radio Receiver Parameters The most important parameters, defining RR usage effectiveness in RM systems, are the following: • • • • • • • •
Operating frequency range Amplitude-frequency response (AFR) Transfer function irregularity (ripples) Voltage standing-wave ratio (VSWR) at RR input Selectivity over the spurious channels Selectivity over the adjacent receiving channels Noise factor and RR limit sensitivity RR sensitivity at the demodulator output
28
• • • • • •
3
Radio Receiver Applications for Radio Monitoring System
Dynamic range and intercept points for 2nd and 3rd order intermodulation Threshold of the blockage effect appearance Crosstalk value Phase noise, stability and tuning speed of the synthesizer Weight and dimensions Manufacturing and exploitation complexity, cost.
Operating Frequency Range The operating frequency range of a radio receiver is the range of possible tuning frequencies within which the main RR characteristics are ensured. At smooth tuning, the range can be defined by the limit frequencies f0 min − f0 max . The relative range is characterized by the coverage coefficient kc = f0 max /f0 min . The measuring receivers, the spectrum analyzers, the selective micro-voltmeters are remarkable for large coverage coefficients.
Amplitude-Frequency Response of the Linear Receive Path Amplitude-frequency response (AFR) of the linear receive path is the frequency dependence of the through transfer function at fixed RR tuning frequency. An example of AFR is shown in Fig. 3.5. AFR can be evaluated quantitatively with the following parameters: the selectivity kf at the given detuning f (see Fig. 3.5); the bandwidth B1 at the given irregularity SB1 ; the squareness coefficient B2 /B1 at the given attenuation levels SB2 , SB1 . Fig. 3.5 AFR irregularity estimation for linear receive path
k0 SB1
B1
SB2
B2 kΔf 0
Δf f0
f
In practice, RR path transfer function varies with the frequency. An example of this function is shown in Fig. 3.6. Usually within the limits of one sub-range, the transfer function irregularity varies smoothly; but, at the interfaces between the subranges, the jump in the through AFR occurs, which corresponds to the commutation of RR paths at tuning, e.g., from the last frequency in the previous sub-range to the first frequency of the next sub-range.
Main Radio Receiver Parameters
29
k k max k0 k min
0
f1
f2
fK-2
fK-1
fK
RR tuning frequency
Fig. 3.6 AFR of RR path: f1 is lower frequency of the frequency range; fk is higher frequency of the frequency range
The transfer function irregularity Sk is evaluated by maximal relative deviation of the transfer function kmax from its average value k0 : Sk = 20 lg [kmax /k0 ]
(3.2)
Where Kmax =max(Kmax –K0 ;K0 –Kmin ). When RR is used for the signal level measurement, AFR irregularity defines a more strict (metrological) characteristic: the limit of the permissible relative error of level measurement, which is the maximal relative deviation of the measured value from its true value.
Voltage Standing Wave Ratio If the receiver input impedance (complex resistance) differs from the cable resistance of the antenna system, then not all power transferred through the cable will arrive at the receiver. Part of this power will be reflected backward. The reflected signal will be added to the incident signal when their phases coincide, and will be subtracted when the reflected signal acts in reverse phase. As a result, in the inlet cable, the set of voltage maximums and minimums enters the intervals equal to the half wavelength. Voltage standing wave ratio (VSWR) is the ratio of the voltage maximum to its minimum: S = Umax /Umin
(3.3)
Since Umax = Uinc + Urefl , and Umax = Uinc − Urefl , where Uinc is the incident signal voltage, and Urefl is the reflected signal voltage, Equation (3.3) can be rewritten in the form: 1 + Urefl /Uinc Umax 1+r S= = = (3.4) Umin 1 − Urefl /Uinc 1−r where r = Urefl /Uinc is the voltage ratio of the reflected and incident signals. Since the power ratio of the reflected Prefl and incident Pinc signals is r = 2
2 Urefl 2 Uinc
=
Prefl , Pinc
(3.5)
30
3
Radio Receiver Applications for Radio Monitoring System
The VSWR equation can be rewritten as Prefl /Pinc . S= 1 − Prefl /Pinc 1+
(3.6)
Figure 3.7 shows VSWR vs. the power ratio of the reflected and incident waves. If a receiver’s input resistance is purely active and equal to the wave impedance of the input cable, S = 1 and reflected power is absent. If the input impedance is not equal to the input cable wave impedance, VSWR is more than 1. If the reflected power is equal to 10% of the incident power, S ≈ 2; if the reflected power is equal to 25% of the incident power, S = 3. In practice, the value S < 3 is usually considered as acceptable for RR input cascades. Fig. 3.7 VSWR vs. the power ratio of the reflected and incident waves
S 5
3
1 0 0.05
0.15
0.25
0.35
0.45 Prefl Pinc
Main Channel and Spurious Channels The frequency bandwidth in which the received signal spectrum falls forms the main receiving channel. The frequency bands, which join to the main channel and which may be occupied by the outside (spurious) signal spectra, form the adjacent receiving channels. Spurious channel formation can be explained by the frequency conversion in the superheterodyne receivers. The frequency conversion process is the high frequency f0 voltage transformation (signal at the tuning frequency) into the voltage at another (intermediate) frequency fIF , without changing the modulation type and character. The frequency conversion is executed in the frequency converter and can be fulfilled both with down-conversion (fIF < f0 ), or with up-conversion (fIF > f0 ). Usually, the tuner includes the local oscillator, which is the low-power oscillator generating the oscillations with fLO frequency, and the mixer, in which the oscillations of signal and local oscillators are mixed and one of the combination frequencies is extracted, for example, by decreasing the frequency (Fig. 3.8): fIF = fLO − f0 .
(3.7)
Main Radio Receiver Parameters
31
Pre-selector AFR
0
IF section AFR
f0
fIF
fLO
f
f
0
fIF
Fig. 3.8 Received signal frequency shift
This combination frequency is indeed the intermediate frequency. Thus, in the tuner, the variation of the signal carrying frequency occurs without the distortion of the information carried with the signal. The frequency mixer creates the combination frequency spectrum, in response to the arrival of two or more signals of different frequencies. In the mixer, the signal spectrum transition occurs into intermediate frequency range without the amplitude and phase ratios of the spectrum components disturbance. However, in a similar way, the radio emission can be received for the frequencies falling higher LO frequency fIF = f0 − fLO .
(3.8)
Thus, the spurious channel with the receiving at image frequency fim fim = fs + 2fIF .
(3.9)
corresponds to the useful channel with signal receiving at fs frequency. All inherent parameters of the tuner for the image channel and the channel on the signal frequency are absolutely the same. Therefore, the image channel is the one of the most dangerous spurious channels (Fig. 3.9). When using the total (sum) frequency fIF = fs + fLO , the image frequency is
fLO fIF
0
f 0
fIF
Fig. 3.9 Image channel formation
fim
f
32
3
0
fsp
0
fIF
Radio Receiver Applications for Radio Monitoring System
f
f
Fig. 3.10 Direct leakage channel formation
fim = 2fIF − fs .
(3.10)
Another spurious channel, which is referred to as the direct leakage channel, is the channel whose frequency is equal to IF (Fig. 3.10). If the signal with IF frequency acts at the tuner input, signal direct leakage without frequency conversion occurs, but with the amplification in the tuner and in the cascades of IF path. It should be noted that the frequency of the direct leakage channel is constant and equal to IF, while the image channel frequency varies at each RR retuning. The direct leakage channel non-related to the frequency conversion is as dangerous as the image channel. However, when evaluating the danger of these spurious channels, we must take into consideration that the direct leakage channel with the chosen fIF is fixed, but the image channel moves after the useful signal channel. Therefore, the probability of interference passing through the image channel is greater than through the direct leakage channel. Measures against the spurious channels are possible in the circuits before the tuner only, namely, in the input circuits, in the pre-selector, or in RFA. To eliminate the interference from RES, the frequency of which is equal or close to IF, the special rejecters are often used at RR input (in signal frequency path). The channels whose frequencies differ from LO harmonics 2fLO , 3fLO ,. . ., kfLO by fIF value fsp = kfLO ± fIF ,
(3.11)
where k is any integer, are the spurious channels as well. The spurious channel at combination frequencies fcom is formed as a result of the interaction of the tuner spectrum components with LO frequency or LO harmonics (Fig. 3.11): mfcom = nfLO ± fIF ,
(3.12)
where m, n are any positive and negative integers. The frequency of the spurious combination channel is
Main Radio Receiver Parameters
33
fcom 2fLO fIF
0
f
f 0
fIF
Fig. 3.11 Combination channel formation
fcom = (1/m)fIF ± (n/m)fLO .
(3.13)
The interference squeal (at demodulator output) or the received signal “twins” appearance (on the spectral diagram) is the type of distortion related to non-linear processes occurring in the tuner. In addition to the intermediate frequency in the tuner output circuit, the combination frequencies close to IF can appear. Thus, if the receiving signal is at a frequency which is k times lower than the frequency of any spurious channel fs = fsp /k
(3.14)
interference from signal k th harmonic forms, due to distortions in the mixer. For the main receiving channel, the input signal frequency shift, by some testing step f , corresponds to the same frequency shift of the output signal to the value f . At this, the shifted sign depends on the specific LO frequency position, with respect to the received signal frequency position for the given tuning frequency. For the image receiving channel, the input signal frequency shift to some testing step f corresponds to the output signal inverse shift in frequency to the value minus f (see Figs. 3.8 and 3.9). This property can be used to distinguish the signal for the main and image receiving channels. For the spurious combination channels – the spurious channels at the frequencies, which are integer times less than RR tuning frequencies – for IF, image frequencies, the input signal frequency shift by some testing step f corresponds to the output signal frequency displacement by the value, multiple to f , proportionally to the values and signs of the coefficients m and n (respectively, to combination order). Transfer functions within the bandwidth of the main channel, the spurious channels at image and intermediate frequencies do not depend on the input signal level (provided that the testing signals are rather small and the overload mode does not occur). At that, the input signal level variation by some testing value U corresponds to RR adequate output signal level variation by the same value U. Transfer functions within the bandwidth of the spurious combination channels, of the spurious channels at the frequencies (integer times less than RR tuning frequency), of IF, of image frequencies, depend on the input signal level. Therefore,
34
3
Radio Receiver Applications for Radio Monitoring System
the input signal level variation, by some testing value U, corresponds to the inadequate RR output signal level variation kU, where k is the multiple depending on the input signal amplitude and its harmonics level, type of tuner non-linearity, value and sign of the testing impact U. Thus, there are many signals at the various frequencies, which are converted into the oscillations at the same IF, and, in the general case, the number of these signals is infinite. From this set, only one signal is useful, and others correspond to the spurious receiving channels; therefore, suppression of the spurious channels and direct leakage channel should be strictly regulated in RR technical requirements.
RR Selectivity Radio receiver selectivity is its ability to extract the useful signal, at which the receiver is tuned, from the spurious signals arriving from the antenna system. In most cases, the interference level in the receiving antenna exceeds the useful signal level, which emphasizes the particular importance of this RR characteristic. The receiver’s ability to extract the useful signal from the spurious ones is based on the usage of distinguishing features between useful and spurious signals, namely, the emission arrival direction and time of activity, the amplitude, frequency and phase. The first feature is used at the spatial selectivity, which is realized with the help of antennas with the sharp pattern. The second feature allows the fulfillment of time selectivity, which is the receiver opening only for the period of the useful signal action. The amplitude, frequency and phase distinction of the useful and spurious signals is based on the amplitude, frequency and phase selectivity, respectively. The frequency selectivity is of primary importance. It can be explained by the fact that, in radio communication systems, the signals differ in frequency and signal separation can be fulfilled with the help of the resonant circuits and filters. One can distinguish two types of selectivity: one-signal and real. The one-signal selectivity is defined by filters AFR of the RR radio frequency path without the non-linear phenomena account at the single input signal (or useful, or spurious). The one-signal selectivity is quantitatively estimated by the ratio of the testing signal level at the interference frequency to its value at the useful signal frequency, for invariable tuning and the same output voltage. This selectivity can be estimated as well by the ratio, showing how much larger the gain of the radio path or the gain of the receiver’s separate cascade is compared with the gain for the spurious signal. Measurement of the one-signal selectivity is used for performance determination at the small enough levels of the input radio signal, which allows for the avoidance of the influence of non-linear processes (e.g., caused by overload) on the measurement results. The circuits of automatic control (of frequency, gain, etc.) are disconnected during these measurements. If there are no defined frequencies, at which the selectivity should be measured, in the RR technical requirements, then it should be measured at the end frequencies
Main Radio Receiver Parameters
Signal generator
35
Attenuator
Selective microvoltmeter
RR
Fig. 3.12 Structural diagram for determination of one-signal selectivity
and in the middle of each sub-range. To determine the selectivity curve by the onesignal approach, one can use the structural diagram shown in Fig. 3.12. The general approach for determining the selectivity curve consists in the following. The receiver and signal generator are tuned to the required frequency. The output signal level and the additional attenuator attenuation value are set in such a manner that the non-linear phenomena are absent during the testing signal receiving at the tuning frequency. The receiver output signal level Un (as the normal level of the receiver output signal) and the receiver input signal level U1 are fixed. After that, the generator frequency is increased by some value f . Then, the level of the generator signal is increased till the value, at which the receiver output signal level will again be normal (Un ). We measure the receiver output signal level U2 at the second point of the measuring curve. The measurements are repeated, increasing the generator detuning f to the necessary value. After that, the researcher changes the generator detuning, by the same steps, into the range of frequencies less than the receiver tuning frequency. On the basis of data obtained, one can plot the selectivity curve (Fig. 3.13). This curve is used for the determination of the bandwidth, the selectivity curve squareness coefficient, and the attenuation in the adjacent channel. The selectivity for the image channel as well as for the direct leakage and the combination channels are determined for the large detuning. The signal attenuation in the spurious channel is the ratio of the input radio signal level, required for the given output signal level, to the useful radio signal level, necessary to obtain the same output signal. Thus, the frequency selectivity characteristic in the spurious channels defines the susceptibility level function for the spurious receiving channels, with respect to the testing signal frequency. The bandwidth B is the band limited by two frequencies, at which the signal level attenuation does not exceed the given limits.
Uosc
Noise level
fosc Fig. 3.13 Radio receiver selectivity curve
fsp
f0
fim
fcom
36
3
Radio Receiver Applications for Radio Monitoring System
The slope of receiver AFR decays depending on the selection filter complexity inside the linear receiving path, and shows the reduction speed of the transfer function outside the bandwidth. The decay slope can be measured in decibels/Hz, in decibels/octave, or in decibels/decade (the octave means the twice frequency variation, the decade means ten-time variation). Sufficient information about the selectivity at one-signal approach measurement can be obtained on the basis of an analysis of the frequency differences corresponding to the signal attenuation by 20, 40, 60, 80, and 100 dB, beginning from the end frequencies of the bandwidth. If the attenuation values obtained in such a manner are close to the lower and higher end of the bandwidth, one can indicate average values only.
Inherent Noise and Receiver Sensitivity The inherent resistance of antenna, its thermal noise and RR input circuits noise are the factors that affect RR sensitivity, i.e. RR ability to ensure that weak radio signals are received. The main influence is exerted by the noise appearing in RR input cascades because this noise is amplified in the same manner as the useful signal. It is well known that each conductor having the inherent resistance creates electrical fluctuations, i.e., a noise in the whole frequency spectrum. This noise is conditioned on the thermal motion of the electrical charge carriers. The random thermal motion of the charge carriers in the conductor causes the random electric potential between its ends. This electric potential oscillates around the average value equal to zero, and its average square is proportional to the absolute temperature. This noise is referred to as the thermal noise. The noise magnitude depends on the conductor ohm resistance, its temperature, and the bandwidth of the transmitted signal. Root-meansquare (RMS) voltage of the thermal noise Un (expressed in Volts) is defined as: Un =
√
4kTBR
(3.15)
where k = 1,38 ·10−23 J/K is Boltzmann constant; T is the temperature (in K); B is the bandwidth (in Hz); R is the resistance (in Ohm). The receiver is a system consisting of the active and passive elements set, and possessing an active resistance. The linear part of the radio receiver, from the input to the detector, can be characterized by the non-dimensional noise coefficient F, which shows how many times larger the input signal and noise power ratio Ps /Pn is than the output signal and noise power ratio Ps,out /Pn,out [1]: F=
Ps /Pn . Ps out /Pn out
(3.16)
At present, it is acceptable to use the noise coefficient, expressed in decibels. The noise coefficient in the ideal noiseless receiver is F = 1(0 dB), because the
Main Radio Receiver Parameters
37
signal and the noise are amplified in the same manner (with the similar gain). In real receivers, the noise coefficient is increased due to the inherent noise and, as a result, the output noise power grows and the output signal-noise-ratio (SNR) decreases. The output power Pn,out can be presented as a sum of two items: Pn G, caused by the amplification of the input (source) noise, and Pinh , caused by the inherent own noise, where G is the receiver power gain coefficient. In this case, Equation (3.16) can be converted in the form: F=
Pinh Ps (Pn G + Pinh ) =1+ . GPs Pn GPn
(3.17)
In order to be able to compare various receivers by their noise properties, the standard value of the resistor R thermal noise power at T = 293 K is used as the input noise power Pn = 4kTBR.
(3.18)
Sometimes, another temperature value (299 or 300K) is used and, at that, the numerical value Pn changes insignificantly. RR sensitivity evaluated by the signal power value Ps only, at which signal receiving is ensured, takes into consideration the RR amplification properties only. It can appear that, by means of the amplification increase, one can ensure the receiving of any arbitrarily weak signals. As a matter of fact, however, the receiver with the greatest amplification inevitably amplifies its own inherent noise, and that restricts its sensitivity. The ratio of signal power to noise power at the RR linear path output characterizes the SNR, which is often referred to as the discrimination coefficient q = Ps,out /Pn,out .
(3.19)
RR ultimate sensitivity is equal to the minimal input signal power Ps = Ps min for the discrimination coefficient q = 1. At that Ps min = FPn
(3.20)
Thus, RR ultimate sensitivity is proportional to the noise coefficient. Consistent reception of the useful signal is ensured at the considerable useful signal power Ps,out excess over the noise Pn,out , i.e., at the discrimination coefficient q > 1. The real RR sensitivity is estimated by the minimal input signal power Ps , at which the required value q > 1 of the discrimination coefficient is achieved: Ps = FPn q.
(3.21)
Let us obtain the calculation formula for determination of the real RR sensitivity. Assuming that the signal source has the inherent resistance Rss , as shown in Fig. 3.14, the noise power at RR input can be written as: Pn =
In2 Rin
=
Un Rss + Rin
2 Rin .
(3.22)
38
3
Radio Receiver Applications for Radio Monitoring System
Fig. 3.14 Equivalent circuit for RR sensitivity determination
Rss
Un
In
Rin
In order to ensure the maximal power in the load, we should satisfy the condition of the inherent source resistance Rss and the load resistance Rin equality: Rin = Rss , then Pn = Un2 /(4Rss ).
(3.23)
Assuming that the noise has a thermal origin and is defined by Equation (3.15), we can determine the real sensitivity of the receiver as: Ps = qFPn = qF
4kTRss B = qFkTB. 4Rss
(3.24)
Power sensitivity can be transformed into voltage sensitivity. For the matched load Ps = Us2 /4Rss and (3.25) Us = 2 qFkTBRss . It should be remembered that, at Rss = Rin , the voltage at RR input is twice as less than the voltage acting at the source output in the no-load condition. To calculate the radio equipment parameters, it is convenient to use the logarithmic unity. The noise coefficient expressed in decibels (noise-factor), is NF = 10 lg F.
(3.26)
The sensitivity, expressed in decibels with respect to mW (dBm), can be presented as: qFkTB = 10 lg q + 10 lg (1.38 · 10−20 T) + 10 lg B + NF. (3.27) Ps = 10 lg 10−3 Let us check how much RR sensitivity changes at varying ambient temperature. It is evident that, in the last equation, only the second item a = 10 lg (1.38 · 10−20 T) ◦ depends on the temperature. At T = 223 K (i.e., –50 C) we get a = −175.1 dBm; ◦ ◦ at T = 353 K (+60 C) a = −173.4 dBm. Thus, for temperature variation by 110 C, ◦ the sensitivity changes less than 2 dB. For room temperature T = 293 K (20 C), Equation (3.24) can be rewritten in the simplified form: Ps = Q − 174 + 10 lg B + NF, dBm
(3.28)
where Q = 10 lg q is the required SNR at RR output (discrimination coefficient) in dB.
Main Radio Receiver Parameters
39
For the ideal receiver without the inherent noise F = 1, and in 1 Hz-bandwidth, the threshold sensitivity, i.e., the sensitivity at output SNR Qout = 0 dB, is equal to –174 dBm. Using Equation (3.25), at the temperature T = 293 K and the input resistance Rss = 50 Ohm, we can calculate the voltage sensitivity, expressed in decibels with respect to μV (dB μV), as: Us = 20 lg (2·106 qFkTBRss ) = Q − 61 + 10 lg (B + NF).
(3.29)
For example, RR sensitivity for the bandwidth B = 10 kHz and the noise-factor NF = 12dB at the output SNR Q = 10 dB will be Us = 10 − 61 + 10 lg 1,000 + 12 = 1 dB μV
(3.30)
or in micro-volts Us = 1.08 μV.
Sensitivity Increase with the Help of Pre-amplifiers Any amplifier assimilates the noise signal as the input signal. At the cascade connection of the electronic units (cascades), each cascade amplifies both the signals and a noise, passed through the previous cascades, adding, at that point, its own inherent noise. Let us determine the noise factor of three cascades connected consecutively as shown in Fig. 3.15. In accordance with Equations (3.16) and (3.17), the total noise factor is Ps
Fig. 3.15 Consecutive cascade connection in RR
Pn
F=
F1, G1
Ps1 Pn1
F2, G2
Ps2 Pn2
Ps /Pn Ps (Pn2 G3 + Pinh3 ) = Ps3 /Pn3 Pn G1 G2 G3 Ps
F3, G 3
Ps3 Pn3
(3.31)
where G1 ,G2 ,G3 are power gain factors of the first, second and third cascades; Pn2 is the output noise of the second cascade; Pinh3 is the inherent (own) noise of the third cascade. Representing the second cascade output noise in the form of a sum of the inherent noise and the amplified input noise, and then, in the same manner, the output noise of the first cascade, we find F=
[(Pn G1 + Pinh1 )G2 + Pinh2 ]G3 + Pinh3 Pn G1 G2 G3
Pn G1 G2 G3 + Pinh1 G2 G3 + Pinh2 G3 + Pinh3 = . Pn G1 G2 G3
(3.32)
40
3
Radio Receiver Applications for Radio Monitoring System
From (3.17) we find Pinh = (F − 1)GPn .
(3.33)
Substituting Pinh into (3.32) we get F=
Pn [G1 G2 G3 + (F1 − 1)G1 G2 G3 + (F2 − 1)G2 G3 + (F3 − 1)G3 ] . Pn G1 G2 G3
(3.34)
Executing the cancellations, we fulfill the final formula for the noise factor for three consecutively connected cascades: F = F1 +
F2 − 1 F3 − 1 + . G1 G1 G2
(3.35)
On the analogy of (3.35), we can form the equation for the noise factor for an arbitrary number of cascades: F = F1 +
F2 − 1 F3 − 1 FM − 1 + + ... + M−1 G1 G1 G2 Gm
(3.36)
m=1
where M is the cascade number. This formula is called Früs’ noise equation [1]. From (3.36), we can see that the whole system noise is defined first of all by the first cascade parameters. The contribution of the other cascades can be practically neglected, if the first cascade gain is large. It should be noted that, in the general case, the noise factor and transfer function of each cascade will depend on frequency, i.e., they will have different values in various frequency ranges. This means that the specific calculations can be executed in definite frequency intervals only. As a rule, a RR is connected to an antenna system by the connecting cable. As any electric device with losses, the coaxial cable has its own noise level [1]. At room temperature, the noise factor of a coaxial transmission line is equal to the losses in it. When frequency grows, the losses in the coaxial cable increase. Figure 3.16 shows the plots of linear attenuation (over 1 m length) vs. frequency, for several types of coaxial cables manufactured in Russia. As we can see from the figure, the signal attenuation value in the cable and, hence, its noise factor increases with transferred signal frequency growth. Attenuation values for the flexible coaxial cables at 1,000 MHz frequency are within the limits 0.1–0.6 dB/m, but, at 2,000 MHz, frequency values are within the limits 0.2–1 dB/m. At sufficient cable length, the cable’s noise factor will be rather significant and that will decrease RR sensitivity. For example, if cable losses at a 2,000 MHz frequency are 0.5 dB/m, a cable 30 m in length will have the noise factor NF = 15dB. The first possible option to decrease the noise factor of the cable line is the usage of the cable with minimal losses. Unfortunately, such cable with minimal losses has a very high cost.
Main Radio Receiver Parameters Fig. 3.16 Coaxial cable attenuation vs. frequency
41
G, dB RK-50-2-22
0.8 0.6 0.4
RK-50-4.8-31(32)
RK-50-7-11
0.2 RK-50-7-34
0
RK-50-7-35
RK-50-9-11
600
1000
1400
1800 f, MHz
The second way is by cable length minimization or, in the ideal case, in RR mounting just near the receiving antenna. If the receiver has small dimensions, this problem can be essentially simplified, e.g., the direction finder’s receiver can be mounted directly on the antenna array basis (see Chapter 8). Finally, the third option to decrease the influence of the cable noise factor is the application of a low-noise amplifier (LNA) based just near the receiving antenna (Fig. 3.17). This LNA should have a noise factor not exceeding several decibels and also the required gain. Low-noise pre-amplifier
Cable line
Radio receiver
Fig. 3.17 Example of low-noise pre-amplifier application for decreasing the influence of the cable line noise factor
Let us examine the example. We assume that LNA is used with the noise factor NF1 = 4dB and the gain g1 = 30dB. The connecting cable has the noise factor NF2 = 10dB and attenuates the signal by g2 = −10dB. The receiver has the noise factor NF3 = 12dB. Let us transform these values to the absolute values. For LNA F1 = 10NF1 /10 = 104/10 = 2.51; G1 = 10g1 /10 = 1030/10 = 1000. Similarly, for the cable and the receiver, we get: F2 = 10; G2 = 0.1; F3 = 15.85. Let us obtain the total noise factor F = F1 +
F2 − 1 F3 − 1 10 − 1 15.85 + = 2.512 + + = 2.67 G1 G1 G2 1000 1000 · 0.1
(3.37)
or in decibels NF = 4.3 dB. If we have no pre-amplifier, the total noise factor would be: F3 − 1 15.85 − 1 = 10 + = 158.69 (3.38) F ∗ = F2 + G2 0.1
42
3
Radio Receiver Applications for Radio Monitoring System
or in decibels NF ∗ = 22 dB. Thus, a pre-amplifier with the inherent noise factor NF1 = 4dB and with gain g1 = 30 dB increases the system sensitivity by = NF ∗ − NF= 22 – 4.3 = 17.7 dB. The question is: how to choose the LNA gain factor correctly, for the given noise factor? When increasing the LNA gain factor g1 , the total noise factor will asymptotically tend to its own noise factor value. Figure 3.18 shows the system noise factor plot versus the LNA gain factor for three types of the connecting cable with the noise factor 5, 10, and 15 dB. The other system parameters were kept unaltered. From these curves, we see that, when using the cable with the noise factor NF2 = 5 dB, the required LNA gain should be near 20 dB; for the cable with the noise factor 10 dB the required gain is 25 dB; and, finally, for the cable with the noise factor 15 dB, the required gain is 30 dB. Thus, it is evident that further LNA gain growth does not improve practically the system noise factor. NF, dB
Fig. 3.18 System noise factor vs. LNA gain factor
25
15 dB
20 10 dB 15 5 dB
10 5 0
5
15
25
35
g1, dB
If the wide-band signals have a large level and borrow the wide frequency band, the pre-amplifier can be overloaded. Hence, the main attention should be placed on its linearity, especially, if there are no pre-selection filters in its input. Moreover, in the measuring systems, LNA with the calibrated gain should be used to minimize the measurement errors.
Pre-amplifier Gain Factor Selection The dynamic range D of the receiver or its separate cascades is understood to be the ratio of maximally possible and minimally possible input signal levels. Usually the dynamic range is expressed in decibels as D = 20 lg
Uin max Uin min
= 10 lg
Pin max Pin min
= 10 lg (Pin max ) − 10 lg (Pin min ). (3.39)
The minimal level values are usually equal to the threshold RR sensitivity. The maximal values are defined by the acceptable non-linear distortion level at the output.
Main Radio Receiver Parameters
43
Let us return to the typical circuit of an antenna system connected to a RR by a connecting cable line. As shown in section “Sensitivity increase with the help of preamplifiers”, in order to reduce the harmful influence of the cable’s inherent noise, it is necessary to apply a pre-amplifier with a low noise factor, just after the antenna system. The gain factor growth decreases asymptotically the total noise factor of the system. For the hypothetical case, when the gain factor is equal to infinity, the noise factor of the whole system is equal to the noise factor of the amplifier. In the above-mentioned example, the LNA with its own noise figure of NF1 = 4 dB and a gain factor of g1 = 30 dB increased the system sensitivity by = 17.7 dB, i.e., it actually extended the system’s dynamic range into the small values range, for this value. On the contrary, with gain factor growth, the system’s dynamic range decreases into the large values range, by the difference between the amplifier gain factor and the value by which the dynamic range was extended in the small values range. For example, in the above-mentioned example, the dynamic range decreases by g1 − = 30 − 17.7 − 12.3 dB. As we can see from Fig. 3.18, beginning from the definite value, the gain factor growth does not practically decrease the noise factor. Hence, in order to avoid excessive dynamic range reduction, the LNA gain factor should not exceed some necessary value, enough for fulfillment of the required noise factor and system sensitivity. One can see from Fig. 3.18, that to establish the total system noise factor NF ≤ 5 dB, and the LNA gain factor g1 ≈ 20 dB, then, for the cable with NF2 = 5 dB, we have g1 ≈ 25 dB, and, for the cable with NF2 = 15 dB, we have g1 ≈ 30 dB.
Receiver Multi-Signal Selectivity Multi-signal selectivity describes the receiver’s ability to extract the weak useful signal in the presence of powerful disturbing signals falling outside the receiving bandwidth. The interference from these signals appears in the mixer. If the mixer fulfilled absolutely exactly the voltage multiplication operation for the signals from the receiver input and the local oscillator, no interference due to out-of-band signals would appear at all. Each input signal would create its own different frequency at mixer output, and the receiver’s multi-signal selectivity would coincide with the one-signal selectivity. Real mixers do not have this feature. Firstly, they mix up the different input signals so that the first one serves as the local oscillator signal for another, thus causing intermodulation noise. Secondly, they detect signals, which lead to crosstalk noise, namely, to modulation transition from the disturbing signal to the useful one. Thirdly, they detect the powerful input signal, which leads to the blockage, i.e., transfer function variation of the linear cascades.
Intermodulation Noise Intermodulation in the receiver is the interference occurrence at RR output when two or more disturbing signals act at its input, the frequencies of which fall outside the main and spurious receiving channels. These interferences are referred to as
44
3
Radio Receiver Applications for Radio Monitoring System
intermodulation. The reason intermodulation occurs is due to the amplitude nonlinearity of the transfer function of RF path elements. Gain-transfer characteristic (GTC) of the receiver or its separate cascades is referred to as the function of the output voltage amplitude (or the active value) versus the input sinusoidal voltage of fixed frequency. Figure 3.19 shows the GTC of the ideal path, by the dotted line, and the real path, by the continuous line.
Fig. 3.19 Gain-transfer characteristic of the path
Uout
Uout comp Uout max
III
II Uout min Un
I Uin 0
Uin min
Uin max Uin comp
The real path GTC can be divided into the following parts: Part I– the part of the signal and noise (interference) superposition (between Un and Uin min points); Part II– the linear part (between Uin min and Uin max points); Part III–the overload part (between Uin max and Uout comp ). GTC of the ideal and real paths coincide at the linear part II from Uin min to Uin max . In this part, GTC is the straight line, the slope angle of which defines the voltage gain factor of the path. In part II, at Uin < Uin min , the real path GTC does not pass through the coordinates origin. Even at Uin = 0 some voltage Un acts at the path output, which is caused by the action of the fluctuations and interference in the path. In part III, at Uin > Uin max , the real path GTC becomes detached from the ideal path GTC, which is related to the overload of the real path at large levels of the input signal. The condition Uin min < Uin < Uin max should be satisfied for normal path operation. Let us analyze the influence of the analog path transfer function non-linearity on the useful signal amplitude variation. Approximation of the path transfer function is quite complex, but the main nonlinear transformation regularities can be understood if we use a simple model, in the form of a non-linear two-port network, where volt-ampere (amplitude) characteristic, i.e., the function of output signal voltage versus that of input signal voltage, has a polynomial form: iout =
∞ k=0
bk ukin ≈ (b0 + b1 uin + b2 u2in + b3 u3in + b4 u4in + ...).
(3.40)
Main Radio Receiver Parameters
45
We shall be limited by the cubic polynomial for the analysis, the combination components occurring as a result of the non-linear transformation: iout =
3
bk ukin = b0 + b1 uin + b2 u2in + b3 u3in .
(3.41)
k=0
The two signal sum can be accepted as the input signal uin instantaneous value uin = u1 + u2 = U1 cos ω1 t + U2 cos ω2 t.
(3.42)
Substituting Equation (3.42) into (3.41) we get, after exponentiation: iout (t) = b0 + b1 U1 cos ω1 t + b1 U2 cos ω2 t + b2 U12 cos2 ω1 t+ +2b2 U1 U2 cos ω1 t cos ω2 t + b2 U22 cos2 ω2 t + b3 U13 cos3 ω1 t+ (3.43) +3b3 U12 U2 cos2 ω1 t cos ω2 t + 3b3 U22 U1 cos2 ω2 t cos ω1 t+ +b3 U23 cos3 ω2 t Using the known trigonometric relations cos2 α =
1 (1 + cos 2α); 2
1 [cos (α − β) + cos (α + β)]; 2 cos (2α−β) cos2 α cos β = 12 cos β + cos (2α+β) + ; 2 2 cos α cos β =
cos3 α =
3 4
cos α +
1 4
(3.44) cos 3α,
formula (3.43) can be presented in the form: iout (t) = b0 + b1 U1 cos ω1 t + b1 U2 cos ω2 t +
b2 U12 2
+
b2 U12 cos 2ω1 t + 2
+b2 U1 U2 cos (ω1 t + ω2 t) + b2 U1 U2 cos (ω1 t − ω2 t) + +
b2 U22 cos 2ω2 t 2
+
3b3 U13 cos ω1 t 4
+
3b3 U12 U2 cos ω2 t 2
+
3b3 U12 U2 cos (2ω1 t+ω2 t) 4
+
+
b3 U13 cos 3ω1 t + 4
3b3 U12 U2 cos (2ω1 t−ω2 t) + 4
+
3b3 U22 U1 cos (2ω2 t−ω1 t) + 4
b2 U22 2 +
46
3
+
Radio Receiver Applications for Radio Monitoring System
3b3 U22 U1 cos (2ω2 t+ω1 t) 4
3b U 3 cos ω t + 3 24 2
+
+
3b3 U22 U1 cos (2ω2 t−ω1 t) + 4
(3.45)
b3 U23 cos 3ω2 4 4
It should be noted that the cubic polynomial use for receiver path transfer function approximation allows the illustration of harmonics and new frequency components occurrence, but it does not ensure the correct calculation of these frequency components, corresponding to practical GTC. Nevertheless, the considered example shows that in the spectrum of the current, passing through the non-linear element – the characteristic of which is given by the third order polynomial – besides the components with frequencies ω1 and ω2 , the additional spectrum components occur, the frequencies of which are presented in Table 3.1. Table 3.1 Combination components Combination frequency order N
Frequencies
1 2 3
ω1 , ω2 2ω1 , 2ω2 , ω1 + ω2 , ω1 − ω2 3ω1 , 3ω2 ,2ω1 + ω2 , 2ω1 − ω2 , 2ω2 + ω1 , 2ω2 − ω1
Spectral component frequencies at non-linear element output are referred to as combination frequencies. The combination frequencies are described by the equation: ω = [n1 ω1 + n2 ω2 + ... + nm ωm + ...]
(3.46)
where ni are any positive or negative integers, including zero. The combination frequencies are usually grouped combining together all frequencies for which N = |n1 | + |n2 | + ... + |nm | .
(3.47)
The number N is referred to as the combination frequency order. There exists the following regularity [2]: the item with exponent N in the path non-linear transfer function causes the combination component occurrence with the maximal order equal to N. If N is an even number, the combination components of the even order occur: N, N − 2, N − 4 till DC component N = 0. If N is an odd number, the combination components of the odd order occur: N, N − 2, N − 4 till N = 1. Intermodulation characteristics are extremely important properties for quality determination because, in the most cases, the receiver must operate in a complex electromagnetic environment in the presence of powerful disturbing signals at different frequencies.
Main Radio Receiver Parameters
47
Output signal
Frequency 0 f2 – f 1
2f1 2f2 f1 f2 2f1 – f2 2f2 – f1 f2 + f1 Input signals Second order products Third order products
3f1 3f2 2f1 + f2 2f2 + f1
Fig. 3.20 Second and third order intermodulation products
Figure 3.20 shows the possible location of intermodulation components of the second and third order, occurring when two sinusoidal signals of a similar level act on the path input. We can see that the even order products are formed farther from the input signals on the frequency axis compared with the odd order products at the frequencies 2 f1 − f2 and 2 f2 − f1 . The parameters describing quantitatively the ratio of useful signal and intermodulation components are more important for the receiver.
Intercept Points on IP2 and IP3 Intermodulation At present, there are three approaches for determining the linearity of radio receivers. Approach 1. Measurement of intermodulation components (IC) of the third or second order, expressed in decibels, with respect to microvolt (dB μV), or in decibels, with respect to milliwatt (dBm). This is the level of disturbing signals acting on the receiver input, which causes the intermodulation components at its output, the level of which is equal to output signal, obtained at introducing the input signal corresponding to the receiver sensitivity level. Approach 2. Intermodulation factor (or the dynamic range on intermodulation) expressed in decibels. For example, if the intermodulation factor is not worse than 70 dB, it means that the disturbing signals should be at the least 70 dB higher than the useful signal in order to create output products of the same level. Approach 3. Intercept points on the intermodulation of third IP3 or the second IP2 order. Sometimes the TOI (third order intercept) abbreviation is used for the designation of the intercept point. Prior to the 1980s, the concepts of intermodulation factor or the dynamic range on the intermodulation were used as a rule for the description of intermodulation properties. However, later on, the almost ubiquitous transition to the intercept point concept was put into practice. This concept turned out to be more convenient due to its universality, since it unambiguously characterizes both the linearity
48
3
Radio Receiver Applications for Radio Monitoring System
and the dynamic range of the receiver (amplifier or any other non-linear circuit). The intercept point value is the rather permanent meaning for the specific receiver, in contrast to the IC level, which depends on the signal level at the input [3, 4]. Third order intercept point IP3 can be calculated for the assumption that the function of IC power of the third order versus the input signal power in the receiver follows exactly the cube law, i.e., at 1 dB increase of the input-disturbing signal level, the third-order products of intermodulation distortions increase by 3 dB. Actually, in accordance with Equation (3.45), the intermodulation third-order product has a level of 3b3 U12 U1 /4 . If two signals with same amplitudes U1 = U2 act at RR input, the output value of the intermodulation product will be proportional to the amplitude cube. In log-scale, this means that the output signal will grow three times faster than the input signal, i.e., the curve has a linear form with the slope 3:1. Second-order intercept point IP2 is defined under the assumption that the function of second-order products power versus the receiver input signals power follows exactly the quadratic law, i.e., at 1 dB increase of the input signal level the intermodulation distortion products of second order will increase by 2 dB. Actually, in accordance with Equation (3.45), the second-order intermodulation product has the level of b2 U1 U2 . If two components with the same amplitudes U1 = U2 enter at the receiver input, the output value of the intermodulation product is proportional to the amplitude squared. In log-scale, this means that the output signal power will grow twice as fast as the input signal power, i.e., the curve will have a linear form with the slope 2:1. At the same time, the useful signal growth at the receiver output should follow the linear law, i.e., at 1 dB increase of the input signal power the output signal power should increase by 1 dB. In log-scale, this means that the output signal power increases at the same rate as the input signal power, i.e., the function has a linear law with the slope 1:1. Figure 3.21 shows the output signal power versus the input signal power for the useful signal (P1 ), the intermodulation product of the second (P2 ) and the third (P3 ) order in log-scale. The third-order intermodulation product power P3 increases three times faster compared to the input signal power P1 . This means that these curves must have an intersection point. It should be noted that in reality the growth rate of these curves reduces as the input signal power decreases. The values do not tend to infinity since the real GTC has the form of the curve with saturation, as shown in Fig. 3.19. However, at small input signals, the curves have practically a linear character. If the curves are extrapolated by the straight lines, they will cross at IP3 point. Similarly, the second-order intermodulation product power P2 grows twice as fast as the useful signal power P1 . Accordingly, the hypothetical intersection point of lines P1 and P2 is referred to as the intercept point on the second-order intermodulation IP2 . In receiver specifications, the second and third-order intercept points are usually defined with respect to the input, i.e., by the input signal power, expressed in decibels with respect to milliwatt (dBm).
Main Radio Receiver Parameters
49
Pout,dBm
Extrapolated
IP2out
IP2 curves IP3
IP3out
Actual curves
P1dB
1dB P1dB
P1
1 dB-compression point
P3 P2 Pin,dB I P3in
I P2in
Fig. 3.21 Second and third-order intercept points
Intercept points IP2 and IP3 usage is the convenient engineering approach, allowing the quantitative estimation of the receiving path linearity, to determine the difference (in decibels) between the useful signal and the intermodulation component level. For example, let the value IP3 be 12 dBm. How much will the useful signal level exceed the intermodulation components level at the input signal power Pin = −10 dBm? The input signal is less than IP3 on input, by IP3 − Pin = 12 − ( −10) = 22 dBm. Hence, the useful signal power P1 is less than IP on the output, by 22 dBm, and the intermodulation components power P3 is less than IP3 on the output, by 66 dBm. The useful signal will be larger than the intermodulation components by 3 = 44dBm. In the general case, the useful signal excess over the third-order intermodulation components can be determined as: 3 = P1 − P3 = (IP3 − Pin ) − 3(IP3 − Pin ) = −2(IP3 − Pin ).
(3.48)
For the useful signal and the second-order intermodulation components, we get 2 = P1 − P2 = (IP2 − Pin ) − 2(IP2 − Pin ) = −(IP2 − Pin ).
(3.49)
For the useful signal and the n-order intermodulation components, we have n = P1 − Pn = (IPn − Pin ) − n(IPn − Pin ) = −(n − 1)(IPn − Pin ).
(3.50)
Moreover, knowing the intercept points, one can estimate the receiver dynamic range, free of intermodulation.
50
3
Radio Receiver Applications for Radio Monitoring System
Intermodulation-Free Dynamic Range Determination Usually, the full receiver dynamic range is defined by Equation (3.39), as the ratio of the largest to the smallest signals received by the receiver. The level of limited sensitivity Ps lim , determined by Equation (3.20), is accepted as the lower limit, and the higher limit is related to the receiver’s characteristics, describing its nonlinearity, such as, for example, the blockage level of 1 dB-compression point (P1dB in Fig. 3.21). The compression point is the absolute limit of the GTC linear part, and it is usually less than IP by 10–20 dB [5]. One additional receiver feature has the greatest practical importance, namely, the dynamic range on the intermodulation. This shows in which input-values range the receiver is able to operate without the intermodulation distortions. Let us look at the plot in Fig. 3.22. The third-order intermodulation dynamic range D3 can be obtained by determining the length of the vertical segment BC. The segment starting point – point B – is the intersection of the straight line P3 with the limited sensitivity level or with inherent noise Ps lim . Point C is situated on the straight line P1 . Since the straight line P1 has the slope 1:1, the triangle ABC is isosceles one. Therefore, the dynamic range D3 can be obtained as the length of the AB horizontal segment from the intersection point of the straight line P1 with the straight line Ps lim till the intersection point of the line P3 with the line Ps lim . As mentioned above, the straight line P3 has the slope 3:1, therefore, the segment BD is three times less than the segment AD. At the same time, the segment AD is equal to the segment DIP . Thus, the third-order intermodulation dynamic range is 2 D3 = (IP3 − Ps lim ). (3.51) 3 Similarly, we get the expression for the second-order intermodulation dynamic range: D2 =
1 (IP2 − Ps lim ). 2
(3.52)
P3
Pout,dB 20
P1
IP3
–20
C
–60 D3 –100
Fig. 3.22 Dynamic range of the third-order intermodulation
D3
A –100
–60 2/3
B –20 1/3
D Ps lim Pin,dB
Main Radio Receiver Parameters
51
In the general case, the dynamic range Dn on the n−order intermodulation is Dn =
n−1 (IPn − Ps lim ). n
(3.53)
where IPn is the intercept point on the n−order intermodulation. Recall that the limited receiver sensitivity can be obtained by Equation (3.20).
Attenuator Influence on the Intermodulation Value A serious question arises: Does the attenuator at the receiver input influence the intermodulation components value? Reducing the receiver input voltage by the attenuator, for example, by 1 dB, decreases the useful signal P1 by 1 dB also, but then the second-order intermodulation products P2 decreases by 2 dB, and the thirdorder products P3 decreases by 3 dB, as shown in Figs. 3.23 and 3.24. This property can be used for estimating the intermodulation component order. Fig. 3.23 1 dB attenuator influence on the second-order intermodulation
1dB
2dB
(f2-f1)
Fig. 3.24 The level variation of the third-order intermodulation components with 1 dB attenuator
f1 f2
(f2-f1)
f1 f2
1dB
3dB (2f1-f2) f1 f2 (2f2-f1)
(2f1-f2) f1 f2 (2f2-f1)
In the general case, by including the attenuator with the transfer function g1 = −A dB, n−order intermodulation components level will be decreased by nA dB. At that, intercept point value on input will be increased by A value, since all straight lines in Fig. 3.21will displace parallel to the right by A value. Since the intercept point displaces to the right by A when including the attenuator, it can appear that receiver linearity will increase. As a matter of fact, this is not so, because, at that point, the receiver’s limited sensitivity becomes worse by exactly the same A value.
52
3
Radio Receiver Applications for Radio Monitoring System
Determining the Intercept Points There are many approaches to determine the intercept points [3]. In most cases, these approaches are based on the measurement of the dynamic range on intermodulation Dn . In one such approach, used for determining third-order intermodulation distortions, two signals enter the input, as shown in Fig. 3.25. One signal displaces from the channel central frequency by 20 kHz, and another by 40 kHz. Third-order intercept point IP is calculated assuming that the third-order IC level depends on the receiver input signal levels following exactly the cube law, i.e., at 1 dB-increase of the disturbing signal level, the third-order IC increases by 3 dB, and the difference by 2 dB. Signal generator 1
Attenuator 1 Adder
Signal generator 2
Radio receiver
Spectrum analyzer
Attenuator 2
Fig. 3.25 Structural diagram of measurement to determine the intercept point on intermodulation
In the general case, to calculate the arbitrary order intercept points on input, one can use the following equations, obtained from (3.48) to (3.50): 3 + Pin 2
(3.54)
IP2 = 2 + Pin
(3.55)
IP3 =
IPn =
n + Pin n−1
(3.56)
where 3 , 2 , n are differences, expressed in decibels, between the useful signal power and the IC power of the third, second and n-th order, respectively; Pin is the input signal power.
Blockage Effect The useful signal blockage (compression) becomes apparent when reducing the gain factor in the input path of the receiver, or, in decreasing SNR, when disturbing signal action, the frequency of which is outside the main channel range. Blockage occurs in the RF path active elements (RFA and tuners), due to the non-linear law of the useful signal transfer function variation, simultaneously with the disturbing signal (Fig. 3.26). Useful signal blockage occurs in this or that receiver cascade if the amplitude signal transfer function has the saturation character, in which case it is as though the
Main Radio Receiver Parameters
53
Fig. 3.26 Explanation of the useful signal blockage effect in the receiver
ftun
fbl
ftun
fbl
output signal increment lags with respect to the input signal increment, in the wide variation interval. The blockage does not occur if the amplitude signal-transfer function has the linear character in the wide interval of the input signal variation. The channel, in which the blocking/disturbing signal acts, is out-of-band; the nominal frequency of such a signal may accept the different values within some frequency band, depending on the disturbing signal level and the RF path circuit’s selectivity before the mixer. At rather large detunings f of the disturbing signal, relative to the receiver tuning frequency, this signal is attenuated by the RF path resonant circuits. To describe the receiver property to detect the useful signal in the presence of the powerful disturbing signal, one may use the concept of the “dynamic range on the blockage” (in decibels): Db = 20
Umax b Us min
(3.57)
where Umax b is the maximal permissible voltage of the disturbing signal, corresponding to the blockage threshold; Us min is the minimal voltage of the useful signal, corresponding to the receiver sensitivity.
Crosstalk Distortions Crosstalk distortions are useful signal spectrum structure variations of the modulated disturbing signal, occurring at simultaneous impact to the receiver, the frequencies of which do not coincide with the main and the spurious receiving channels. Such a distortion-occurring process is defined by the non-linear variation of the amplitude signal transfer function in the RF path active elements. The “crosstalk distortions” concept refers to the useful signal with amplitude modulation (AM), when the modulation components of the disturbing AM signal occur in its structure (Fig. 3.27). In this case, the RF path non-linear element – just as for the blockage in RFA – can be presented by way of the power polynomial model, the only difference being that the input signal is imitated by the sum of two AM signals. In order to simplify the analysis, one can limit oneself to the third order polynomial, as at the blockage. After some transformations we can obtain the crosstalk
54
3
Radio Receiver Applications for Radio Monitoring System
Fig. 3.27 Explanation of the crosstalk distortions process
ftun
ftun
fctd
distortion factor. This factor represents the ratio of the spectral component level in the useful signal structure, occurring as a result of the crosstalk distortions, to the useful signal level, for the given parameters of the disturbing and useful signals. To describe the receiver property necessary to detect the useful signal in the presence of the powerful disturbing signal – up to the crosstalk distortion threshold – we can use the concept of dynamic range on the crosstalk distortions. The channel, in which the disturbing signal acts, and which creates the crosstalk distortions, is out-of-band; the nominal frequency value of such a signal may differ within some frequency band, depending on the disturbing signal level and on the RF path circuit’s selectivity before the tuner. As mentioned above, the “crosstalk distortions” concept is often attributed to the disturbing signal affect on the useful AM signal. Nevertheless, it can be more widely understood. The crosstalk distortions may become apparent in the form of the useful signal phase variations (distortions), i.e., in the form of interference in the phase (generally angle) modulation systems. In this case, the term “phase crosstalk distortions” is expedient. Phase crosstalk distortions are not essential for AM systems. They become apparent in the form of interference, in systems where the useful information consists in the phase structure of the received signal.
Phase Noise and Retuning Rate of the Panoramic RR Frequency synthesizers (FS) are used in receivers to vary the tuning frequency. In the general case, FS form the discrete frequency set by means of coherent frequency conversion of the single reference oscillator, usually crystal. The long-term relative stability of any frequency at the output of such a coherent synthesizer is equal to the long-term frequency stability of the reference oscillator. Summarizing the various FS versions, we shall list the most important FS characteristics. Then, we shall take into consideration the widespread case, when the oscillation from one set of equidistant frequencies occurs in the FS output, at each time moment. The main FS characteristics are: the output signal frequency stability, the operating frequency range, the discreteness of the frequency or phase retuning of the output signal, the output signal type, the level of spurious discrete components, the phase noise level, and the switching time.
Main Radio Receiver Parameters
55
The FS operating frequency range is defined by the operating frequency range of the receiver. If a FS is used as the first local oscillator, its frequency should differ from the input frequency of the receiver by the intermediate frequency value. Instead of the frequency range, one may apply the coverage factor of the operating frequency range, which is equal to the ratio of the maximal frequency of the operating frequency range to the minimal frequency of the same range. The frequency retuning discreteness (the step of frequency grid) is defined by the synthesizer’s purpose. At fixed-frequency communication and at the frequency jump tracking, the step of synthesizer frequency grid, used as a local oscillator, is defined by the step of the transmitter frequency grid (from hundreds of hertz to tens of megahertz). At the programmed Doppler shift compensation and with the use of FS as a tracking system (on the phase or in time), the required frequency grid step may be very low (till some hundredth and some thousandth part of hertz). In this, and some other, cases, the phase (not frequency) retuning discreteness of the output oscillation has a higher profile. As a result, the phase jumps are inadmissible during the transition from one frequency to another, and these jumps cannot exceed the definite value (usually some tenth part or parts of a degree). The output signal view is defined to a great extent by the signal processing character in the device in which a FS is used. For the synthesizer to be used as a local oscillator of the analog receiver, the sinusoidal form of the output signal is usually required. For a FS in digital-analog tracking systems, the pulse form is preferable. Finally, in the path with completely digital signal processing, a FS must generate the number sequence (codes) corresponding to the sinusoidal signal samples in the fixed (equidistant) time moments. In technical specifications, the phase noise spectral density is usually given in decibels, by hertz, with respect to carrier level at the given offset from the carrier frequency (or dBc, for example, –120 dBc/Hz at 10 kHz offset). The phase noise level of the FS output oscillation, with respect to the generated signal level (the carrier level), usually falls within –60 to –120 dB/Hz at 10 kHz offset from the carrier frequency. At frequency multiplication with FS, the spectral density increases proportionally to the multiplication factor, applying hard restrictions on the reference oscillator noises. The frequency switching time (permissible) varies widely depending on the synthesizer’s purpose. In particular, when a FS is used in a communication receiver at different frequencies, the switching time may be equal to one second. In that case, during the frequency switching, not only are phase jumps permitted, but the complete short-term miss of signal is permitted as well. On the other hand, at FS application in the phase-locked loop, and in some other cases, it is desirable to ensure the complete absence of the transients. For panoramic receivers, the frequency switching time defines the receiverretuning rate over the operating frequency range. For modern panoramic receivers, the switching time is equal to units of milliseconds. We should mention that the less the frequency switching time is, the more difficult it is to ensure the low phase noise level.
56
3
Radio Receiver Applications for Radio Monitoring System
The frequency accuracy of the receiver frec includes both the initial error of the given tuning frequency setting and the receiver tuning instability. The setting error depends on the setting method and the tuning frequency indication method, and the tuning instability depends on the tuning frequency offset due to the system warming-up, the climatic and mechanical impacts, the supply voltage variations, etc. The high frequency accuracy of the receiver is necessary for the preset communication entry, the communication maintenance without adjustment. For measuring receivers, frequency accuracy is the governing factor for the accurate measurements of the radio signal frequency. The technical specification may fix the receiver’s frequency accuracy, or, separately, the setting error and the tuning instability. In some cases, tuning instability is defined in parts, due to the necessity of separate calculation and testing: the selfwarming-up, the temperature variations, the shocks and vibrations, the variations of the supply power voltage. The absolute frequency instability of the reference oscillator (RO) is the oscillation frequency deviation f at its output during the definite time interval, caused by the external destabilizing factor influences, against the specified nominal frequency f0 : f = f − f0 . The strictest requirements for frequency accuracy are placed on the receivers, intended for radio signal receiving with single-sideband (SSB) modulation (frec = 5 − 10Hz) and the signals with differential phase modulation (frec = 0.5 − 1Hz). This follows from the fact that the relative frequency accuracy of the receivers should have the value 10−7 − 10−8 . When the receiver uses the frequency stabilization system with one RO, it complies with the requirements for relative frequency accuracy defined by this oscillator. The relative RO frequency instability is the ratio of the absolute frequency instability to the specified frequency: δ = f /f0 . To reduce from the total frequency error the error portion caused by the initial frequency setting inaccuracy, the possibility of RO frequency correction on the basis of the external frequency standard or the operation from the external RO, which is more accurate and stable, should be provided. Tuning instability of the receiver can also occur via the permissible variation of the receiver’s local oscillator frequencies in the time interval after the self-warmingup. The temperature coefficient of the local oscillator is the important parameter, i.e., the relative frequency offset at air temperature 1◦ C variation around the receiver. The long-term RO frequency instability is the total frequency deviation caused by its slow variation due to element aging, the external destabilizing factor influence. It is defined during the long-term period, namely, an hour, round the clock, a month, a year.
Digital Radio Receivers General Principles of Digital Radio Receiver Implementation The digital RR is the RR that performs the signal processing in analog and digital forms [6–9].
Digital Radio Receivers
57
The complete or partial digital signal processing (DSP) is executed in the digital RR (DRR), intended for receiving the analog signal. In accordance with DSP applications, DRR can be divided into two groups: • Receivers, where the received signal conversion into the digital form is not executed, and the separate units are implemented on the digital element base, e.g., the control devices, the monitoring devices, the information image devices, the systems for establishment of communication, the automatic control systems, the digital frequency synthesizers, etc. • Receivers, where the signal is converted into the digital form and part of the main receiving path is fulfilled on the digital element base, including the digital filter for the main signal selection, the digital demodulators, the digital devices for signal recognition and parameter measurement, and the auxiliary units. The first group of RRs is the most numerous, at present. The majority of modern, professional, RRs have separate units implemented on the digital base of their structure. However, receivers in the second group are also considered receivers, where the preliminary signal filtering, its amplification, and the frequency conversion to the intermediate frequency are executed in the analog domain; and, after that, the IF signal is subjected to the analog-digital conversion and all further signal processing is executed in the digital domain. Practically all, modern, RRs intended for radio-monitoring problems can be classified as the second group. The generalized structural diagram of DRR consists of five functional units, as shown in Fig. 3.28. The amplifying-converting path (ACP) accepts the signal from the antenna, provides filtering against the interference, displaces the input signal spectrum to the IF, at which time the analog-digital conversion is executed. The auxiliary units: the automatic gain control (AGC) system, the attenuators, the limiters, etc., which affect From reference oscillator Frequency synthesizer From antenna
Amplifyingconverting section
DSP section
Secondary power supply
Fig. 3.28 Generalized structural diagram of DRR
Control and display device
To customer
58
3
Radio Receiver Applications for Radio Monitoring System
the amplitude characteristics of the amplifying section, but do not introduce distortions into the received information, may be included in ACP structure. The main signal processing is executed in the DSP section. It includes the filter, defining to a considerable degree the noise immunity of the receiver, the demodulator, and the circuits of post-detector signal processing. The frequency synthesizer converts the frequency of the external or inherent reference oscillator and, from this signal, generates the frequency grid necessary for operation of the ACP tuners. The synthesizer permits the receiver to adjust on another input frequency. The separate synthesizers may belong to the tracking system structure. Moreover, the synthesizer can generate the frequency grid required for the DSP unit’s operation. In the autonomous mode, the control and display unit executes the given algorithm of the receiver operation (switching on, switching out, search and choice of the signal, adapting to the changing operating conditions, etc.) and allows the operator to control the receiver manually or automatically. The secondary power supply is intended for the energy conversion of the primary power source, e.g., stationary grid (220 V) or on-board grid, into the form suitable for application directly in RR.
Types of ARM Receivers In spite of the generality of these operation principles, it is possible to single out several characteristic types of RRs, as listed in Table 3.2. These types are used, at present, for ARM problems solving. High real sensitivity and selectivity, as well as the application of methods providing noise immunity and reliability under the conditions of strong pulse, fluctuation, and concentrated over the spectrum interference, are typical for scanning receivers. As a rule, the scanning receiver can be assigned to the first DRR group. The selective micro-voltmeter is a voltmeter equipped with an adjusted narrowband filter, and, thanks to this filter, the voltmeter can measure frequency band voltage, up to the separate spectrum components. The most advanced selective microvoltmeters differ from spectrum analyzers by the manual adjustment only, as well as by the absence of panoramic representation. Selective micro-voltmeters allow us to measure the signal level at the antenna path output in the given bandwidth for the wide operating frequency range. The spectrum analyzer (SA) is a universal measuring device designed for the investigation of the signal spectral structure and the measurement of its parameters. The spectrum analyzer structure coincides with the superheterodyne receiver structure. Depending on the input signal sensor type (with the appropriate matching circuits), the SA can be used in various areas of science and technology. In particular, when using the antenna (as the radio signal sensor), the SA plays the role of the panoramic receiver. Upon additional application of the antenna to the SA – the circuits of preliminary selection – such a SA is then capable of solving the problems of the panoramic measuring receiver.
Digital Radio Receivers
59
Table 3.2 Types of radio receivers used for radio monitoring Spectrum imaging Calibration Demodulator
Signal parameter
Defined by signal bandwidth for which receiverreception is designed (from hundreds of Hz to hundreds of kHz) Adjustable (from hundreds Hz to hundreds kHz)
Usually no
No
Yes
No
Usually no
Yes
Desirable
Yes
Usually adjustable
Yes
Usually yes
Desirable
Yes
Wide. Usually from hundreds kHz to tens MHz Adjustable (from tens Hz to tens MHz)
Yes
No
Usually yes
Usually yes
Yes
Yes
Usually yes
Yes
Preselector Bandwidth Scanning radio receiver
Yes
Selective Yes microvolt meter Spectrum Usually anano lyzer Panoramic Yes receiver
Panoramic Yes measuring radio receiver
In spite of all the mentioned SA advantages, its application for solving the ARM tasks is not always expedient, since the basic technology of the analyzer implementation is, as a rule, intended for realization of universal functional possibilities. In the basic configuration, a SA, as a rule, has no preliminary signal selection units, therefore, its application for ARM problems is rather difficult. The panoramic receiver is a RR with a wide bandwidth (from hundreds kHz to some tens MHz), capable of representing the signal spectral structure, and with the high speed of spectral analysis (from hundreds MHz to tens GHz per second). As a rule, the panoramic receiver has the demodulators of AM, FM, PM, and SSB signals and the interface for connection to a PC. The panoramic measurement receiver is a panoramic receiver with high metrological characteristics for the measurement of the level and other radio signal parameters. The measuring receiver is the “heart” of the advanced radio-monitoring unit. At present, the hardware of ARM equipment is limited by the antenna complex and the measuring receiver. As a rule, the set of peak, quasi-peak and RMS detectors designed for signal level measurements is included in the measuring receiver structure. The measuring receiver must provide the frequency resolution, from several hertz to tens of kilohertz, at the signal spectral analysis, and must operate under PC control.
60
3
Radio Receiver Applications for Radio Monitoring System
The main type of measuring receiver is the voltage and power measuring system for RF signals and noise, which operates for the given receiving antenna parameters as the field-strength meter. It is a superheterodyne receiver with a measuring device at the output, which has high sensitivity (up to 10–15 W) and selectivity (50–60 dB). As a rule, the measuring receiver is a multi-range receiver with bandwidth adjustment in the IF section. The linear receiving path operates to the linear and quadratic amplitude detector with the given time constants of the load circuit, which allows for the possibility to measure the averaged rectified, effective, and peak values of sinusoidal and noise signals.
Development of Russian Arm Systems First- and Second-Generation Systems Prior to the 1990s, automated radio monitoring systems used in Russia were based on foreign scanning receivers. In the first-generation systems, the foreign receivers were used without serious updating, but, in the second-generation systems, serious modernization in receivers was made, which allowed for the improved technical performance of ARM systems, in which they were applied. In the first-generation systems, the receiver update was reduced to the additional buffer arrangement, so that the analog IF outputs could pass the signals to the external digital processing unit. Moreover, a switch was added as an AGC circuit break, to ensure the possibility of operating in the multiple-pass panoramic coverage mode with the permanent path transfer factor. More serious updates were provided in the second-generation systems, including the arrangement of additional units of “fast,” properly developed and manufactured, synthesizers, permitting the time of receiver adjustment to the given frequency to be reduced substantially and the panoramic coverage rate to be increased.
Fig. 3.29 Central unit of ARK-PK3KU automated radio monitoring system (AOR-3000 radio receiver)
Development of Russian Arm Systems
61
At that time, the following ARM equipment manufacturing was launched, namely, the handheld direction finders, the multi-channel RM systems, the various ARM systems, the distributed systems of remote RM to distant premises and the reveal of the information leakage channel, multi-functional RM and DF systems, and so on. Figures 3.29, 3.30, 3.31, 3.32 and 3.33 show some of these systems.
Fig. 3.30 Central unit of ARK-PK3KU automated radio monitoring system (AOR-5000 radio receiver)
Fig. 3.31 Control unit and receiver of the handheld ARK-RP1 direction finder (AOR-3000 radio receiver)
62
3
Radio Receiver Applications for Radio Monitoring System
Fig. 3.32 ARK-KPC six-channel radio monitoring system (two IC-8500 radio receivers and four IC-PSR1000 radio receivers)
Fig. 3.33 ARK-RD4 four-channel radio monitoring system (four IC-PCR1000 radio receivers)
Radio Receivers of the Third and Fourth Generation Experience gained from application of the hardware-software ARM systems of the first and second generation showed the total accuracy of the selected hardwaresoftware solutions, but, at the same time, the following restrictions were discovered:
Development of Russian Arm Systems
63
• Large irregularity of the gain factor and the inherent noise within the operating frequency range, which could not allow the system to be applied for measuring purposes • Insufficient dynamic range in the wide-band path • Impossibility of optimal receiver control in the system structure, leading to operation rate reduction • Low tuning rate, which could not allow the 50 MHz/s limit to be overcome • Impossible operation in the RM structure required the wide-band IF output • Large number of staggered frequencies falling into IF bandwidth • Absence of double-channeled coherent receiving possibility from two receivers, without serious updating • Extent of the electromagnetic compatibility problems for the receivers and for other system units • Structural features of receivers, prohibiting application within the systems, due to the complicated requirements for the mechanical and climatic operation conditions. Taking these shortcomings into consideration, Russia decided to launch the development of a new DRR generation, which would ensure the following features in the interests of ARM systems: • Possibility for use in multi-channel coherent receiving systems • Dynamic range within wide-band path on the second- and third-order intermodulation not less than 70 dB • Retuning duration from one frequency to another not more than 15 ms for frequency setting accuracy 500 Hz • Digital processing bandwidth not less than 2 MHz • Spectral analysis rate not less than 100 MHz/s, for the spectral sample discreteness 3 kHz • Increased firmness to the mechanical and climatic impacts permitting DRR application in the equipment of various purposes based on the transport carrier • Minimization of the consumed power. As a result of the intensive development of new circuitry and design solutions, in 1999 a new Russian DRR of the third generation came into the world. One of its implementations is shown in Fig. 3.34. The developed ARK-CT1 receiver had the acceptable technical specifications, including dynamic range on the third-order intermodulation not worse than 70 dB, synthesizer switching time of about 10 ms, which, at 2 MHz bandwidth, enabled a digital spectrum analysis rate of 140–150 MHz/s at the spectral sample discreteness 3.125 kHz. The irregularity within the receiver bandwidth did not exceed 1.5 dB. The receiver’s consumed power was not more than 30 W. On the basis of that RR, the third-generation ARM equipment was manufactured including RM, DF, and CEE equipment, described in subsequent chapters. High specifications of the receiver shown in Fig. 3.34, and its further modernization
64
3
Radio Receiver Applications for Radio Monitoring System
Fig. 3.34 DRR ARK-CT1 of the third generation
Fig. 3.35 ARK-D1TP panoramic measuring receiver
permitted the creation of the panoramic measuring receiver (Fig. 3.35), certified in 2002 by the authorized Russian committee as measuring equipment. The constantly growing requirements for RM, CEE and DF complexes, in combination with the successfully-solved problem of full-scale production of the singlechannel and double-channel DRR of the third generation, made for the development of the fourth DRR generation in Russia, with an extended set of functional possibilities and higher specifications. The main requirements for such DRR are listed below: • Operating frequency range 0.01–3 GHz • Synthesizer retuning time not more than 5 ms • Presence of built-in digital unit for radio signal demodulation and technical analysis • Increased one-signal and multi-signal selectivity especially in the upper end of the operating frequency range
Development of Russian Arm Systems
65
• Extended possibilities regarding digital signal processing, due to the application of multi-processor units and the rate increase of information exchange with the control computer • Receiving section bandwidth not less than 4 MHz • Spectrum analysis rate not less than 500 MHz/s • Implementation of the modular construction principle, in particular, rackmounted realization on the basis of the Compact PCI standard constructions • Possibility for the coherent combination of receivers, along with the opportunity to equalize the amplitude-phase characteristics of the receiving paths. At the design step, special attention was devoted to the development of functionally and constructively completed modules, which could be easily combined in the various system types by means of different unifying plates and under-frames. As a result of this development, the new ARK-CT3 digital receiver was delivered into the Russian market in 2003. The frequency range of the new receiver is 9 kHz–3 GHz. The synthesizer switching time is about 5 ms, and a bandwidth is equal to 5 MHz, which allows for the possibility to calculate spectral panorama with a rate of more than 700 MHz/s. The dynamic range on the third-order intermodulation is not worse than 75 dB. This new receiver was used for the creation of ARM equipment of the fourth generation, including the ARK-D7K double-channel RM and CEE system, in ARK-MK1, ARK-MK2 mobile and stationary direction-finding systems, and in a new ARM station called ARGUMENT. The external view of the ARK-MK1 system central unit, on the basis of two coherently connected receivers, is shown in Fig. 3.36.
Fig. 3.36 Central module of ARK-MK1 system with the double-channel DRR of ARK-CT3 Type
The external remote-controlled radio signal converter, providing the higher limit enhancement of RM operating range till 18 GHz, was developed and designed in parallel with this receiver development. This frequency converter assures suppression of the spurious receiving channel up to 40 dB, has a noise factor of not more than 14 dB, can operate both with the internal directional antenna system, and with the external antenna. Its external view is shown in Fig. 3.37. In 2004, this external converter was certified as measuring equipment.
66
3
Radio Receiver Applications for Radio Monitoring System
Fig.3.37 ARK-KNV4 external frequency converter
Fifth-Generation Radio Receivers Unfortunately, equipment based on the DRR ARK-CT1 and ARK-CT3 required rather large weight and dimensions, in order to effectively apply it in the carried systems. Taking into consideration the cumulative experience on development and manufacture of the direction finders and the equipment for handheld DFs, as well as the experience of equipment-mounting on various ground and air carriers, in 2003,
ARK-CT1 Digital Radio Receiver
67
the following main requirements for the portable DRR of the next (fifth) generation were formulated in Russia: • Receiver design should include two modules: radio frequency (RF) – very high frequency (VHF) – ultra high frequency (UHF) radio signal frequency converter and the digital signal processing module • Basic size of the module plate should be accepted as 100 × 160 mm, which is the world standard for industrial equipment • Module implementation must ensure its integration into the user’s equipment • DRR should be used in multi-channel coherent receiving systems • DRR should have sufficient stable metrological characteristic for application as measuring equipment. The requirements for class B ARM equipment were selected as the technical requirements for new DRR. At that time, some additional restrictions were imposed concerning the consumed power, namely, the analog radio signal frequency converter should consume not more than 8 W, and the DSP module not more than 10 W. In the middle of 2004, after 12 months of intensive development, the first versions of the fifth generation DRR arrived and were named ARGAMAK. In 2004–2005, on the basis of the ARGAMAK family, the fifth generation of RM, DF and CEE technical equipment was developed, including the stationary, mobile, portable and carried equipment.
ARK-CT1 Digital Radio Receiver Let us consider the structure of Russian DRR of the third generation of ARK-CT1 type with the bandwidth of 2 MHz. Constructively, this receiver consists of two units: the analog radio receiving path of CT1 type and the digital signal processing unit of AC01 type, as shown in Fig. 3.38. Technical characteristics are presented in Table 3.3. Fig. 3.38 Structural diagram of ARK-CT1R
ARK-CT1 IF 10,7 MHz Input
CT1 unit
DSP unit
LPT COM
PC
The structural diagram of the RR section is shown in Fig. 3.39. The main section parts are: the pre-selectors unit; the mixers unit; the frequency synthesizers unit with the reference oscillator; and the control unit. To achieve high selectivity on the combination channels, it is necessary to ensure good signal selection in the pre-selector. The usage of the band-pass filters with fixed bandwidth is acceptable only in the case of rather narrow operating ranges of the receiver. Since the main area of this receiver application is radio monitoring in
68
3
Radio Receiver Applications for Radio Monitoring System
Table 3.3 Technical specifications of ARK-CT1 DRR Parameter name
Value
General parameters Operating frequency range of the basic set, MHz Input attenuator, dB Maximal permissible input signal, dBm Noise factor, dB: In range 25–1,000 MHz above 1,000 MHz Relative frequency instability of the reference oscillator ◦ ◦ Relative error of frequency setting (–20 C to +50 C) Tuning time of synthesizer, ms, not more than LO phase noise at offset 10 kHz, dB/Hz Selectivity and non-linear distortions Interference suppression at IF, dB, not less than Selectivity on the image channel, dB, not less than Dynamic range on the 3rd and 2nd order intermodulation, dB, not less than Gain factor irregularity within operating frequency range, dB, not more than
20–2020 10, 20, 30 23 not > 14 not > 16 5 · 10−6 10–6 10 –95 70 70 70 ±3
IF signal Analog IF signal frequency, MHz Bandwidth before IF output, MHz Demodulator frequency setting discreteness, Hz
10.7 2 1
Operating temperature, weight, dimensions, power consumption ◦ Operating temperature interval, C Supply voltage, V Consumed power, VA, not more than Dimensions (width × height × depth), mm Weight, kg
–10 to +60 24–30 18 300 × 65 × 255 not > 4
the wide frequency range, a set of ten tracking filters is used in pre-selector, which are switched by the analog switchers (AS), for the frequency sub-ranges: 20–35, 35–60, 60–100, 100–170, 170–240, 240–333, 333–465, 465–700, 700–1012, 1,012– 2,020 MHz. The medium frequency coverage factor is about 1.58. Within the frequency range of 20–1012 MHz the band-pass filters (BPF) are used, and in the range of 1,012–2,020 MHz the high-pass filter (HPF) with the cutoff frequency defined by the receiver tuning frequency. The filters bandwidth in the range of 20–1,012 MHz on –3dB-level is about 10%, and the bandwidth on –70 dB-level is 50–150% from the receiver tuning frequency. Figure 3.40 shows the experimental curves of the pre-selector transfer functions obtained at the testing of the receiver commercial specimen at the tuning frequencies 20 and 701 MHz. The control over switchers and pre-selector filters is fulfilled by the special controller. During factory adjustment for each individual receiver, a record of the controlling voltages, made into the controller, is executed in sixteen points of each
ARK-CT1 Digital Radio Receiver
69
Pre-selector unit
S w i t c h
Antenna input
Mixers unit
HPF (1012–2020 MHz)
Amp
HPF
Amp
BHPF
BPF (700–1012 MHz)
Amp
HPF
Amp
BPF
BPF (465–700 MHz)
Amp
HPF
Amp
BPF
BPF (333–465 MHz)
Amp
HPF
Amp
BPF
BPF (240–333 MHz)
Amp
HPF
Amp
BPF
BPF (170–240 MHz)
Amp
BPF
BPF (100–170 MHz)
Amp
BPF
BPF (60–100 MHz)
Amp
BPF
BPF (35–60 MHz)
Amp
BPF
Amp
BPF
BPF (20–35 MHz) Attenuator
S w i t c h
S w i t c h
Mixer 2 BPF
Amp
BPF
BPF
Amp
BPF
Switch Mixer 2
BPF
Amp
BPF
Mixer 3
BPF
Amp
Analogous IF output 10,7 MHz
Pre-selector unit controller
From ACO1 unit
M i x e r 1
Amp
Amp
Amp
Amp
Osc
Osc
Osc
Osc
Synthesizers unit
Synthesizers unit controller
Control unit
Fig. 3.39 Structural diagram of CT1 unit K/K0,dB –10
K/K0,dB –10
–30
–30
–50
–50
–70
–70
a)
10
20
30 f, MHz
b)
700
900 f, MHz
Fig. 3.40 Pre-selector frequency responses at the tuning frequencies of 20 MHz (a) and 701 MHz (b)
frequency sub-range. Obtaining the array in such a manner ensures filter fine-tuning at each tuning frequency of the receiver. In receiver operation mode, the voltages applied to the filter variable capacitance diodes from the outputs of the controller DAC unit are calculated by means of linear interpolation of the stored controlling values. Indexes, by which the controlling value selection is executed, are calculated on the basis of the receiver tuning frequency. A similar approach is used to maintain the regularity of the pre-selector transfer function. The transfer function irregularity does not exceed 3 dB at each frequency within the receiver operating range. To achieve the pre-selector tuning automation, the specific process software applications are used together with the specific hardware. The external view of pre-selector of ARK-CT1 DRR with removed shielding is shown in Fig. 3.41. As shown, the attenuator at the receiver input regulates the input signal attenuation by 0, 10, 20, or 30 dB. The next important element in the receiving path determining the resolution capacity and the receiver’s tuning rate is the frequency synthesizer unit. The single reference frequency 12.8 MHz from the reference oscillator with the crystal frequency stabilization, and with the temperature stabilization, is applied to the synthesizer unit. DRR retuning with 10 kHz-step is executed by the first LO
70
3
Radio Receiver Applications for Radio Monitoring System
Fig. 3.41 ARK-CT1 RR pre-selector
retuning, and the 500 Hz-step tuning is executed by the third LO retuning. The tuning with 1 Hz-step is fulfilled in the DSP unit. We can attribute the suppression method of the spurious harmonics penetrating into the IF section to the characteristic properties of the synthesizer units of the ARK-CT1 and ARK-CT2 receivers. This can be achieved by means of the simultaneous frequency offset of the 1st and 2nd LO at the given receiver tuning to the selected in advance offset. As a result, within the whole receiver operating range, from 20 to 2,020 MHz, the spurious harmonics are absent. Another synthesizer peculiarity is the frequency correction possibility of the built-in reference oscillator. The relative frequency instability of a RO in the temper◦ ature range from 0 to +50 C achieves 3 × 10-6 . RO program frequency correction, stipulated for the application as being in the widened temperature range, provides a reduction of temperature instability not less than three times. To increase the receiver frequency stability, the external reference oscillator, on 10 MHz or 12.8 MHz, can be connected to the receiver. The switching between internal and external RO is executed by the program approach. The synthesizer unit of the receiver provides the frequency setting, with the accuracy not worse than 250 Hz for less than 15 ms. One of the important factors defining receiver resolution capacity is the phase noise of the frequency synthesizer. In ARK-CT1 DRR, the phase noise level is mainly caused by the first LO noise, which increases with its frequency growth. Due to the fact that range coverage is provided by the mutual operation of all LOs, the noise distribution over the tuning range depends on the tuning frequency, but does not exceed 95 dB/Hz. Figure 3.42 shows a typical picture of spectral density of the phase noise power with respect to generated signal frequency offset. The receiving path is designed on the basis of the superheterodyne receiver with three frequency conversions. The fourth frequency conversion is fulfilled in the analog-digital signal-processing unit. The first IF value depends on the frequency of the received signal and is equal to 712 MHz or 302 MHz. The second IF value is equal to 45 MHz, and, lastly, the third IF value is equal to 10.7 MHz. DRR has the buffered third IF output, which can be used for the additional connections, for example, for radio signals decoders. Receiver selectivity on the adjacent channel is defined by the characteristics of the IF filters used. Therefore, when designing the DRR analog path, much attention
ARK-CT1 Digital Radio Receiver Fig. 3.42 Phase noise spectrogram at IF3 output for 460 MHz tuning frequency
71 N, dBc/Hz –20 –40 –60 –80 –100 –15 –10
–5
0
5
10
15 f, MHz
was paid to the development of the IF filters. The resulting filter parameters are listed in Table 3.4. Such filter applications ensure the suppression of the image and spurious channels, occurring as a result of the spectra transition, by not less than 70 dB. The typical value of the combination interference suppression level is equal to 90 dB. Table 3.4 IF filters selectivity characteristics Parameter name AFR irregularity in bandwidth, dB, not more than –1 dB-bandwidth, MHz –70 dB-bandwidth, MHz
IF1 302 MHz
IF1 712 MHz
IF2 45 MHz
IF3 IF4 10.7 MHz 1.6 MHz
±0.5
±0.5
±0.5
±0.5
±0.5
17 60
24 90
6 20
3.2 6.4
2 3.2
The structural diagram of the ACO1 unit is shown in Fig. 3.43. The unit consists of two main assemblies of the 3rd IF 10.7 MHz frequency converter into the 4th IF 1.6 MHz; ADC; the digital signal processor; the controller unit. ACO unit 10.7 MHz input from CT1 unit
CSP
ADC
AO unit 10.7 MHz channel
Converter
1.6 MHz channel
DSP
+12V Controller USB
To controller of CT1 unit
To PC
Fig. 3.43 Structural diagram of ACO1 unit
COM
Buffer Power supply +27V
+27V
DAC
From power supply To CT1 unit
LPT
To PC CSP testing output
72
3
Radio Receiver Applications for Radio Monitoring System
The signal at 1.6 MHz frequency is exposed to analog-digital conversion in 12-bit ADC with the sampling frequency 6.4 MHz. The fourth IF value 1.6 MHz is related to ADC sampling frequency. The selected relation between the input frequency 1.6 MHz and the sampling frequency, equal to 1:4, allows the algorithms to be used to reduce the computing operation number when obtaining the digital signal complex envelope. The operations related to the spectrum transition from the 3rd IF to the 4th IF, are shown in Fig. 3.44. AFRBPF
Spectrum transition
AFRRR
AFRIF
0
1.6 2
AFRRR
0
1.6 2
3.2
6.4
10.7 RR IF
Image channels of the sampled signal (without filtering)
F,MHz
Operating bandwidth
3.2
6.4
F,MHz
Image channels of the sampled signal (with Transition LF band filtering) Transition HF band
Fig. 3.44 Transition to IF 1.6 MHz
Before AD conversion, the signal in the 1.6 MHz channel is filtered additionally in the band-pass filter, which has a 2 MHz bandwidth. With respect to the bandwidth central frequency, at 2.6 MHz offset, a suppression of not less than 80 dB is ensured. The filter transfer function irregularity within the bandwidth does not exceed 1 dB. From the band-pass filter, the signal passes at ADC input. After ADC, the digital signal enters the digital signal processor, manufactured by Analog Devices Ltd. The processor’s main purpose consists of signal spectral analysis on the basis of fast Fourier transform (FFT). The length of a typical spectrum is equal to 1,024 complex samples. The spectral sample discreteness is 3.125 kHz. To reduce the Gibbs effect, a Kaiser-Bessel window is used to weight the input time sample. At selected sampling frequency, the equivalent noise bandwidth of this window is equal to 6 kHz. The calculation time for the 1,024 sample complex spectrum does not exceed 5.5 ms. If necessary, the time sample length used in calculation
ARK-D1TP Digital Panoramic Measuring Receiver
73
may achieve 16,000 samples. In a DSP unit, additional kinds of processing can be executed, such as signal detection, phase difference determination, peak, quasi-peak and rms detection, and digital demodulation. Information exchange with the PC is fulfilled through the printer parallel port. At that time, the EPP, Bi-Di or SPP protocols can be used. The ACO1 unit has an additional feature: DSP program loading is executed from the PC. This presents the possibility for fast DSP program changes directly during the unit’s operation. The ARK-CT1 digital radio receiver has within its structure a secondary impulse power supply, ensuring that the receiver can operate from power sources with the voltage range 9–33 V and not corrupt the sensitivity of the receiver analog paths. The inherent temperature regime in RRs is provided by the hardware-software controller, by adjusting the internal warming elements and the external forced aircooling used in combination with the heat radiators mounted on the receiver’s case. ARK-CT1 DRR can successfully operate under conditions of considerable mechanical impacts. It is stable to vibration loading in the frequency range from 10 to 55 Hz, shift amplitude 0.15 mm and to shock load with 10 g peak acceleration at shock impulse duration 16 ms. On the basis of the ARK-CT1 receiver, it was possible to develop the ARKCT2 double-channel coherent receiver, which consists of two identical ARK-CT1 receiver paths and the double-channel analog-digital processing unit. The above-mentioned properties of ARK-CT1 DRR made it possible to use it as a hardware base for the ARK-MK1–ARK-MK6 mobile and deployed RM and DF systems, and in ARK-D1T single-channel CEE complexes and ARK-D7 doublechannel CEE complexes (see below).
ARK-D1TP Digital Panoramic Measuring Receiver The high stability of ARK-CT1 DRR characteristics gave rise to the possibility to create the ARK-D1TP panoramic measuring receiver on its basis [10]. The ARK-D1TP panoramic measuring receiver is a Russian receiver certified as measuring equipment by the authorized state institution. As the ARK-CT1 DRR, this receiver is designed for operation in the frequency range from 20 to 2,020 MHz in the automated RM system structures. To widen the operating frequency range till 18 GHz, one can connect to this receiver the ARK-KNV4 external converter, which is also certified as measuring equipment. The receiver is manufactured in the form of portable, desk-size equipment and consists of two units: the CT1 unit and the ACO1 unit. The main technical specifications of ARK-D1TP are listed in Table 3.5. The possibility of using this DRR as measuring equipment is achieved in the first place due to the automation of the AFR adjustment process in DRR during factory adjustments and the possibility of equipment calibration during normal operation. In the D1TP panoramic measuring receiver the possibility of forming and using the calibration system files “as a whole” is realized. In this case, the calibration peculiarity is the calibration files usage, which contains the correction factors array for the frequency and temperature lists.
74
3
Radio Receiver Applications for Radio Monitoring System
Table 3.5 Technical specification of ARK-D1TP panoramic measuring receiver Parameter name
Value
General parameters Operating frequency range of the basic set, MHz Operating frequency range in maximal configuration (with ARK-KNV4 converter), MHz Limit of permissible error of level measurement, dB Frequency stability External reference oscillator frequency, MHz VSWR on input Maximal measured signal at receiver input, dBm Panoramic analysis speed in operating range, MHz/s Selectivity and non-linear distortions Dynamic range of measured radio signals, with built-in attenuators, dB Dynamic range on 3rd and 2nd order intermodulation, dB Image and spurious channels attenuation, dB Sensitivity restricted by noise (S+N)/N till 6 dB, dBm At frequencies 20–1,012 MHz
20–2,020 0.009–18,000 ±3 2 × 10–6 12.8 Not more than 3 –7 (100 dBμV ) 150 100 70 70
At frequencies 1,012–2,020 MHz
–110 (–3 dBμV , 0,71 μV ) –107 (0 dBμV , 1.0 μV )
Weight, dimensions, power consumption Supply voltage, V Consumed current, A Weight, kg Dimensions (width × height × depth), mm
27 ±3 1.2 7.5 340 × 130 × 260
The complex calibration is fulfilled with the help of the standard signal sources, attenuators and connectors, the errors of which are 3–5 times less than the main permissible error of the complex. The calibration is executed at several reference points of the frequency range by means of the transfer function variation till the nominal reading achievement. The amendment calculation between two reference points is conducted on the basis of the linear interpolation method, which essentially simplifies the correction algorithm. Having increased the reference point number, one may decrease the bias error up to the level defined by the calibration process error. The calibration allows for the execution of a serviceability check for the hardwaresoftware system and the reduction of its bias error. The interpolation allows us to obtain, with the given margin of error, the correction values falling in the interval between two reference points. The interpolation accuracy depends on the argument variation interval (t2 − t1 ) and the function increment F(t2 ) − F(t1 ). At linear interpolation, the value F(t12 ) in the arbitrary point t12 between the points t1 and t2 is determined as:
ARK-CT3 Digital Receiver
75
F(t12 ) = F(t1 ) +
(t12 − t1 )[F(t2 ) − F(t1 )] . t2 − t1
(3.58)
Calibration by the linear interpolation method between the reference points is executed in the ARK-D1TP receiver at the stage of the pre-selector unit adjustment. To satisfy the standard requirements, the peak, quasi-peak and rms detectors are software realized. They operate over the signal time sampling on IF. Before the signal is detected, it is exposed to filtering in the band-pass digital filter. The digital filter bandwidth can be defined from the list. In this list, there are bandwidths of 9 and 120 kHz, which are recommended by state standards. The possibility is stipulated to change the time constants of charge or discharge for the quasi-peak detector. For this, the inertial part of the quasi-peak detector is implemented in the form of the 1st-order filter digital model with the transfer function K(p) =
1 , 1 + pτ
(3.59)
where p is a complex variable, and τ is the filter time constant. When using the integration by trapezium method, the transfer function of the discrete filter will have the form:
K(z−1 ) =
1 2 1−z−1 Td 1+z−1 τ
+1
=
Td −1 Td +2τ (1 + z ) Td −2τ −1 +1 Td +2τ z
(3.60)
where z is a complex variable, Td is the IF signal sampling period, and τ is the time constant. The difference equation of this digital filter can be written as: y[k] =
Td − 2τ Td (x[k] + x[k − 1]) − y[k − 1] Td + 2τ Td + 2τ
(3.61)
where x[k] is the signal sample at filter input, y[k] is the signal sample at detector output, k is a number of time sample. The time samples period is equal to Td . If the input signal x[k] increases, filter coefficients are used that are calculated on the basis of the recharge time constant; if the x[k] signal decreases, filter coefficients are used that are calculated by the discharge time constant.
ARK-CT3 Digital Receiver Compared to the ARK-CT1 DRR, the ARK-CT3 receiver has the more advanced characteristics of receiving path linearity, selectivity, the operating frequency range, the bandwidth and the frequency-retuning rate. The dynamic range on the 3rd order intermodulation is 75 dB, the operating frequency range is from 9 kHz to 3 GHz, the receiving path bandwidth is 5 MHz, the synthesizer retuning time does not exceed 5 ms. This receiver uses a module construction based on the standard Compact PCI, with module printed plate sized 100 × 160 mm. The basic receiver type has two
76
3
Radio Receiver Applications for Radio Monitoring System
Table 3.6 Technical specifications of ARK-CT3 DRR Parameter name
Value
General parameters Operating frequency range of the basic set, MHz Input attenuator, dB Maximal permissible input signal, dBm Noise factor, dB: In the range 25–1,000 MHz Above 1,000 MHz Frequency instability of reference oscillator ◦ ◦ Relative error of frequency setting (–20 C to +50 C) Tuning time of the synthesizer, ms, not more than LO phase noise at 10 kHz offset, dB/Hz Selectivity and non-linear distortions Suppression of IF interference, dB, not less than Selectivity on image channel, dB Dynamic range on the 3rd and 2nd order intermodulation, dB Transfer function irregularity in operating frequency range, dB, not more than IF signal Analog IF signal frequency, MHz Bandwidth before IF output, MHz Demodulator frequency setting discreteness, Hz Operating temperature, weight, dimensions, power consumption ◦ Operating temperature interval, C Supply voltage, V Consumed power, VA, not more than Dimensions (width × height × depth), mm Weight, kg
0.009–3,000 10, 20, 30 23 not > 14 not > 16 1 · 10−6 1 · 10−6 5 –95
70 70 75 ±3
41.6 4 or 5 1
–20–+50 24–30 50 450 × 140 × 300 not > 6.5
coherent receiving channels. This receiver’s external view in rack-mount implementation is shown in Fig. 3.36. ARK-CT3 DRR technical specifications for two coherent channels are listed in Table 3.6. ARK-CT3 DRR consists of the range switch and two identical channels of signal receiving (channels A and B), which are connected to a double-channel DSP unit. The structural diagram of ARK-CT3 DRR is shown in Fig. 3.45. The signals from the antennas, or from the antenna switch, pass to the doublechannel range switch, which connects them to the appropriate channels A or B depending on tuning frequency. In the 25 MHz–1 GHz range the input signals are connected to input 1 of the channels, in 1–3 GHz, they are connected to input 2. The receiver control is executed through the RS-485 serial interface. The structural diagram of one receiving path is shown in Fig. 3.46. The following units are included in the channel structure:
ARK-CT3 Digital Receiver
77 Output A IF 41.6 MHz
Input 1
Input B 25–3000 MHz
Range switch
Input A 25–3000 MHz
Channel A
Input 2
Supply 9–16 V
RO LO LO output1 output2 output LO LO RO input1 input 2 input
Output B IF 41.6 MHz
Input 1 Input 2
Channel B
Control RS-485
Fig. 3.45 Structural diagram of ARK-CT3
BPF 3 (285–465 MHz) BPF 4 (160–285 MHz) BPF 5 (85–160 MHz)
Switch 7/1
Switch 1/7
Switch 1/2
Switch 2/1
Attenuator 0..–30dB
Input 1 25–1000 MHz
Buf
BPF 2 (465–700 MHz)
IFA 1 299.2 ÌÃö
Mix
IFA 2 708.8 ÌÃö
Switch 2/1
IFU RFU-1
BPF 1 (700–1000 MHz)
Buf
Mix
Buf
BPF 38.6-44.6 IF output
Buf
Mix
Buf
RO
41.6 MHz VCO 1 257.6
PLL system
Switch 2/1
Buf
Buf
Ðâ LO output 2
Switch 2/1
BPF 6 (45–85 MHz) BPF 7 (25–45 MHz)
RO
Switch 2/1
VCO 2 667.2
PLL system
Buf
Ðâ LO input 2 LO output1
RFU-2
Buf
BPF 3 (2250–2500 MHz)
Buf Buf
BPF 4 (2000–2250 MHz) BPF 5 (1750–2000 MHz)
Switch 8/1
1000–3000 MHz
Switch 1/8
Input 2
Attenuator 0..–30dB
BPF 1 (2750–3000 MHz) BPF 2 (2500–2750 MHz)
Mix
Buf
VCO 3 1131–1400
BPF (1826–2262)
Buf
VCO 2 913–1131
BPF (1550–1826)
BPF 6 (1500–1750 MHz)
VCO 1 733-913
BPF 7 (1250–1500 MHz) BPF 8 (1000–1250 MHz)
BPF (2262–2800) BPF (913–1131)
Switch 3/1 BPF (73311–1913)
Buf
BPF (1131–1400)
Switch 3/1
Synthesiz ers RO unit
Switch 2/1
Switch 2/1
Power supply
RO
PLL system
RO
PLL system
–5V +5V +3.3V+12V+30V
Switch 2/1
PLL system
Control buses and lines Control unit
Switch 1/3
RO Reference oscillator
BUF
CPSU
LO input 1
RO output RO input Control RS-485 Power supply
9 - 16 V
Fig. 3.46 Structural diagram of ARK-CT3 (one receiving channel)
• RFU1 and RFU2 radio frequency units, fulfilling the preliminary frequency selection, signal amplification, and the frequency conversion to the first IF, which is equal to 299.2 MHz or 708.8 MHz (the frequency plan) depending on the frequency tuning • Synthesizer unit (SU) intended for first LO signal formation for RFU1 and RFU2 mixers • IF unit (IFU) fulfilling the main frequency selection, signal amplification and frequency conversion to the second IF, which is equal to 41.6 MHz • Control and power supply unit (CPSU) intended for power supply voltage generation necessary for all units, for receiving and processing the commands from
78
3
Radio Receiver Applications for Radio Monitoring System
the DSP unit, for the unified reference frequency 12.8 MHz generation necessary for synthesizer operation, including in SU and IFU units. The input signal, at frequencies of 25–1,000 MHz, enters the attenuator of RFU1 and then on to the pre-selector assembly. The set of seven sub-range frequency channels is arranged in the pre-selector, which can be selected by seven-channel input and output switches. Each channel has two adjustable band-pass filters controlled by a micro-controller, and the buffer cascade. The signal from the buffer cascade passes to the mixer input. To its second input, the LO signal enters the frequency from 733.8 to 1,299.2 MHz. The input signal at the frequency of 1–3 GHz enters the input attenuator of RFU2 and then on to the pre-selector assembly. The RFU2 pre-selector has eight frequency channels, which can be selected by eight-channel input and output switches. Each channel includes two non-adjustable band-pass filters and the buffer cascade. The signal from the buffer cascade enters the mixer input. The signal from the adjustable LO of the synthesizer unit, at a frequency from 1,550 to 2,800 MHz is applied to the second input of the mixer. The synthesizer unit is intended for reference frequency grid generation for RFU1 and RFU2 mixers. SU consists of three adjustable LO with the adjustable frequencies 73–913 MHz, 913–1,131 MHz, 1,131–1,400 MHz. All local oscillators are implemented on the basis of the frequency synthesizer. The unified reference frequency signal enters the SU from the RO. The LO signal is applied to the RFU1 mixer input, and the signal to the RFU2 mixer is applied after frequency doubling. To arrange the channel’s coherent operation in the direction-finding structure, the possibility is ensured for both to pass the internal LO signals to the synchronization output connectors and to obtain the external oscillator signals from the input connectors. The intermediate frequency unit (IFU) consists of two channels from the first IF processing: the synthesizers and the switches of input and output data. The input switches fulfills the IF signals switching from RFU1 and RFU2 outputs to one of the first IF processing channels (IFA1 or IFA2). The first IF and first LO frequency values are shown in Table 3.7, for various tuning frequencies. Table 3.7 First IF and first LO frequencies of ARK-CT3 Tuning frequency, MHz
LO frequency, MHz
IF frequency, MHz
from 25 to 464.8 from 464.8 to 1,001.2 from 1,001.2 to 1,250.8 from 1,250.8 to 2,550.4 from 2,550.4 to 2,750 from 2,750 to 3,000
from 733.8 to 1,173.6 from 767.4 to 1,300.4 from 1,710 to 1,959.6 from 1,550 to 2,799.6 from 2,201.2 to 2,450.8 from 2,041.2 to 2,291.2
708.8 299.2 708.8 299.2 299.2 708.8
IFA1 and IFA2 have similar structures and consist of the first IF filter and buffer amplifier, and the second mixer. The signals from the buffer amplifiers
ARK-KNV4 External Remote-Controlled Converter
79
are applied to the mixer inputs. Another input of the mixers absorbs the signals from the appropriate non-adjustable LOs of IFU synthesizers module (257.6 and 667.2 MHz). Similarly to SU in IFU, to arrange the coherent channel operation in the direction-finding equipment structure, it is possible both to pass the internal LO synchronization signals to the output connectors and to obtain the external source signals at the input connectors, for further application as the local oscillator signals. The control and power supply unit (CPSU) consists of the impulse voltage converter generating the voltage set necessary for other units operation; the receiving and processing micro-controller for the commands from the DSP unit, and the reference oscillator generating the unified reference frequency 12.8 MHz for the frequency synthesizers included in the SU and IFU structures. To ensure that both channels operate from the unified reference frequency there is the possibility in CPSU both to pass the reference oscillator synchronization signal to the output connector and to obtain the reference oscillator signal from the external source (another channel) from the input connectors.
ARK-KNV4 External Remote-Controlled Converter The ARK-KNV4 radio signals converter operates in the structure of the wide-band automated RM and CEE systems. It is intended for radio signal transition from the range of 1–18 GHz to the intermediate frequencies 299.2 and 708.8 MHz. This converter can be used as portable equipment, or as a stationary device, mounted on a mast or a tripod. The converter connects to ARK-CT1, ARK-CT2 or ARK-PR5 DRR. The main technical specifications of the converter are listed in Table 3.8. The operating temperature range of the ARK-KNV4 converter is from –10◦ C to +50◦ C;. Relative humidity of the environment should not exceed 80%. The external view of the ARK-KNV4 converter is shown in Fig. 3.37, and Figs. 3.47 and 3.48 show the converter mounted on a tripod and on a dielectric mast, together with the horn measuring antenna. The structural diagram of the ARK-KNV4 converter is shown in Fig. 3.49. This converter consists of the active antenna unit, pre-selector and first IF unit, synthesizer 1, mixer 1, synthesizer 2, the second IF unit, the reference oscillator and the control unit. The built-in antenna system of the ARK-KNV4 contains eight wide-band active antennas implemented as the active phased array with linear signal polarization. The signal from the antenna system, which corresponds to the switch tuning frequency, and which, with the help of switches “9” to “1” passes through the group path to the mixer. The ninth switch input is used for the transition to operation mode from the external antenna. The signal transfer factor from the external antenna input to IF output is 0 dB. Thus, the ARK-KNV4 converter, together with the measuring antenna, can be used for the field strength measurement.
80
3
Radio Receiver Applications for Radio Monitoring System
Table 3.8 Main parameters of ARK-KNV4 radio signals converter Parameter name Received frequencies range, GHz: At operation from the internal antenna At operation from the external antenna input Output signal frequency, MHz Relative LO frequency instability, not worse than –3 dB-level bandwidth, MHz, not less than Converter sensitivity across the field, at operation from the internal antenna in the bandwidth 10 kHz, μV/m, not worse than Transfer factor at operation from external antenna, dB Transfer factor error at operation from external antenna, dB, not > Relative level of 3rd-order intermodulation noise at the 10 MHz offset for the receiver 120 kHz bandwidth, dB, not more than Spurious receiving channels suppression at operation from internal antenna, dB, not less than Noise factor (without accounting for the transfer factor of antenna circuit), dB, not worse than Medium level of inherent noise within the bandwidth 120 kHz, W, not more than Dynamic range, dB Through AFR irregularity, dB, not more than Phase noise at 10 kHz-offset, dB/Hz, not more than Intercept point on the 3rd-order intermodulation, dBm SWR on external antenna input at input impedance 50 , not worse than SWR on output, not worse than Supply voltage, V Consumed power, VA, not more than Dimensions (width × height × depth), mm, not more than Converter weight, kg, not more than
Value
3–18 1–18 299 or 708.6 2.10–7 10 50 0 6 64 45 14 4.10–11 70 ±6 –80 –5 2 3 12±3.0 or 27±3.0 19 250 × 220 × 90 2.0
From the mixer output the signal passes through the first IF unit. Here, the signal is amplified and restricted in bandwidth. After that, the first IF signal enters the second IF unit, which carries out the additional frequency selection, the amplification, and the signal transfer to the output frequency of the second IF. Synthesizers 1 and 2 generate the signals required for the mixers. After filtering, amplification, and frequency conversion, the signals from ARKKNV4 output at IF 299.2 MHz or 708.8 MHz enter the receiver input, for example, ARK-D1TP or ARGAMAK. The intermediate frequency value is defined by the ARK-KNV4 tuning frequency, in accordance with the frequency plan accepted at the development stage. The control unit accepts the commands from the external devices through the serial bus RS-485, transfers the data to the active antenna unit, to the synthesizer units and to the second IF unit. The control unit of ARK-KNV4 has a nonvolatile storage device, in which the adjustable parameters are stored, namely, the frequency correction coefficients for the reference oscillator, attenuator adjustment data, etc.
ARK-PR5 “Argamak” Digital Radio Receiver
81
Fig. 3.47 ARK-KNV4 converter on the tripod
These parameters are automatically loaded after switching-on the converter or after the hardware reset signal is received. The ARK-KNV4 external remote-controlled radio signal converter is included into the measuring equipment register, by State Standard of the Russian Federation.
ARK-PR5 “Argamak” Digital Radio Receiver Compared to ARK-CT1 and ARK-CT3, the ARK-PR5 ARGAMAK digital radio receiver has several times as less weight and dimensions. Therefore, its accuracy and sensitivity parameters are better than those of the ARK-CT3, and the synthesizer tuning time on frequency is shorter than 2 ms. This receiver consists of two modules: the ARK-PS5 RF-VHF-UHF radio signal converter and the ARK-CO DSP module, as shown in Fig. 3.50. Each module is located on the multi-layer printed board with dimensions of 100 × 160 mm. The implementation of each module allows for the possibility of
82
3
Radio Receiver Applications for Radio Monitoring System
Fig. 3.48 ARK-KNV4 converter on the dielectric mast
ARK-KNB4 IF1 unit Active antenna unit First IF unit
Active antenna 1
Second IF unit
IF output
Active antenna 4 Active antenna 5
Switch 9 to 1
Active antenna 3
Group section
Active antenna 2 Mixer
Synthesizer 2
Reference oscillator
Synthesizer 1
Active antenna 6 Control unit
Control RS-485
Active antenna 7 Active antenna 8
Fig. 3.49 Structural diagram of ARK-KNV4 converter
their joint or separate application in the equipment. The important DRR feature is the possibility of its application – without any revisions – in the structure of multichannel coherent systems. Moreover, DRR has stable characteristics, including an interface to connect the external reference oscillator, which allows it to be used
ARK-PR5 “Argamak” Digital Radio Receiver
83
IF Input
ARK-PS5
ARK-CO
USB 2.0
PC
(RS485) Control
Fig. 3.50 Enlarged structural diagram of ARGAMAK SRR
as measuring equipment. One more helpful feature is the possibility to select the receiver bandwidth from 2, 5, or 10 MHz. ARK-CO2, ARK-CO5, and ARK-CO10 DSP modules with bandwidths 2, 5, and 10 MHz, respectively, can be used together with this receiver, depending on the bandwidth. Depending on the implementation version, the ARGAMAK DRR can be accommodated in a separate case, as shown in Fig. 3.51, inside the protective case shown in Fig. 3.52, or can be built directly into the equipment, e.g., into the antenna array of the ARK-MK11 DF system, or into the case of the ARK-RP4 handheld direction finder, etc.
a)
b) Fig. 3.51 ARK-PR5 DRR: (a) Front view; (b) Rear view
84
3
Radio Receiver Applications for Radio Monitoring System
Fig. 3.52 ARK-PR5 DRR in protective case with emergency battery
The ARGAMAK DRR has analog IF outputs with bandwidths of 2, 5, or 10 MHz, for connection to the various decoding or demodulating units. ARK-PR5 DRR specifications are listed in Table 3.9. The ARK-CO DSP module has an audio line and adjustable outputs from the digital signal demodulator. The receiving of commands and the data transfer to the PC are executed via a USB 2.0 serial interface. The calculated spectral panoramas or the time signal samples are transferred to the PC. The rate of spectral panoramas obtained depends on the receiver, and DSP unit bandwidths and may exceed 3,000 MHz/s, at a bandwidth of 10 MHz (see Table 3.9). The continuous time sample of arbitrary duration can be transferred to the PC for the received signal with a bandwidth up to 2 MHz. For 5 and 10 MHz signal bandwidths, time samples of limited duration are transferred. Achievement of the DRR’s low weight and dimensions became possible due to the application of advanced passive electronic components with 0402, 0603, and 0805 standard size, and the integrated circuits in MLP, QFN leadless cases, which allow a surface mounting density of more than 10 elements per square cm to be achieved. The ARK-PR5 receiver plate, with dimensions 100 × 160 mm, has about
ARK-PR5 “Argamak” Digital Radio Receiver
85
Table 3.9 ARK-PR5 DRR technical specifications Parameter name General parameters Operating frequency range of basic set, MHz Input attenuator, dB Maximal permissible input signal, dBm Noise factor, dB: In range 25–1,000 MHz Above 1,000 MHz Frequency instability of reference oscillator ◦ ◦ Relative error of frequency setting (–20 C–+50 C) Tuning time of synthesizer, ms, not more than LO phase noise at 10-kHz-offset, dB/Hz In range 25–1,000 MHz Above 1,000 MHz Selectivity and non-linear distortions Intercept point on the 3rd-order intermodulation IP3, dBm IF interference suppression, dB, not less than Image attenuation, dB Dynamic range on the 3rd and 2nd order intermodulation, dB Transfer factor irregularity within operating frequency range, dB, not more than IF signal IF analog signal frequency, MHz Bandwidth before IF output, MHz DSP unit Demodulator frequency setting discreteness, Hz Built-in demodulator Detectors for signal level measurement Spectrum calculation speed at usage of, MHz/s: ARK-CO2 DSP module ARK-CO5 DSP module ARK-C010 DSP module Outputs, control and data transfer interface Low frequency audio output Audio output for earphones Control and data transfer interface Operating temperature, weight, dimensions, power consumption Operating temperature interval, ◦ C Power supply voltage, V Consumed power, VA Dimensions (width × height × depth), mm Weight, kg, max
Value 0.009–3,000 0–30, step 2 23 not more than 12 not more than 14 5 × 10–7 5 × 10–7 2 –95 –85 0 70 70 75 ±3 10.7 or 41.6 2, 5, or 10 1 AM, FM, PM, SSB, CW Peak, Quasi-peak, rms 600 1,500 3,000
Is present Is present USB 2.0 –20–+50 9–16 15 108 × 42 ×200 1.5
86
3
Radio Receiver Applications for Radio Monitoring System
1,800 mounted elements. In printed plate topology, special reference marks are provided for automated assembling; and, checking points are used for the automated adjustment. To manufacture the multi-layer printed circuit board, the FR-4 and RO4350 materials were used, which allows for the ensuring of parameter recurrence over specimens, to take these parameters into account in the mathematical models. At each of the design stages, mathematical modeling was widely used, permitting the receiver unit characteristics to be balanced, and the best location of the printed-circuit wiring components to be selected, without the manufacture of a preproduction model. Figure 3.53 depicts the external view of the ARK-PS5 HF-VHF-UHF radio signal converter module for the ARGAMAK DRR, without the shields, and Fig. 3.54 shows its structural diagram.
Fig. 3.53 External View of ARK-PS5 Radio Signal Converter Module
The ARK-PS5 module represents a device with dimensions 100 × 160 × 20 mm, based on the single, printed circuit board in “Euroboard” standard. The purpose of this module is to receive radio signals in the frequency range of 9 kHz–3 GHz. The output signal is an IF signal at frequency 10.7 MHz or 41.6 MHz. IF value selection for each of the two outputs is executed by the software. For reception in the UHF range of 25–3,000 MHz at the output IF 41.6 MHz, two frequency conversions are used, at the output IF 10.7 MHz, three frequency conversions are used. A single frequency conversion is used for reception in the range of 0.009–30 MHz at output IF 41.6 MHz. But, this frequency conversion should be doubled for output IF 10.7 MHz. The signal at output IF arrives at two independent output connectors and for each connector an individual IF value can be assigned. To arrange the synchronous operation of the LOs in the structure of several ARKPR6, four connectors are mounted at its printed circuit board. Using them, one can ensure the following synchronization modes: • Completely autonomous operation from the internal reference oscillator at 12.8 MHz frequency with 5 · 10−7 accuracy • Autonomous operation of frequency synthesizers from the external reference oscillator, the signal of which enters one of the four synchronization connec-
IF1 In
BPF 4.4 (225–465 MHz) BPF 4.5 (110–225 MHz)
VCO 1.2 1700-2400
BPF 4.6 (53–110 MHz)
Channel  706.8 MHZ
Frequency conversion unit
BA
VCO 2.1 258.4
Mixer 1
BPF 4.7 (25–53 MHz) Frequency conversion unit VCO 3 44.1–71.6
Switch 2/1
BA
Filter unit IF1 Channel  300 MHZ
BA LO2 out/in
RO PLL system
BA
VCO 2.2 665.2
Channel 0.009-30 MHz
BA
Switch 2/1
BPF 4.3 (465–850 MHz)
Switch7/1
Switch 1/8
Att 0…–30 (0.009-30 MHz) dB
25-3000 MHz
PLL system
Switch 2/1
RO
BPF 4.2 (0.85–1.6 GHz)
BA
Switch 3/1
BPF 4.1 (1.6–3.0 GHz)
Switch 1/2
VCO1.1 730–1300
Channel 25–3000 MHz
Switch 2/1
Frequency conversion unit
Switch 2/1
Preselector unit
BA
RO PLL system
Switch 4/1
BPF 4.11 (9–1500 kHZ)
LO3 out/in
–3V 9–16 V
+15V +5V +3.3V +25V
Power supply unit Control buses and lines
Control RS–485
BPF 41.6 (5 MHZ
Filter unit IF1
Frequency conversion unit
BA
Mixer 4
RO +5 V RO
BPF 10.7 (2 MHz)
PLL system BA
Control unit Reference oscillator
IF outputs switch
Switch 2/1
BPF 4.10 (1.5–4.5 MHz)
Mixer 2 Mixer 3
Switch 2/1
BPF 4.9 (4.5–12 MHz)
Switch 2/1
Switch 1/4
0.009-30 MHz Àtt 0…–30 dB
Switch 2/1
BPF 4.8 (12–30 MHz)
ARK-PR5 “Argamak” Digital Radio Receiver
LO1 out/in
IF 41.6/10.7MHz
LO4 out/in. RO Switch 2/1
VCO 4 52.3
87
Fig. 3.54 Structural diagram of ARK-PS5 of ARGAMAK DRR HF-VHF-UHF radio signal converter
88
3
Radio Receiver Applications for Radio Monitoring System
tors to increase the frequency setting accuracy, and for the complex frequency synchronization of several ARK-PS5 modules • Autonomous operation from the internal reference oscillator with reference frequency signal delivery to one of the four synchronization connectors for the complex frequency synchronization of several ARK-PS5 modules • Synthesizer operation from the internal reference oscillator with LOs signal delivery to four external connectors, to ensure the frequency synchronization of several ARK-PS5 modules, up to accuracy within the phase (e.g., operation in the direction finder’s complex structure) • The mode of completely-driven LOs with disconnected internal reference oscillators and frequency synthesizers, to ensure the synchronization of several ARK-PS5 modules, up to accuracy within the phase (e.g., operation in the direction finder system’s structure). The ARK-PS5 module represents a device with dimensions 100 × 160 × 20 mm, based on the single, printed circuit board in “Euroboard” standard. The purpose of this module is to receive radio signals in the frequency range of 9 kHz–3 GHz. The output signal is an IF signal at frequency 10.7 MHz or 41.6 MHz. IF value choice for each of the two outputs is executed by the software. For reception in the UHF range of 25–3,000 MHz at the output IF 41.6 MHz, two frequency conversions are used, and, at the output IF 10.7 MHz, three frequency conversions are used. A single frequency conversion is used for reception in the range of 0.009– 30 MHz at output IF 41.6 MHz. But, this frequency should be doubled for output IF 10.7 MHz. The signal at output IF enters two independent output connectors and, for each connector, an individual IF value can be assigned. To arrange the synchronous operation of the LOs in the structure of several ARKPR6, four connectors are mounted at its printed circuit board. Using them, one can ensure the following synchronization modes: • Completely autonomous operation from the internal reference oscillator at 12.8 MHz frequency with 5 · 10−7 accuracy • Autonomous operation of frequency synthesizers from the external reference oscillator, the signal of which enters one of the four synchronization connectors to increase the frequency setting accuracy, and for the complex frequency synchronization of several ARK-PS5 modules • Autonomous operation from the internal reference oscillator with reference frequency signal delivery to one of the four synchronization connectors for the complex frequency synchronization of several ARK-PS5 modules • Synthesizer operation from the internal reference oscillator with LOs signal delivery to four external connectors, to ensure the frequency synchronization of several ARK-PS5 modules, up to accuracy within the phase (e.g., operation in the direction finder’s complex structure) • The mode of completely-driven LOs with disconnected internal reference oscillators and frequency synthesizers, to ensure the synchronization of several ARK-
ARK-PR5 “Argamak” Digital Radio Receiver
89
PS5 modules, up to accuracy within the phase (e.g., operation in the direction finder’s complex structure). In all of the above-mentioned modes, the pre-selector and other unit tuning is independently executed through the control bus for each ARK-PS5 module, in accordance with the commands from the controlling PC or another device. Let us consider the structural diagram of ARK-PS5 ARGAMAK DRR module in Fig. 3.54. Upon input of the HF, VHF, and UHF pre-selector sections, the attenuator units are applied; they are intended for the input signal attenuation within the limits 0–30 dB with 2 dB-step. The pre-selector section units of 25–3,000 MHz are used for the preliminary selection, to suppress the signals in the spurious receiving channels and to improve the signal-noise ratio (SNR) at the first mixer input. The input signal arrives through the input switch to the inputs of the adjustable filters, which select the input signal in the given frequency ranges. There are seven adjustable wide-range filters (BPF1–BPF7) switching by the input and output switches, for the following frequency sub-ranges: 25–53; 53–110; 110–225; 225–465; 465–850; 850–1,600; 1,600–3,000 MHz. Within these frequency sub-ranges, each filter is adjusted by the variable capacitance diodes, which are controlled by DAC. Pre-selector units of 0.009–30 MHz consist of four band-pass filters for the frequency sub-ranges 9–1,500 kHz, 1,500–4,500 kHz, 4.5–12 MHz, 12–30 MHz. At that point, the lowest sub-range filter is non-adjustable, but three other filters are adjustable with the bandwidths of 1 and 2 MHz. Pre-selector DAC and switches are controlled by the micro-processor control unit. The individual pre-selectors adjustments for each module are stored in a nonvolatile storage device, which allows for the achievement of the small transfer factor irregularity – both within the bandwidth and within the full operating range. The frequency converter unit is used for the radio signal frequencies transfer to the intermediate frequencies, for its amplification and image channels suppression, and consists of three frequency conversion units and IF filters. The signal from the VHF-UHF range pre-selector unit enters the frequency conversion input 1, which converts the input signal to the intermediate frequencies 300 and 706.8 MHz, in accordance with the frequency plan. The signal enters the first mixer (M1) input, and the signal from the adjustable LO is applied to another mixer input. The LO signal is applied through the switch, from one of three sources: from external LO or from one of two internal oscillators: VCO1.1 or VCO1.2, selected in accordance with the frequency plan. As a result of the signal conversion, the signal of the first IF1 with the frequencies 300 MHz or 706.8 MHz is generated at mixer output, which arrives further to the IF1 filter unit. The values of the first intermediate frequency IF1 and the adjustable LO frequency of the frequency conversion unit 1 are listed in Table 3.10, for various receiver frequencies. From the filter unit IF1 output, the signal passes to frequency conversion unit 2. This unit converts the signal from the first IF to the IF2 signal at 41.6 MHz. The
90
3
Radio Receiver Applications for Radio Monitoring System
Table 3.10 First IF and LO Frequency Values Tuning frequency, MHz
LO frequency, MHz
IF frequency, MHz
from 25 to 464.8 from 464.9 to 1,000 from 1,000.1 to 1,599.1 from 1,599.2 to 2,600.6 from 2,600.7 to 3,000
from 731.8 to 1,171.8 from 764.9 to 1,300 from 1,706.9 to 2,305.9 from 1,899.2 to 2,300.6 from 1,893.9 to 2,293.2
706.8 300 706.8 300 706.8
signal from filter unit IF1 enters the second mixer input, and the signal from one of the LOs, selected in accordance with the frequency plan, is sent to the second input of this mixer. The IF2 signal at 41.6 MHz, from the mixer output, enters the IF2 filter unit and arrives further to the frequency converter unit 4 and to the IF outputs switch. The signal from the HF range pre-selector passes to the frequency converter unit 3 where it converts into IF 41.6 MHz. The frequency converter unit 4 converts the second IF signal into the third IF 10.7 MHz. The output IF signal enters two independent output connectors, and, for each, an IF value of 41.6 MHz or 10.7 MHz can be assigned by software. All LOs of the frequency converter unit are made on the basis of the frequency synthesizers. The unified reference signal frequency of 12.8 MHz, applied to the synthesizers, is generated by the temperature-controlled internal reference oscillator with relative instability 5 · 10−7 or by the external reference oscillator at frequency 12.8 MHz. The control unit provides the receiving command and controls the voltage and commands the transfer into the receiver units. The control unit consists of the microprocessor-controlled (MPC) unit, along with two DACs. The commands from an external source, e.g., from the ARK-CO module, arrive at the control unit via the serial bus of RS-485 protocol. DAC modules generate the control voltages for pre-selectors and the crystal reference oscillator. The power supply unit for the receiver converts the DC input voltage 9–16 V into a set of DC output voltages (–3 V; +5 V; +3.3 V; +15 V, +25 V), required for the unit’s power supply. The structural diagram of the power supply unit is shown in Fig. 3.55. The power supply unit consists of the secondary power supply (SPS) units, the input and output power supply filters, which reduce the output voltage rippling, and the voltage regulator, which provides the power supply on duty for the microprocessor. Let us consider the second module included in ARGAMAK DRR, namely, the DSP module. The structural diagram of the ARK-CO DSP module is shown in Fig. 3.56, and a photo of the printed circuit board with the mounted components is shown in Fig. 3.57. In this photo, we can see the ARK-CO module without the shields. After adjustment, its modules are exposed to the careful shielding. In the CO unit, the analog filters are provided, allowing unambiguously representation of the signal in digital form. Signal demodulation for an acoustical
ARK-PR5 “Argamak” Digital Radio Receiver
91 Power supply unit
External power supply
Power supply filter
9–16 V
Secondary power supply module 1
Power supply filter
–3 V
Power supply filter
+15 V
Power supply filter
+25 V
Power supply filter
+5 V
In/Out PS +5 V
Secondary power supply module 2
Voltage regulator Power supply filter
+3.3 V
Fig. 3.55 Structural diagram of power supply unit
check-up is executed by soft-hardware means that allow the number of permissible modulation types to increase without hardware modernization, and allow the weight and dimension parameters to change. The possibility of radio signal recording in vector form exists, for further technical analysis. This module provides simultaneous operation in the panoramic analysis and signal demodulation modes. The ARK-CO module has two channels that allow for its coherent signal processing. The module provides high performance when direction finding, radio signal demodulation, and when recording in vector form. It serves as the connecting link Analogous output Input filters
1 Channel1In 1 Channel 2In
2 Channel1In 2 Channel 2In
S w i t c h
S w i t c h
Band-pass filter 10.7±1 41.6±2.5 41.6±1 41.6±5 MHz Band-pass filter 10.7±1 41.6±2.5 41.6±1 41.6±5 MHz
Frequency converter 12.3–10.7 41.6–38.4 41.6–40.0 41.6–35.2 MHz Frequency converter 12.3–10.7 41.6–38.4 41.6–40.0 41.6–35.2 MHz
Analogous-digital unit Band-pass filter 1.6±1 3.2±2.5 1.6±1 6.4±5 MHz Band-pass filter 1.6±1 3.2±2.5 1.6±1 6.4±5 MHz
Signal processor
Interface
PLD
Signal processor
PLD
Digital receiver
Signal processor
Codec
A D C
RS485 Nº1 RS485 Nº2
A D C
fdig
fïp
USB
Digital receiver
I2C
Synthesizer unit f ref1 f ref2 12.8 MHz
9–16 V
Switch
Switch
Synthesizer
Reference oscillator
Synthesizer
Reference oscillator
Buffer
Power supply unit
+5 V –5 V +3.3 V +2.5 V
Fig. 3.56 Structural diagram of ARK-CO5 DSP unit
I2C Control processor with ROM
Audio unit
Control
Amplifier
Control
Buffer
Line output Head-phones
92
3
Radio Receiver Applications for Radio Monitoring System
Fig. 3.57 External view of ARK-CO5 DSP module
in RM complex control systems, ensuring the information exchange between PC or other controlling device, DSP processors, the radio signal converters and the additional equipment. Similar to the ARK-PS5 receiver, the ARK-CO unit is implemented as a module with dimensions 100 × 60 × 20 mm. An IF signal at frequency of 41.6 MHz or 10.7 MHz enters it. At IF 41.6 MHz, the unit provides analog signal processing with a bandwidth of 5 MHz or 10 MHz, but, at IF 10.7 MHz, it processes the signal with a 2 MHz bandwidth. This unit executes input signal filtering, its conversion into digital form, further signal processing, e.g., demodulation or spectrum calculation, and the preliminary amplification of the output audio signal. Moreover, the unit provides external device control via RS-485 serial interface, e.g., the ARK-PS5 receiving section. The module has two processing channels; each channel has two switched inputs. There are also two analog input-outputs and the digital differential input-output at 12.8 MHz frequency, to synchronize the operation with other devices. The module can operate in autonomous mode or can be controlled by the external PC through a USB2.0 bus. Moreover, the module has two serial ports of RS-485 standard for communication or control by the other devices. As we can see from the structural diagram, the ARK-CO module consists of three digital signal processors of Analog Devices-type AD2185 for the digital processing of the received signals and two controlling processors with Intel 8051 core to control the audio signals and the data exchange between the PC, the digital processors, and the ARK-PS5 module or other devices. The selection of the mentioned digital signal processors can be explained by the requirement to reduce the consumed power and the interference level created by these electronic devices. The presence of the built-in reference clock-frequency synthesizer can be considered a module feature that provides the possibility to connect the external high-stable reference oscillator, as well as the possibility to improve the signal AD conversion stability and hence, to increase the signal parameter measurement accuracy, e.g., its carrier frequency.
Conclusion
93
The DRR AD6620 – one for each channel – fulfils the demodulation, or filtering operation, and the signal decimation, to execute the vector analysis. The ARK-CO module can operate in three main modes: in autonomous mode from the remote control panel, under PC control through USB interface, and through RS-485 interface. The last mode can be used to create the net of complexes controlling through radio-modems.
ARGAMAK-I Panoramic Measuring Receiver Based upon the ARGAMAK DRR, the ARGAMAK-I panoramic measuring receiver was developed, and was certified as measuring equipment. The features of this receiver are the specifications required for measuring equipment. These specifications are ensured by the minimization of sensitiveness functions to parameter variations during the design phase, and also by using the digital technologies of path corrections. So, the first adjustable IF introduced allows us to virtually exclude the number of staggering frequencies at the panoramic analysis. The stability of the absolute section transfer function is achieved by means of its digital correction, taking into account the operating temperature. The application of the temperaturecontrolled reference oscillator provides reference frequency stability upon temperature change; and, the ability to adjust the frequency – with the help of specific software – eliminates the aging effect of the crystal resonator.
Conclusion In the present chapter, our main attention is placed on structural diagrams, the construction features, and the parameters of superheterodyne receivers with single or several frequency conversions. This receiver type is the most convenient for radio monitoring problems in the wide frequency range from units to thousands of megahertz and for digital signal processing at intermediate frequencies. The effectiveness of radio receiver applications in radio monitoring systems is defined by its main parameters: operating frequency range, amplitude-frequency response, standing-wave ratio at receiver input, selectivity on adjacent and spurious receiving channels, receiver sensitivity in the given bandwidth, linearity, phase noise of the frequency synthesizer, adjustment (re-tuning) speed, bandwidth, time of spectrum calculation, weight and dimension, and the exploitation complexity. In this chapter, we show that noise factor is, at present, the most universal parameter for RR sensitivity determination in the given frequency band, for estimation of the 2nd and 3rd order intermodulation intercept points, which are the most suitable for determining receiver linearity. Thus, knowing the noise factor and the intermodulation intercept points, one can easily find out the dynamic range of the receiver. The various RRs in use now for radio monitoring applications are discussed. These include the scanning receiver, the selective micro-voltmeter, the spectrum analyzer, the panoramic receiver, and the panoramic measuring receiver. We prove
94
3
Radio Receiver Applications for Radio Monitoring System
that the digital panoramic measuring receiver is the most suitable for radio monitoring applications. Finally, a short developmental history of hardware-software RM systems in the Russian Federation during last 20 years is considered in this chapter. The structural diagrams and the construction features of the Russian ARK-CT1, ARK-CT3, ARKPR5 panoramic digital receivers, and ARK-KNV4 converter, which are widely used in radio monitoring, measuring, and direction–finding systems are described.
References 1. Davies, J., and Carr, J.J., Newnes Radio and RF Engineer’s Pocket Book. 2nd Edition, Butterworth-Heinemann, 2000, 594 pp. 2. Baskakov, S.I., Radio Engineering Circuits and Systems (in Russian). Moscow, Vyschaya Skola, 2005. 462 p. 3. Kurganov, A.N., and Pavlyuk, A.P., Linearity Features of Measuring Radio Receivers (in Russian). Trudy NIIR, 2003. 4. Raucher, Ch., Janssen, V., and Minihold, R., Fundamentals of Spectrum Analysis. Munchen, Rohde & Schwarz, 2001. 5. Red, T.E., Arbeitsbuch fur den HF-Techniker. München, Franzis-Verlag GmbH, 1986, ISBN 3-7723-8151-0. 6. Poberezhskiy, E.S., Digital Radio Receivers (in Russian). Moscow, Radio i sviaz, 1987, 184 pp. 7. Rabiner, L.R., and Gold, B., Theory and Applications of Digital Signal Processing. Prentice Hall, New Jersey, 1975. 8. Digital Radio Receiving Systems: Reference Book. Zhodzizhskiy, M.I., Mazepa, R.B., Ovsiannikov, E.P. etc. (in Russian). Under edition of Zhodzizhskiy, M.I. Moscow, Radio i sviaz, 1990, 208 pp. 9. Rembovsky, A.M., Automated Radio Emission Monitoring – Problems and Facilities (in Russian). Special technologies. 2002. Special Edition. pp. 2–6. 10. Sergeev, V.B., Sergienko, A.R., and Pereversev. S.B., ARK-D1TP Panoramic Measuring Receiver. Special technologies. No. 3, 2004, pp. 50–57.
Chapter 4
Single-Channel and Multi-Channel Radio Signal Detection
Introduction Complexity of radio monitoring-problem solutions, as applied to the various types of signals, can differ considerably. Thus, for example, the detection of wide-band signals in the absence of a priori information is a serious problem. At the same time, narrow-band signal processing, which, as a rule, is described by a large SNR, can be solved successfully on the basis of a rather simple model of examined processes requiring a minimum of a priori information. Below, we consider the digitalprocessing algorithms for the narrow-band radio signal group, observed together on the wide-band additive noise background [1]. Let us assume the following conditions for radio environment investigation: • Analyzed random process has an unknown number of narrow-band components, generated by RES • A priori information on the carrier frequency, and on the type and parameters of the examined signal modulation, are absent. We know only that the signal spectrum width does not exceed some limit set in advance • Noise intensity data are absent, however, within the processed frequency bandwidth, restricted by the possible sample frequency values, the noise power does not change practically. Further, in this chapter, we shall interpret the signal detection task as the determination of the number of simultaneously-observed narrow-band signals and their location along the frequency axis. Let us formulate this task as follows. An unknown number M of narrow-band radio signals um (t), whose spectrum width dfm is limited and cannot be more than some definite value dfrch (radio channel width), acts within the frequency band F on the additive normal white noise ξ (t) background with unknown intensity σξ2 · The observed process can be written in the form: uin (t) = ξ (t) +
M
um (t, fm , dfm )
(4.1)
m=1
A. Rembovsky et al., Radio Monitoring, Lecture Notes in Electrical Engineering 43, C Springer Science+Business Media, LLC 2009 DOI 10.1007/978-0-387-98100-0_4,
95
96
4
Sin (f)
df1
Single-Channel and Multi-Channel Radio Signal Detection dfM
df2
dfrch
f1
f2
F
fì
f
Fig. 4.1 Spectral power density of observed process
where f m is the central frequency of the um (t) signal spectrum. The signals do not overlap in requency, and are equally likely to be located in any part of the analyzed range. Their spectrum form is unknown (see, e.g., Fig. 4.1). It is required to determine the number M of the narrow-band signals included in the uin (t) structure, to estimate their spectrum width and central frequencies on the basis of observation results of one or several process (4.1) samples. As the theoretical basis of the stated-problem solution, we use the statistical synthesis theory of joint optimal algorithms [2]. In terms of this theory, the above-stated problem may be formulated as the problem of compound vector → → T λ = σξ2 , M, λ M estimation, where M is the signal number, falling into the ana M is the vector of the unknown parameters collection of the lyzed frequency band, λ signals being resolved. Vector x samples of the observed co-ordinates of the input random process, calculated by the registered realization uin (t), are the basis for vec determination. The probabilistic features of x samples depend on the true value tor λ M ). λ, and redefined by the conditional probability density W( x|λ) = WM ( x|σ 2 ,M,λ ξ of sigThe search object is such a detection rule at which the received estimation λ nal number and parameters is the best of all those possible, on the basis of some criterion selected in advance. Independent from the selected optimization criterion, the solution of the stated problem is related to the research of the behavior of the above-mentioned condi called the likelihood functions. At the selection tional probability densities W( x|λ), of the time sample collection of uin (t) process, each value included in x depends in a complex way on parameters of all resolved signals, and determination of the estimation is rather time-consuming. On the contrary, in the freoptimal vector λ quency domain, the dependence of the spectral sample probability characteristics on the resolved signal parameters is local, and the samples themselves can be considered as independent [3], which leads to the likelihood function factorization, and simplifies both the optimal algorithm search process and the obtained processing
Single-Channel Signal Detection
97
procedure itself. Due to this, we shall use the spectral samples of the uin (t) process as a vector of observed co-ordinates x.
Single-Channel Signal Detection Discrete Fourier transform (DFT) N−1 1 c˙ (n) = uin (kT)e−i2πnk/N N
(4.2)
k=0
is the generally-accepted spectral characteristic of signals represented by the fixed volume samples. However, the complex sample c˙ (n) collection is not the optimal vector of the M , included in the W( x|λ) observed co-ordinates x because the parameter vector λ structure, contains the unknown amplitudes and phases of the spectral components of all signals. At likelihood function maximization, with respect to these parameters (in the absence of a priori information about the phase spectrum of the detected signals), the ratio of the real and imagined parts of c˙ (n) will affect on only the noninformative vector of the phase spectral components for the solved problem. The signal set detection results depend on the amplitude vector of the analyzed process components only, therefore, one can take the sample collection 1 |˙c(n)|2 R R
XR (n) =
(4.3)
r=1
of the process uin (t) energy spectrum as the vector x. If the spectrum (4.3) is obtained by R samples of observed process uin (kT), each N samples in volume, the signal um (t) can be presented in this spectrum dnm = N · T · dfm
(4.4)
by the spectral samples, beginning from the sample which is more right than nm = int[(fm − 0.5dfm )NT], where int[.] is the function of the integer part of the number-taking. The spectral sample sub-set, which satisfies the inequality nm ≤ n ≤ nm + dnm and characterizes the m−th signal, we designate, in the future, as θm . In fact, these samples represent the independent random variables following the non-central χ 2 - distribution: Wncχ 2 (x; a, λ, J, δ) = x−a J/2−1 −[δ+(x−a)/λ] J x−a 1 e = λJ/2 (J/2) 0 F1 2 , δ λ λ where 0 F1 (α,z) is the generalized hyper-geometric function. These distribution parameters are:
(4.5)
98
4
Single-Channel and Multi-Channel Radio Signal Detection
a = 0, λ =
σξ2 RN
, J = 2R, δ =
2 RNSmq
4σξ2
,
(4.6)
where σξ2 is the additive noise power; q = n − nm is the sample sequence number within the useful signal spectrum; and, Smq is the amplitude of the signal um (t) component at the appropriate frequency. The noise samples of the averaged energy spectrum (4.3) represent the random variables, following the central χ 2 −distribution 1 Wχ 2 (x;a,λ,J) = λ(J/2)
x−a λ
J/2−1
e−(x−a)/λ
(4.7)
with J = 2R degrees of freedom and parameters a = 0, λ =
σξ2 RN
.
(4.8)
The corresponding samples sub-set we shall designate as θn . Taking into account the selected observation vector x, we can formulate the detection problem as follows. There exists an averaged energy spectrum (4.3) of the random process, which consists of noise, and may be of several useful narrow-band signals um (t). Taking into consideration the mutual independence of the XR (n) samples, this spectrum can be described by the likelihood function = W( x|λ)
M
m=1
⎡ ⎣
dnm
q=1
⎤ Wncχ 2 (xnm +q |Smq )⎦
Wχ 2 (xn )
(4.9)
n∈θn
where Wχ 2 (xn ) is the central χ 2 − distribution with parameters (4.8), and Wncχ 2 (xnm +q |Smq ) is the non-central χ 2 − distribution with parameters (4.6). It is required to determine the number M of the narrow-band signals included into the uin (t) structure, and also to estimate the values nm and dnm collection defining the frequency limits of the detected signals. At the stated problem solution, it is necessary to take into account the following: • This problem is distinguished by its essential a priori uncertainty, i.e., not only the resolved signal parameter values, but also their distribution laws are a priori unknown. To overcome this a priori uncertainty, the statistical theory of optimal algorithms synthesis recommends the use of the adaptive approach [4], in accordance with which we should use its maximal believable estimates instead of the unknown variables. • If we have a priori information concerning the probabilities of PM arriving at the specific number M of narrow-band components in the process (4.1) structure, the resolution and estimation of signals can be executed on the basis of the
Single-Channel Signal Detection
99
maximal a posteriori probability criterion. The appropriate processing algorithm a posteriori probabilities of arriving at the should maximize over the vector λ averaged energy spectrum x for the set of hypotheses H0 , H1 ,. . .,HM max , concerning the presence of a specific number of um (t) signals in the uin (t) structure. Unfortunately, this maximization can be arranged only in the form of an iteration procedure, and that is why this optimal algorithm will be extremely complicated for the calculations. • The optimal procedure corresponding to the case of probabilities PM absence, based, for example, on the maximal likelihood criterion, is merely slightly simpler for calculation because the necessity of the mutual maximization of the multi-dimensional likelihood function over the parameters set is kept, in this case. • Together with the optimal approaches to the signal detection problem, which have a great complexity, one can offer a number of quasi-optimal methods for this problem solution, based on the concept of quasi-full resolution [5]. This method allows the resolution problem to be considered as a problem of the complex detection of some signal set at which the signal resolution problem can be described by the normalized probability of false detection Pfd1 and by the probability of missing the separate, arbitrarily-selected, signal Pms1 . At that point, the algorithm ensuring, at the fixed probability of false detection Pfd1 , the minimal value of the probability of the missing signal Pms1 , is the optimal one (as per the Neiman-Pirson criterion). In practical implementation of RM systems, the operating-rate factor is extremely important. The threshold detection procedures discussed below permit us to simplify essentially the calculating complexity of the processing, at the expense of some decreasing of obtained estimation accuracy. In this connection, the exact same quasioptimal algorithms are the most claimed in practice. The simplification offered by the threshold quasi-optimal procedure, consists in the fact that the global maximization of the likelihood function (4.9) is changed by the preliminary division of all sample sets to “noise” θn and “signal” θm (m ≥ 1) sub-sets, on the basis of the differences between the distribution laws of noise and signal spectrum samples. Let H1 be a hypothesis asserting the affiliation of a XR (n) sample, having the value xn , with the spectrum of some signal um (t), the amplitude of which is Smq at the frequency n/NT; and let H0 be a hypothesis of the sample with a value xn affiliation with the frequency axis segment, where there are no useful signals. In this case, in accordance with the above-mentioned statistical characteristics of the noise and signal spectra samples, the hypotheses H1 and H0 likelihood ratio has the form: L(xn ) =
Wncξ 2 (xn |Smq ) Wξ 2 (xn )
= λ1−R e−δ 0 F1 (R, δ
xn ). λ
(4.10)
At any δ>0 with xn growth, the likelihood ratio is monotonically increased. This means that the optimal rule of spectral samples division into sub-sets θn and θm consists in the comparison of xn with the some threshold xthr :
100
4 Single-Channel and Multi-Channel Radio Signal Detection
H1 > Xn < X1¨ 1ˆ ð H0
(4.11)
The threshold level unambiguously defines the error probabilities contained in the assignment of some number of maximal noise spectrum samples to the sub-set θm , and some missing signal components, in this case, falling into sub-set θn . Since it is assumed that the decision about the number of observed useful signals is made on the basis of the assigned sub-sets θm of the signal spectrum components, the selection of xthr exclusively affects on the parameters of the resolution algorithm as a whole. The threshold algorithm of the narrow-band signal detection includes the following steps: 1. The set of signal spectrum components is assigned on the basis of the rule (4.11). 2. Signal spectrum samples, located in order of their increasing number n, are divided into several sub-ranges Qj , 1≤ j ≤ J. At that point, the signal sample with the least number XR (n1 ) always belongs to the sub-range Q1 , and the affiliation of the subsequent samples is determined by the iterative grouping rule XR (nk+1 ) ∈
Qj , if XR (nk ) ∈ Qj and nk+1 − nk ≤ dnrch Qj+1 , if XR (nk ) ∈ Qj and nk+1 − nk > dnrch
(4.12)
where dnrch = N · T · dfrch is the maximal possible sample number representing the signal spectrum in one radio channel. 3. For each sub-range, the minimal and maximal number of samples falling in this sub-range is determined, and the number of useful signals in each sub-range is determined by the rule: Mj = int[1 + (nmax j − nmin j )/dnrch ]
(4.13)
where int[.] is the symbol of the integer part of the number-taking. 4. For Mj signals formed in the sub-range Qj , the most probable variant of the sample set nmin j ...nmax j is division into the sub-sets θm , and, for each of these sub-sets, the central frequency and the signal-spectrum width estimation is executed by the correlation approach. Let the frequency band of the simultaneous analysis correspond to the averaged spectrum-sample set with the numbers nmin ≤ n ≤ nmax . From the above-mentioned description of the threshold algorithm, we find that, for false signal resolution to occur, it is enough that at least one signal from the noise spectral sample exceeds the level xthr1 . If the number of such samples is equal to Nn , the probability of false signal detection is:
Single-Channel Signal Detection
101
Pfd1 = 1 − P{XR (n) < xthr1 , n ∈ θn } = 1 − FχN2n (xthr1 )
(4.14)
where Fχ 2 (x) is the function of noise sample distribution. To ensure acceptable detection quality, it is necessary to provide a low false-detection probability, at least Pfd1 < 0.01, and hence Fχ 2 (xthr1 ) ≈ 1, and thus Pfd1 ≈ Nn [1 − Fχ 2 (xthr1 )]. Evidently, before the analysis is finished, the exact number of noise samples Nn is unknown; however, to assure fulfillment of the requirements for the permissible false-detection probability, it is quite reasonable to accept Nn as equal to the analysis range width expressed in the samples Noper = nmax − nmin . In this case, the permissible probability of threshold excess, by the separate noise sample of the averaged spectrum, will be defined by the value ε = 1 − Fχ 2 (xthr1 ) ≈ Pfd1 /Noper .
(4.15)
For central χ 2 −distribution, the argument χε2 for which P{χ 2 ≥ χε2 } = ε, is approximately defined by the equation
2 χε2 ≈ λJ 1 − + xε 9J
2 9J
3 (4.16)
where J is the degree of distribution freedom, and xε is the percentage point of Gaussian distribution, which can be calculated using the approximate expression: xε = t −
c0 + c1 t + c2 t2 1 + d1 t + d2 t2 + d3 t3
(4.17)
where t = ln (1/ε2 ); c0 = 2.515517, c1 = 0.802853, c3 = 0.010328, d1 = 1.432788, d2 = 0.189269, d3 = 0.001308. Using the mentioned approximations and Equation (4.8), we get the following rule for the calculation of the threshold xthr1 , which divides the spectrum samples into noise and signal sub-sets at the single-channel processing: xthr1 (σξ2 )
=
σξ2 N
1 1− + xε 9R
1 9R
3 (4.18)
where xε is defined by (4.17), and the probability of ε is defined by Equation (4.15). The algorithm offered above, for the sample division into noise and signal sub-sets, assumes the noise intensity σξ2 as a known quantity, on the background of which the narrow-band components are observed, in particular, see (4.18). However, in accordance with the initial statement of the detection problem, this intensity is a random parameter, and not only the value σξ2 itself, but its distribution law, too, is unknown. The attempt to use the optimal maximal-likelihood estimate of noise levels leads to an algorithm that is complicated, in terms of the calculations. In this vein, let
102
4 Single-Channel and Multi-Channel Radio Signal Detection
us analyze the following approach to the noise level estimate, one that is a bit less labor-intensive. For the central χ 2 − distribution with J degrees of freedom and parameters (4.8), the expected value is defined as: MXn =
σξ2 Jλ = . 2 N
(4.19)
Hence, if the narrow-band signal’s um (t) location on the frequency axis is known, and the number Nn of noise spectrum samples is large enough, it is expedient to estimate the noise level σξ2 by means of the noise spectrum components only, rejecting the signal components. Of course, in this case, the quality of the σξ2 estimation decreases; but, if the signal component number is a small portion of all the spectral samples, the accuracy reduction is negligible and the need to determine the unknown amplitudes of the signal components Smq is eliminated. In practice, however, the sub-set of the noise samples θn , as a rule, is not known in advance and can be determined only approximately. Let us analyze two different methods that allow the division of the noise samples sub-set, at little computing expense. 1. If we ignore the presence of signal components and define the initial noiseintensity estimate by the equation 2 = NMXn = σest1,0
N Noper
nmin +Noper −1
xn
(4.20)
n=nmin
where nmin is the sample corresponding to the left frequency-band border of the simultaneous analysis, and Noper is the range width in the samples, then the obtained value (at M > 0) will obviously be too high. Nevertheless, the following equation will be true for signals with considerable intensity, even for a similar inaccurate estimation, 2 ) ≈1, P XR (n) n∈θm > xthr1 (σest1,k
(4.21)
that allows us to attribute these samples to the signal ones. Ascertainment of the fact that the energy spectrum-samples portion is affiliated with the signal sub-set allows the substitution of the rough estimate (4.20) by the more accurate 2 σest1, k =
N xn Noper − Ns
(4.22)
n∈θ / sig
where Ns is the number of signal samples found to exceed the threshold, and k is the iteration number. As a rule, after two to three similar iterations, the estimate (4.22) becomes already rather close to the true value of the noise power.
Single-Channel Signal Detection
103
2. Another approach to estimate noise level is based on the fact that, in any signalfree frequency segment, the expected value of the energy spectrum samples coincides with the variable σξ2 /N, but, at the “signal” segments, it increases. If we smooth the spectrum – whose width B W does not exceed the maximal (on frequency) interval Bmax between signal spectra (Fig. 4.2) – over frequencies, by the window, such smoothing will cause noticeable spectrum distortion only near the central frequencies of the signals um (t), but, at noise intervals, it will decrease the spectral sample variance by BW -times, without changing the expected value.
XR(n) B0
dn1
B1 = Bmax
dn2
B2
dnM
BM
Approximated result of smoothing
dnrch
nmin
n1
n2
nM
n max
Fig. 4.2 Window width selection for smoothing in frequency
At BW > 20, the smoothing effect becomes seriously apparent, and both minimal and maximal samples (among “purely noise”) differ slightly from σξ2 /N. The minimal samples belonging to the θn sub-set are the least simultaneous among all samples of the smoothed spectrum. As a result, it is not necessary to determine the useful signal location in the frequency range, in order to estimate noise intensity. The main complexity occurring at the practical application of the offered approach is the fact that, before finishing the signal resolution procedure, information on the size of the widest interval Bmax between the useful signal spectra is absent. If the maximal possible signal number M0 which can be present in the analyzed frequency spectrum, is known, we may ensure that, among the signal spectra, at least one interval can be found, which exceeds BW min =
nmax − nmin − M0 dnrch . M0 + 1
(4.23)
However, the estimate (4.23) assumes the presence of all M0 signals and their location in the frequency through strictly similar intervals, which is unlikely. In this connection, it is expedient to accept the maximal frequency interval between the spectra as being equal to the BW estimate satisfying the equation:
104
4 Single-Channel and Multi-Channel Radio Signal Detection
FB max (BW |M0 ) = δ
(4.24)
where δ ≈ 0 is the probability of the absence of an inter-spectral interval in the analyzed frequency range, which exceeds BW . Since Equation (4.24) is oriented on the maximal possible number M0 of narrowband signals, in reality, the probability of situation Bmax < BW occurring will be much less than δ, i.e., practically zero. The final curves permitting us to find the Bmax value are shown in Fig. 4.3. From the presented data, we can see that a δ increase to 10–15%, compared with the case δ = 0, which corresponds to the rule (4.23), allows the smoothing window width BW to grow by 2–2.5 times.
Bw Nn 0.35
δ = 0.15
0.25
δ = 0.10
0.15
δ = 0.05
0.05
δ=0 0
5
15
25
M
Fig. 4.3 Recommended width of smoothing window BW
Thus, the second method to estimate noise intensity assumes the calculation of the smoothed energy spectrum X˜ R (n) =
1 BW
n+(B W /2)
xi i=n−[(BW −1)/2]
(4.25)
where the smoothed window width BW is defined on the basis of (4.24), and usage of the least of the obtained values for determination of noise intensity: 2 ˜ R (n). = N · min X σest2 nNoper
(4.26)
Analysis shows that both of the above-considered, quasi-believable, estimations of noise intensity, found on the basis of maximal component elimination (1st type estimations), as well as determined by minimal smoothed energy spectrum (2nd type estimations), have the bias (systematic) error, the magnitude of which grows with the increasing of the simultaneously-observed signal number. Nevertheless, the character of this error is different: the 1st type estimations, as a rule, are too high,
Characteristics of Single-Channel Detectionof Narrow-Band Signals
105
while the 2nd type errors are too low. Practice shows that the best results can be achieved by combining the two previous procedures into a unified procedure. To perform it, noise intensity is preliminarily estimated by rule (4.26) and, after that, is made more exact on the basis of (4.22). At large volume of the presented spectral data, and for not very large loading of the analyzed frequency range by the narrowband signals (less than 30%), the accuracy of such type of algorithm does not yield to the strictly-optimal estimation procedure. Now, assuming that the error at noise-level determination is negligibly small, we can estimate signal detection quality by the single-channel algorithm formulated above as follows. The signal components of the averaged energy spectrum follow the non-central χ 2 −distribution: −δ
Fncχ 2 (xn ) = e
∞ k δ γ (0.5n + k,xn ) k! (0.5n + k)
(4.27)
k=0
where (x) is gamma-function, γ (a, x) is non-full gamma-function, and other designations and parameters are as introduced above [see (4.6)]. Since signal presence is registered after the threshold is exceeded by only one spectral sample, applying to the signal described by dnm a set of spectral components with Smq amplitudes, the probability of missing the signal is defined as Psm1 =
dnm
Fncχ 2 (xthr1 |Smq ) ·
(4.28)
q=1
Evidently, the increasing of the spectral sample number representing the signal causes the decreasing of its signal-missing probability. In this connection, it is expedient to analyze the dependence of the signal-missing probability on radio environment parameters for the most complicated case, when the signal is represented by only one spectral sample. The curves corresponding to this case are shown in Fig. 4.4. Using samples of the observed, random process uin (t) of large size (N>>1), we can ensure each signal spectrum representation, by a large enough number of spectral samples and in accordance with (4.28), to provide a rather high quality of signal detection.
Characteristics of Single-Channel Detection of Narrow-Band Signals Let us apply the above-mentioned results to the detection of a specific signal of small duration, whose position on the frequency axis is unknown. To do this, let us estimate the probability P{tdet ≤ Ts } of this signal-detection during the definite time interval, and its dependence on the signal’s properties and the ARM system’s features.
106
4 Single-Channel and Multi-Channel Radio Signal Detection Psm1
Pfd1 = 0.001
{ { R=1
0.1
Pfd1 = 0.005
R=2
Pfd1 = 0.01
Pfd1 = 0.001
0.03
P fd1 = 0.05
Pfd1 = 0.005 0.01 P fd1= 0.01 0.003 Pfd1= 0.05 0.001
a)
6
14
h2
22
Psm1
{
R =16 0.1
{
R =8
0.03
P Pfd1 = 0.001 0.001 fd1= Pfd1= 0.005
0.01
Pfd1 = 0.01 Pfd1= 0.05
0.003 0.001 0 1.5
2.5
3.5
4.5
h2
b) Fig. 4.4 Qualitative characteristics of single-channel algorithm at small (a) and large (b) number of spectrum averaging actions R
At panoramic spectral analysis, an ARM system scans cyclically L frequency bands with width F, one of which contains the signal under detection. The analysis of the separate frequency band by the ARM system assumes the taking and processing successively of R samples, after which the ARM receiver is re-tuned on a new frequency. As a result, the processing of each frequency band requires τproc1 = τret + Rτsamp
(4.29)
where τret is the time of ARM receiver re-tuning on a new frequency; and, τsamp is the time of taking (and processing) one sample. As a consequence, an ARM system’s single cycle of analysis has the length
Characteristics of Single-Channel Detectionof Narrow-Band Signals
107
τcyc = Lτ proc1
(4.30)
and ARM system performance in this mode is defined by the variable: g=
LF F = · Lτproc1 τret + Rτsamp
(4.31)
For signal detection, it is necessary, firstly, that, before time interval Ts ends, the ARM system, at least once, has time to re-tune on the frequency band containing this signal, and, secondly, that the child spectral peak after this emission was higher than the detection threshold – in at least one of the observation cycles (if there were several peaks). The small duration of the detected signal assumes that the maximal permissible duration of detection procedure Ts should not exceed several seconds. Nevertheless, time of taking and processing of the single sample is defined by portions of milliseconds, and the coincidence of ARM equipment re-tuning time with the signal beginning or finishing is unlikely; therefore, at ARM system tuning on the appropriate frequency band corresponding to the signal, we can consider that the signal will be observed in this analysis cycle during all R samples. Let us introduce the designation k = int[Ts /τcyc ]
(4.32)
to represent the minimal possible number of analysis cycles, for which the ARM system will have the time to fulfill, during time Ts . Thus, the actually observed cycle number Kcyc is equal to k or k + 1, and probabilities of the appropriate events are determined as P{Kcyc = k} = 1 − v
(4.33)
P{Kcyc = k + 1} = v
(4.34)
where v=
Ts Ts Ts − int −k = τcyc τcyc τcyc
(4.35)
is the fractional part of the ratio of observation interval and the analysis cycle duration of the ARM system. If, during the Ts time interval, the detected signal falls Kcyc times into the ARM system scanning band, using the averaged, over R, spectral estimate samples, the probability of its missing the signal at all of these cycles is Psm (R, Kcyc ) =
nfin
n=nst
Kcyc Fncχ 2 (xthr |J = 2R, δ =
h2n )
(4.36)
108
4 Single-Channel and Multi-Channel Radio Signal Detection
where xthr is the detection threshold determined by (4.18); nst ,...,nfin is the range of the spectral sample numbers corresponding to the detected signal; and, h2n is the SNR on power, at the frequencies of these samples. If the sample averaging is not used, and the signal detection is fulfilled on the basis of each sample individually, the probability of signal-missing in all Kcyc cycles is Psm (R, Kcyc ) =
nfin
RKcyc Fncχ 2 (xthr |J = 2, δ =
h2n )
.
(4.37)
n=nst
Thus, the full detection probability can be calculated as P{tdet < Ts } =
v[1 − Psm (R,1)] at Ts ≤ τcyc (1 − v)[1 − Psm (R, k)] + v[1 − Psm (R, k + 1) at Ts > τ (4.38)
where R is a samples number accumulated by the ARM system, without the retuning to the new scanning band; k and v are defined from (4.32) and (4.35), and, for calculation of Psm (..), we must apply (4.36) when using the averaged spectrum, or (4.37) for the non-averaged spectrum.
Single-Channel Detection of Radio Signals With POFT Let us apply the above-obtained results to the detection problem for signals with programmable operating frequency tuning (POFT). Such signals possess high energy security and are used for the transfer of a large number of messages, alternatively re-adjusting among them by the random rule. For POFT signals, the averaged power (over the large time interval) falling on each frequency is comparable with the natural additive noise, which is the serious obstacle for this power detection. At the same time, the instantaneous power accompanying the data transmitting at the given frequency is rather large; therefore, at coincidence of the radio environment analysis moment with the time interval of using the frequency, the radio emission event can be registered. When using the modern, high-performance, ARM systems in the panoramic spectral analysis mode, the intensity peaks corresponding to typical radio signals with fixed operating frequency are observed during many analysis cycles, as a rule. The peaks caused by POFT signals are almost always singular because the probability of recurring usage of some frequency – strictly after the time interval equal to one ARM system analysis cycle length – is very small. Thus, the presence of the activity peaks periodically observed in some frequency sub-set, which can be described by the large enough amplitude and by the approximately constant spectrum width, is the characteristic sign of POFT signals used within the frequency range controlled by an ARM system. Of course, the single spectral peaks can be caused by the noise also, but, in this case, the peak amplitude,
Single-Channel Detection of Radio Signals With POFT
109
as a rule, is not large, and the spectrum width fluctuates randomly. As a result, if the observation at the given frequency of at least two single peaks with the spectrum width corresponding to a POFT signal is selected as a criterion of some frequency affiliation to the POFT frequency set, then the probability of the mistaken classification is small.
Probabilistic Features of the Frequency Observation Time The time diagram of the mutual operation of a POFT signal transmitter and an ARM system is shown in Fig.4.5. This diagram assumes that the ARM system scans cyclically L frequency bands in which M frequencies are located randomly, and that these positions are used at the formation of the POFT signal. As in section “Characteristics of Single-Channel Detection of Narrow-Band Signals”, we assume that, at the analysis of the single scanning band by the ARM system, before the re-tuning to the new frequency, R samples are accumulated and mutually processed. However, now, the interval τem1 of the continuous usage of the separate frequency by the POFT signal is very small (it coincides with the time of the sample set-taking, up to the order of magnitude). Therefore, the observation time ξ of the separate frequency by the ARM system is random variable and the maximal possible ξ value is τmax = min τem1 ;Rτsamp .
(4.39)
Let us select any arbitrary m− th position from the total M number of frequencies used with a POFT signal, and examine the one arbitrary activity case for this frequency, having analyzed the random observation time ξ by the ARM system of
Frequency position of POFT signal
F, MHz
L scanning bands of RM system
POFT signal operating frequency variation
fM
… l-th scanning band
fm
f3 f2 f1
τ ret t0 t0 + tobs1
Rτ samp t0 + τcyc
Fig. 4.5 Time diagram of POFT signal detection
t,s
110
4 Single-Channel and Multi-Channel Radio Signal Detection Partial or maximal registration region
Fig. 4.6 Calculation of distribution law of active position observation time ξ
–τem 1
0
Region of missing the emission of the frequency position R•τsamp
t, s τcyc–τem1
Moment of analysis start for l-th frequency band
the given radio signal emission. Since both the moments of radio signal emission and the transition to the necessary frequency band analysis are random variables, let us attach the time axis to the moment of the monitoring start for exactly the same (l−th) frequency band, which contains m− th frequency (see Fig. 4.6). The emission start we consider equiprobable within the ARM system analysis cycle (4.30). At the activation moment falling into the interval from −τem1 to Rτsamp for m− th frequency, the time interval of this frequency usage at least partially coincides with the analysis time of the l− th spectrum band and hence, the registration of this event is potentially possible. If the active position falls into another part of the cycle, the frequency observation is principally impossible, due to the emission and monitoring interval non-overlap. As a result, the probability density ξ can be written as
τem1 + Rτsamp δ(z)+ Wξ (z) = 1 − τcyc 2 |τ −Rτ | , 0 ≤ z ≤ τmax + em1 τcyc samp δ(z − τmax ) + τcyc
(4.40)
If the sample number η, during which one succeeds in observing the selected position on the POFT signal, is of interest, we can follow the relations
0 < r < rmax 2τsamp /τcyc , P{η = r|τem1 ≥ Rτsamp } = τem1 − (R − 1)τsamp /τcyc , r = rmax
(4.41)
0 < r < rmax 2τsamp /τcyc , P{η = r|τem1 < Rτsamp } = τem1 + (R − 2rmax + 1)τsamp /τcyc , r = rmax (4.42) where rmax =
R, τem1 ≥ Rτsamp int[0.5 + τem1 /τsamp ,τem1 < Rτsamp
is the maximal possible number of samples within the overlap interval.
(4.43)
Single-Channel Detection of Radio Signals With POFT
111
Probability of Separate Frequency Registration Let each accumulated sample be processed independently from the others and a decision concerning the signals presence at any frequencies is made if the detection threshold is exceeded in at least one of these samples. In this case, for each specific number of samples η = r, during which, at the next emission, the ARM system will observe some m− th frequency, the conditional probability of successful spectral peak registration can be represented in the form: Pr = 1 −
nst +n
Fncχ 2 (xthr |δ = h2n )
nfin
r Fncχ 2 (xthr |δ = h2n
(4.44)
n=nfin −n
n=nst
where xthr is the detection threshold of signal samples defined by (4.18), nst , nfin are the start and the end samples included in the monitored spectrum peak, n is the permissible measurement error of the signal spectrum width (in samples), h2n is the SNR on power at the frequency of the tested sample. The freedom degree number of the non-central χ 2 − distribution J = 2, parameter λ of this distribution is equal to σn2 /N. Combining, on the basis of total probability relation, the cases, differing by the number of the observation samples η, we get the following equation for unconditional probability of m− th frequency registration at its next emission: Preg1 =
rmax
P{η = r}Pr
(4.45)
r=1
where the probabilities P{η = r} depending on τem1 are calculated on the basis of (4.41) or in accordance with (4.42). The case, when the samples accumulated during the time of l− th frequency band observation are used for the averaged spectrum calculation, is the alternative one to that considered above. Due to the mistiming of the emission and monitoring moments, m− th frequency, including to POFT signal, is observed during the random time ξ ≤ Rτsamp instead of the time intervalRτsamp . In this connection, the spectral peak caused by the activity of this position will have the lesser intensity, namely, the specific value ξ = z will correspond to the following probability of the true registration of the appropriate spectral peak: P(z) = 1 − ×
nst +n n=nst n fin
Fncχ 2 xthr |δ =
n=nfin −n
z 2 Rτsamp hn
Fncχ 2 xthr |δ =
z
×
2 Rτsamp hn
(4.46)
where, contrary to (4.44), the parameters of the non-central χ 2 − distribution are defined by the relations J = 2R, λ = σn2 /(RN).
112
4 Single-Channel and Multi-Channel Radio Signal Detection
Taking into consideration the continuous character of the random observation time ξ , the total probability of m− th frequency registration at its next emission should now be described by the equation: τmax
Preg1 =
P(z)Wξ (z)dz
(4.47)
0
Estimate of the Total Number of Registered Frequencies If the transferred message duration is Ts millisecond, the total number of frequencies, which will chaotically change each other, and appear and disappear during the emission, can be obtained as A=
Ts (1 + γ )τem1
(4.48)
where γ is the reserve factor on the POFT signal transmitter re-construction (γ ≈ 0.1). Let us examine the regular a− th message segment with one frequency duration. The probability that the specific m− th frequency will be used, and, moreover, registered, on this segment is equal to pa = Preg1 /M
(4.49)
where Preg1 is the unconditional probability of frequency registration during its next emission, calculated in accordance with (4.45) or (4.47). It can be shown that the registration cases on the specific m− th frequency will represent Poisson event flow. In accordance with this, the probability that the specific m− th frequency will be registered at least twice during all data transmitting time, will be defined as p2+ = 1 − (1 + Apa )e−Apa .
(4.50)
This probability is extended equally to all M frequencies used by the POFT signal, and their registrations happen relatively independently from each other. As a result, the total number of frequencies Mreg , which can be registered during the current data transmitting time, will represent the binomial random variable described by the distribution series k k p2+ (1 − p2+ )M−k . P{Mreg = k} = CM
(4.51)
Single-Channel Detection of Radio Signals With POFT
113
Optimization of ARM System Parameters Practical application of the calculation formulas obtained above shows that, for modern ARM systems and POFT signals with message length in some seconds, the most probable number of revealed frequencies is not large. At the same time, there is a set of parameters, which are selected at the design stage and/or ARM system application stage, which essentially affect on the success of the given problem’s solution. Improvement of the results, in particular, can be achieved using multi-channel ARM equipment or a receiver with lesser re-tuning time to the new frequency, because, usually, time spent on completing the transients is comparable or even exceeds the data accumulation (processing) time. At the same time, there is also one important parameter, the variation of which does not require any material expenses. It is the sample number R, which is processed before the re-tuning on the new frequency. The typical curve of the most probable number of the revealed frequencies as a function of this parameter is shown in Fig. 4.7. Mreg
16
1 12
8
2
4
0
3
9
15
21
R
Fig. 4.7 The most probable number of frequencies revealed for POFT signal with the length 40 s at τret = 4 ms, τsamp = 0.32 ms, L = 60, M = 32, τem1 = 3.125 ms, h2n = 3; 1 – for the averaged energy spectrum; 2 – without averaging
We shall take into consideration the following factors. 1. At small number R of the averaging samples, a significant portion of time is spent on the re-tuning from one frequency to another, because, many times, the receiver re-tuning interval τret exceeds the time of the observation and processing of the spectral sample τsamp 2. At the averaged sample number growth till R ≈ τem1 /τsamp , the portion of time that is wasted unproductively on receiver re-tuning decreases, and the probability
114
4 Single-Channel and Multi-Channel Radio Signal Detection
that registration of the spectral peak increases allows ensuring a Mreg value close to the maximum. 3. Further growth of the R parameter is accompanied by improvement of the ratio of data processing time to the receiver frequency re-tuning time, on the one hand; but, on the other hand, the energy of the short-term observed POFT signal fragment lengthens now to the whole length of the averaged spectra set, which causes a decreasing of the effective SNR. As a result, the long segment of R values is formed, within which the Mreg value virtually does not change. 4. Finally, at redundantly large R, the influence of SNR reduction, determined by (4.46), “outweighs” the growth of the sample taking time, and system operation effectiveness starts to decrease essentially. The loss in the number of revealed positions observed at each individual data processing session, is caused by the small probability of weak signal detection (h2n = 3), when using the non-averaged spectrum. At large SNR, the situation partially changes, and rejection of the spectrum average for large R may allow the detection of more frequencies, compared with the average application. The reason for such deflection is the fact that, at large SNR, the probability of successful spectral peak registration – even on the basis of the single sample – seems to be very large, and independent sample analysis eliminates the typical averaged spectrum negative effect of a “decreasing of effective SNR value”. Nevertheless, in similar cases, the initial segment of the curve Mreg (R) keeps the type shown in Fig. 4.7, i.e., at small sample number, the averaged spectrum application is evidently more effective. From a practical point of view, the overstating of averaging samples is disadvantageous. It causes a reduction in the information renewal rate, in the monitored frequency band, and, therefore, the search of such number of samples R, accumulated before the re-tuning to the new frequency, is of considerable interest. Then, simultaneously with the high effectiveness of the POFT signal-frequency reveal, we can keep the high operation rate. The sample number Rˆ is found optimal, which ensures a probability value (4.51) close to maximum, in conformity with the large range of initial data. The fulfilled analysis shows that, for the most part, the sample number Rˆ is defined by SNR. The value Rˆ is more than 1, only for fulfillment of
R 10 6 Fig. 4.8 Optimal number of averaging samples vs. SNR, in cases differing from τsamp ) and the narrowband m ≤ 1 (i.e., ω ≤ ) frequency modulation. In the first case, as a rule, one considers the components with n ≤ m + 1 numbers. This corresponds to the FM signal spectrum width at sinusoidal modulation within 99% of signal energy concentrates f = 2(f + F) ≈ 2f ·
(6.9)
At small FM index values (from 1 to 2.5), we shall use the formula f = 2F(1 + m +
√ m) ·
(6.10)
Modulation in Communication and Broadcast Systems
145 Δf
A F
f0
m=5
f
Δf
Fig. 6.6 FM signal spectrum
Beyond this band, the component amplitude is 100 times less that the nonmodulated carrier amplitude. At m δ perm
No
Fig. 12.7 Structural diagram of the algorithm for the monitored zone radius R2 calculation when estimating computing facility security against information leakage through CEE channels
In Fig. 12.7, we can see that the radius R2 of the monitored zone is defined, in an iterative way, as the distance at which, for each computing facility’s blocks and for all frequency intervals Fi , the signal/noise ratio does not exceed the maximal permissible by RMD value δperm . Laboratory investigation of the computing facilities includes also the calculation of zone radii r1 and r1 , representing the minimal permissible distances from the computing facility to the auxiliary technical means and systems and its cable connections having exit outside the monitored zone. The radius r1 is calculated with respect to the lumped random antennas, while the radius r1 with respect to the distributed random antennas. The calculations of these variables are executed similarly to the calculation of zone radius R2 , but with the substitution of the field strength of normalized noise Enn (f ) with the sensitivity levels of lumped and distributed random antennas, which are defined by RMD. Since SMO-PRIZ application repeats the recalculation of the security indices directly, as far as the data arrival, the final value of the R2 radius is available just after completing the data entry concerning the blocks ensemble of the analyzed computing facility. In this connection, the operator can check the variation of information security indices directly during the data entry, and the testing protocol is ready for printing just after finishing the information entry.
488
12
Protection Against Information Leakage Via CEE Channels
Thus, the software support of the ARK-D1TI system ensures the full-scale investigation of the computing facility properties, in the automated mode. The calculation results and the protocols formed as a result of the investigations completely correspond to the requirements of RMD enacted at present.
Information Security Monitoring The purpose of both calibration testing and effectiveness estimation of enacted security measures is to determine RMD correspondence of the spurious electromagnetic emission and pick-up levels on the boundary of the monitored zone. Security against information leakage through CEE channels is considered acceptable if all SNR obtained at the analysis of the computing facilities’ blocks do not exceed the limit value corresponding to the category of the testing information object. All investigations of information security on information objects are really executed on the basis of the algorithm, whose structural diagram is presented in Fig. 12.8. The differences consist in the calculation details for the information security indices only. A peculiarity of calibration tests is their execution directly on the information object and the necessity to consider the object’s location features with respect to the potential places of radio monitoring equipment location, as well as its real observed signal attenuation at electromagnetic field propagation to the boundary
Start
1. Calculation of signal component intensity for CEE informative components for all frequencies f j and all blocks of the computing facilities 2. Calculation of propagation and attenuation parameters for signals 3. Selection for the calculation as initial the computing facility block m = 0.
Calculation of maximal observed SNR Δ m for signals of m − th block of the computing facility in the places of possible location of interception equipment and its comparison with the permissible limit δ
Finish
No
Have another blocks?
Yes
m = m+1
Fig. 12.8 Structural diagram of information security estimation algorithm processed by the computing facility
Information Security Monitoring
489
of the monitored zone, etc. The theoretical consideration of such peculiarities is rather complicated and insufficiently accurate [4], therefore, the calculation of security indices during calibration tests is fulfilled by the determination and usage of real signal attenuation factors ⎧ ⎪ ⎨
Edj /ERm j Kpj = Edj K0j (R) ⎪ ⎩ E K (R ) Rm j 0j m
at Rm = R, (12.22)
at Rm < R,
where Edj is the signal field strength corresponding to the frequency fj and formed by the auxiliary generator near the computing facility; ERm j is the field strength measured at distance Rm from the computing facility; R is the distance to the place of possible location of the interception equipment. At estimation of the information’s security against leakage through a CEE channel, calculation of the information security index m is fulfilled in accordance with the equation ⎧! ! ⎪ 2 ⎨" " " m Esmj 2 Knm " m = max # # i ⎪ 2 Kpj ⎩ j
Fi
[Enn (f )]2 df
⎫ ⎪ ⎬ ⎪ ⎭
(12.23)
where Esmj is the signal CEE component strength (measured in μV/m) radiated by the m–th testing block of the computing facility at the frequency fj , and calculated in accordance with (12.20); m is the off-duty factor of the testing signal for the m–th block of the computing facility; Knm is its capacity factor; Enn (f ) is the field strength of the normalized noise, corresponding to the current type of interception equipment; Fi is the frequency interval (12.21), to which the tested CEE components belong. At investigation of information leakage danger due to CEE pick-up, calculation of the signal component intensities of the induced voltages is executed in accordance with the rule Usmj =
/
2 − U2 Uimj nmj
(12.24)
where Uimj is the voltage induced in the testing line at frequency fj , at operation of the m−th block of the computing facility in the active mode;Unmj is the voltage created in the same line by the natural electromagnetic field, at a switched-off computing facility. The rule for calculation of information security index m takes the form: ⎫ ⎧! ! ⎪ ⎪ 2 2 ⎬ ⎨" " " m Usmj " 2 # Knm # [he (f )Enn (f )] df m = max (12.25) ⎪ i ⎪ 2 Klj ⎭ ⎩ j Fi
490
12
Protection Against Information Leakage Via CEE Channels
where Klj is the real signal attenuation factor in the testing line; the effective height he characterizes the properties of the tested line as a random antenna (this function is approximated on the basis of the measurement results); parameters m ,Enn (f ) and Fi remain the same as for the analysis of the CEE channel. In cases when, for improvement of information security at the information object, the system of active noisiness (SAN) is used, the rule (12.25) for calculation of the security index m takes the following form: ⎧! ⎫ ! " 2 ⎨" " Enk 2 ⎬ " m Esmj 2 Kn Knm # , m = max # i ⎩ 2 Kpj Kpnk ⎭ j
(12.26)
k
where Enk is the noise field strength value created by the SAN at the k–th frequency; Kpnk is the real noise attenuation, created by the SAN generator; Kn is the factor of SAN noise quality. Other parameters, including the rule (12.20) for calculation of CEE signal component strength Esmj , completely correspond to the earlier discussed cases. Lastly, the rule for the calculation of information security index m , mentioned in the algorithm in Fig. 12.8, takes the following form at effectiveness estimation in conformity with the pick-up channel: ⎧! ⎫ 2 1 ⎨" ⎬ " m Usmj 2 2 Unlk Kn Knm m = max # , ⎭ i ⎩ 2 Knj j
(12.27)
k
where Unlk is the RMS value of the noise voltage created in the line by the SAN generator at the k–th frequency; the calculation of informative component intensity for the induced signals is fulfilled as per the rule (12.24), and all other designations correspond to the earlier discussed cases.
SMO-PRIZ Application Operation for Information Security Monitoring Figures 12.9 and 12.10 show the SMO-PRIZ application windows for the information security-monitoring mode, in cases when the SAN system is used on the information object and cases when it is not. As earlier, the initial columns of the table presented in the left part of the window contain the results of the intensity measurements for CEE informative components. Since the example in Fig. 12.9 illustrates the pick-up analysis, the information on the intensity of the induced signals is presented now by the measured voltages, and the tested line is characterized by the effective antenna height and by the attenuation factor: unified for all types of radio monitoring equipment. Some table columns represent the calculated (in accordance with Equation (12.24)) level of signal component, the number of frequency interval, to which the
Information Security Monitoring
491
Fig. 12.9 Main window of the SMO-PRIZ application, at information security analysis against leakage through the pick-up channel (SAN system is absent)
Fig. 12.10 Main window of the SMO-PRIZ application, at effectiveness estimation of used information security measures against leakage through the pick-up channel (SAN system is activated)
current CEE component belongs, and, in “SNR” columns, the calculated SNR for these frequency intervals at the boundary of the monitored zone. The SMO-PRIZ operation mode at effectiveness estimation of the used information security measures (see Fig. 12.10) is, in many respects, similar to its operation at information security estimation, and differs only by the additional data account characterizing the operation of the security system. These data are represented in the right part of the main window of the SMO-PRIZ application. The additional tool panel located directly over the data table of the SAN system, and the “SAN” section of the main menu of the application allow the saving of the SAN characteristics on
492
12
Protection Against Information Leakage Via CEE Channels
the hard disk and the earlier saved (or created by the external applications) tables to be involved in the calculation. An example presented in Fig. 12.10 shows that the noisiness system application allows provision of essentially smaller values of the observed SNR (see contents of the “SNR” columns). In that way, the reliable requirement fulfillment of the “Regulations of information security” in all frequency ranges can be realized. Above-mentioned examples illustrate the application of the ARK-D1TI radio monitoring system for solution of the problems of information security estimation, processed by computing facilities. Application of the SMO-TESTER and SMODX applications allows the detection of the CEE informative component ensemble, and the SMO-PRIZ application is capable of calculating all information security indices required by RMD. At that, SMO-PRIZ calculates the information security indices directly during the data acquisition. The operator has complete control over the calculation procedure, and the testing protocol is available for printing, just after the completion of information entry. Moreover, if necessary, the SMOPRIZ application for the information security calculation can be used together with not only the ARK-D1TI system, but with other radio measurement equipment. For automation of the measurement result transmission, the SMO-PRIZ package includes a special module intended for external application data conversion into the format of the SMO-PRIZ database. This makes it possible to use the SMO-PRIZ program for the measurement data processing of other programs different from SMO-DX. At the same time, for the considered software package, the following two shortcomings are typical: • The SMO-DX application uses only the classical double-stage procedure for CEE informative component search (TDS algorithm), which may lead to essential time expenses at analysis execution • The mutual operation of SMO-DX and SMO-PRIZ does not allow, in full, automation of the process of execution and saving of the investigation results. In connection with these mentioned shortcomings, in 2006, a new software package – SMO-THESIS – was developed in Russia, which expands the functions of SMO-DX and SMO-PRIZ applications.
Purposes and Functions of SMO-THESIS Application The SMO-THESIS application for information testing was developed in 2006, and allows the provision of laboratory (bench-top) testing of the computing facilities, as well as attestation of the fact that the information object is processed by the computing facilities. To detect CEE informative components, this application may use both the TDS method and the TDM method. Using the TDM method ensures the essential time reductions necessary for location determination on the frequency axis of all CEE informative components. SMO-THESIS automates also the intensity
Purposes and Functions of SMO-THESIS Application
493
measurement of the noise emission created on the monitored computing facility by the noise generator, at using the system of active noisiness and some other indices anticipated by RMD. SMO-THESIS application is capable of providing most information security investigations in the automated mode. The automated mode (Fig. 12.11) assumes that the SMO-THESIS package takes the testing control on itself. It is enough for the operator in this case to be guided by the requests and program recommendations. A similar approach essentially simplifies the operator actions and ensures a high testing rate. In addition to the automated mode, the SMO-THESIS package offers to the experienced operator the possibility of “manual” rechecking of the obtained data. At any moment, the operator has the possibility to investigate in detail (having stopped the automatic task execution) an interesting section of the frequency range. Having distantly controlled the condition of the equipment under test, the operator can visually monitor the current spectrum variation and also track “orally” the characteristics caused by the testing equipment. The demodulator used for listening can be chosen from a large set of variants, directly during the testing. Lastly, there is the possibility to form the sequence of testing acoustic signals, ensuring the check of acoustic-electrical conversion presence in the monitored equipment.
Fig. 12.11 Investigations in automated mode
494
12
Protection Against Information Leakage Via CEE Channels
At determination of information security indices on the information object, the SMO-THESIS application automates the estimation of the real signal attenuation factor, the signal attenuation factor at propagation pick-up lines, and also the effective heights of random antennas. Using these parameters, for each potential information leakage channel, the value of SNR is calculated at the monitored zone boundary, allowing comparison of the real information security with RMD requirements. Let us consider, in particular, the determination of the signal attenuation factor in the line used for the estimation of information security against leakage via the pickup channel of the information object. To determine the attenuation factor, one can connect two high-frequency voltage probes to the analyzed line, at distance l, each from the other (Fig. 12.12). Into the line at each j–th frequency, the signal from the auxiliary source (the generator of noise signals or the generator of sine signals with the random initial phase) is inserted. Measurement of the induced voltages in the points of connecting the probes allows the determination of the attenuation factor at j–th frequency Klj = Ur1j /Ur2j
(12.28)
where Ur1j is the voltage near the computing facility; Ur2j is the value at the boundary of the monitored zone. Having obtained the data from the equipment, the operator can switch the SMOTHESIS package to the nominal mode intended for the calculation of the information security indices against leakage through the CEE channel. The indices calculation is executed in accordance with the RMD changed in 2005. The SMO-THESIS package allows the calculation of:
Boundary of the monitored zone Computing facility
R
l
r1
B
A
Signal generator
Voltage probe
Complex ARK-D1TI
Fig. 12.12 The circuit for the line attenuation factor measurement
Voltage probe
Purposes and Functions of SMO-THESIS Application
495
• The radius of the monitored zone for the computing facilities, necessary to avoid information leakage through CEE channels • Information security indices, which are processed by the computing facilities, against the leakage through CEE channels on the auxiliary technical facilities and systems • Effectiveness estimation of used information security measures against the leakage through CEE channels. At determination of the monitored zone radius on the basis of the given test ensemble and the selected (by operator) frequency range sets, the SMO-THESIS package provides the following: • Detection of the frequency list of informative CEE • Estimation of detected components’ intensity • Calculation of the monitored zone radii R2 , r1 and r1 , ensuring information security against leakage through the CEE channel and the pick-up channel • Formation of testing protocols corresponding to RMD. In the course of estimating the adopted security measure’s effectiveness, the SMO-THESIS package allows the estimation for the indices of SAN operation and the calculation of the observed (on the monitored zone boundaries) value of the SNR. An example of the SMO-THESIS package’s operation in this mode is shown in Fig. 12.13. In this case, the recorded testing protocol can help to reveal the reasons
Fig. 12.13 Calculation of security indices and saving of investigation results
496
12
Protection Against Information Leakage Via CEE Channels
for the detected discrepancy and to provide the measures for the elimination of these shortcomings of the used system of information active noisiness. The investigation results are saved in the internal database of SMO-THESIS. The information on the various tested objects is saved independently, which allows – without considerable restrictions – the interruption and renewal of testing to be executed for the specific information object and also (if necessary) to use the information obtained during the previous testing.
Conclusion In this chapter, the peculiarities of informative CEE detection in the technical facilities of information processing are considered. Descriptions of the two following algorithms of the informative components are given: the typical double-stage algorithm and the single-stage algorithm of mutual detection and checking of the components with regard to information presence. The single-stage algorithm has the better indices on time expenses, but, for its implementation, ARM equipment with high spectral resolution is necessary. The ARK-D1TI multi-functional radio monitoring system is the modern radio measurement equipment that allows the effective estimation of information security against leakage through CEEP channels. The system software includes a modified version of electromagnetic environment analysis application (SMO-DX) and a special application for the information security calculation (SMO-PRIZ), which permits the execution of the calculation of the monitored zone radius for the computing facilities, the information security indices against leakage through CEEP channels, and the effectiveness estimation of adopted security measures. SMO-PRIZ is certified and fully corresponds to RMD of 2005. To increase the effectiveness of information security investigations, SMOTHESIS was developed, in which the algorithm of mutual detection and property determination of the informative CEE is realized. At investigation of information leakage danger through CEEP channels, the detection of CEE components is provided for the equipment under test, as well as the automatic detection of informative emission components, the measurement of their intensity, and the calculation of information security indices.
References 1. Burmin, V.A., Bykovnikov, V.V., and Tupota, V.I., Multi-Functional Complex for Monitoring of Information Security Effectiveness (in Russian). Special technologies. 2002. Special Edition, pp. 63–75. 2. Kuznetsov, Yu.V., and Baev, A.B., Methods of CEE Measurement: Comparative Analysis (in Russian). Konfident. No. 4–5, 2002, pp. 54–57. 3. Tupota, V.I., Kozmin, V.A., and Tokarev, A.B., Multi-Functional Complex ARK-D1TI Application for Estimation of Information Security Against the Leakage via CEE Channel (in Russian). Special technologies. No 2, 2006, pp. 51–56. 4. Frolov, V.Yu., Pogorelov, A.A., and Petrov, V.V., Calculation of Standard Attenuation Factor of Electromagnetic Field on the Basis of Electromagnetic Field Attenuation Model in Free Space (in Russian). Zaschita informatsii. INSIDE Publisher, No 3, 2005, pp. 82–85.
Conclusion
Respectful reader! The authors of this book hope that, after becoming acquainted with it, you have begun to understand deeply the peculiarities of the problems, which are solved by radio monitoring systems, and that the book has given you a new-found appreciation for the technical features of such systems. The book introduced you to the principles of the digital radio receiver structure, the radio direction-finding of radio emission sources, the localization of the technical channels of information leakage, and the equipment for special investigations of compromising electromagnetic emanation and pickups, as well as the mathematical methods underlying the operation of radio monitoring equipment. There is information in the book related to research, engineering, and applied science, which we feel will be useful for undergraduate and post-graduate students of universities and colleges, who study the fields of radio engineering and electronics, as well as for experts from state and commercial organizations, who deal with radio monitoring and information security. We attempted to describe both the theoretical problems of automated radio monitoring equipment, as well as the specific equipment and the software samples. The authors are active employees of the Russian company, IRCOS, and directly participate in the development and manufacturing of radio monitoring equipment. Due to this, the authors are well versed in its features, and therefore, the descriptions and discussions in this book were based on examples of the equipment manufactured by this and other Russian companies. The long-term successful experience of developing radio monitoring equipment and systems confirms the high potential of Russian experts in this area. The technical solutions used in the equipment and systems are protected by Russian patent, are confirmed by application and independent engineering expertise, are implemented into modern radio engineering systems, and have been honored with various Russian awards. Engineering development occurs very fast and these technical equipment parameters, which today seem state-of-the-art, after several years will be standard for this field. Nevertheless, the principles and structure approaches for the automated radio monitoring equipment described in this book are more conservative, to our A. Rembovsky et al., Radio Monitoring, Lecture Notes in Electrical Engineering 43, C Springer Science+Business Media, LLC 2009 DOI 10.1007/978-0-387-98100-0_BM2,
497
498
Conclusion
mind, than the technological developments that will allow its application in further elaborations to come. Of course, we were not able to address all the current problems of radio monitoring, and various questions will certainly be raised in the mind of the interested reader. Which problems are not included in this book? First, we think that, in a number of cases, we were not able to include enough recommendations regarding the practical applications of the discussed equipment. Unfortunately, we could not include with the book a CD with models of radio signals, radio environment, and equipment. Also, we could not consider in the book a number of new algorithms for signal processing, which increase the effectiveness of radio monitoring problem solutions. We are aware that there are shortcomings in the present book, but it is our hope that they will not affect interest in it from the experts and the profiled institutions. In further editions, these shortcomings can be eliminated with time, and new materials can be added to the book. We think that with the help of experts in this field, from various states and countries, the material in these chapters can be improved, and the authors welcome recommendations and honest evaluations. We thank responding readers for this in advance. The authors would like to express special gratitude to those users of the equipment discussed in the book, who checked its effectiveness both in civil and military fields and whose opinions and suggestions have stimulated the execution of additional investigations and developments and helped in the selection of most of the prospective engineering solutions. The authors anticipate gratefully all future reader remarks and suggestions on the book’s contents, and those comments can be directed to Springer USA or directly to us.
Subject Index
A Accelerometers, 353–355 Acoustic-electrical conversion, 493 Acoustic loud-speakers, 433 Active radio channels, search of, 130 Adcock antenna, 238–239, 270 Adjustable filters, 89 Akvedook P168E, 328 Algorithm “on accumulated spectrum”, 376 cluster, 400 correlation-interference, 384 “on exceeding the reference spectrum”, 376 of radio signal spatial selection, 454 raster, 399 single-channel, 105 testing and detection mutual (TDM), 172, 478–479, 481 testing and detection separate (TDS), 478, 492 “with fixed threshold”, 376 Amplifying-converting path (ACP), 57 Amplitude-frequency response, 27–28 Amplitude modulation, 53, 142, 145, 157–158, 162, 164, 166–167, 203–205, 219–220, 222, 272 Amplitude modulation variety, 142–143 Amplitude-phase, 142, 309 distributions, 311, 313–314 modulation, 142 shift-keying (APSK), 156 signal distribution, 310 Amplitude signal-transfer function, 53 Analog-digital conversion, 57, 72, 139, 242 Analog-digital converter (ADC), 10, 251 Analog-digital processing (ADP), 280 Analog-digital processing unit, 11, 14, 17, 70, 73, 279, 297, 299, 304, 346
Analog-digital processor (FFT processor), 433 Analog modulation, 140–147 amplitude modulation, 140 frequency modulation, 143 phase modulation, 146 Analog switchers (AS), 68 Angular velocity sensors, see Gyroscopes Antenna array (AA), 457 Antenna effect, 12, 239, 297, 304–405, 440, 451 Antenna gain coefficient, 405, 408 Antenna-mast devices, 337 Antenna-receiver unit (ARU), 295 Antenna-switching, 271 Antipodal signals, 152 Aperture sampling, 250 ARCView, 386 ARGAMAK, 67, 80–81, 83–84, 86, 89–90, 93, 125–126, 128, 257, 263, 304, 306–307, 334–336, 341, 383–384, 390, 407, 449–450, 462, 464 “Argument” mobile station, 344, 369, 393, 407, 418, 424 ARK-CO5 DSP unit, structural diagram of, 87 ARK-CT3 digital receiver, 65 ARK-D11, technical specifications of, 127 ARK-D1TI measuring complex, 482–488 CEE informative components, 484 monitored zone radius calculation, 486 ARK-D1TP receiver, 73, 75, 376 ARK-KNV3 radio signal converter, 447 ARK-KNV4 external radio signal frequency converter, 325 ARK-MT1 telescopic dielectric mast, 348 ARK-PR5 DRR technical specifications, 85 ARK-RD8M multi-channel panoramic radio receiver, 128, 341 functional diagram of, 128 technical specifications of, 129
499
500 ARK-SPM spot-jamming generation, 452 ARM station, see “Argument” mobile station Atmospheric duct, 165 Attenuator adjustment data, 80 Attenuator attenuation value, 35 Attenuators, 57, 74, 208 Audio records playback, 131 Audio signal modulation, 201 Autofahrer Rundfunk Data (ARI), 167 Automated operation modes, 132 Automated technical analysis of signals, 219–233 Automatic determination of radio transmitter location, 322 Automatic frequency hopping (FH), 164 Automatic gain control (AGC), 57 Automatic link establishment (ALE), 164 Automatic radio compass, 255, 263–265 direction finder, 266–270 monitoring, mode of, 259 signal-analysis process, 222 signals recognition, 235 Automatic signal analysis, 233–235 Auto request (ARQ) technology, 163 Azimuth calculation, 277 Azimuth estimation, 292 B Band-pass filter (BPF), 67–68, 72, 78, 89, 206–208 Bearing curves, 308–309 Bearing directions, 397 Bearing modes, 304 Bessel function, 120, 144 Bi-Di protocols, 73 Binary phase shift-keying, 221 Boltzmann constant, 36 BPSK signal, see Binary phase shift-keying British TV-Radio Company (BBC), 170 Broadband radio modems, 179 Broadband wireless local loop, 180 Bureau on Radio Communication of ITU, 200 C Calibration coefficient, 406 Calibration curve of measuring antenna, 407 Calibration tests, 488–489 Carrying capacity, 8, 321–322, 330, 332 Cartography, 334, 398 CCIR standard, 166, 205 CEE informative components, 476–477, 486 Cellular radio communication system, 249, 328, 331–333, 390
Subject Index Channel keeping method, 172 Chirp-signal, 146, 437 Citizen’s Band (CB), 158 Clock-frequency synthesizer, 92 CLOVER protocol, 164 Coaxial cable, 40, 440–441, 450 Coded designation of system, 187–188 multiplexing method, 186 transmitted data type, 186 Code division multiple access (CDMA), 176 COFDM signals, 197 Coherently-related local oscillator, 11, 335 Coherent synthesizer, 54 Comb, 14 Combined shift keying, 139 Comb sections, 14 Complex transfer coefficients (CTC), 283 Compromising electromagnetic emanation (CEE), 5–6, 471 Conference European of Post and Telecommunication (CEPT), 136 Continuous dynamic channel selection (CDCS), 178 Control and power supply unit (CPSU), 77, 79 Correlation interference method, 461 Correlation interferometric meter (CIM), 241, 251, 278 CRT monitor testing mode parameters, 475 Crystal receiver, 24 Crystal resonator, 93 Cube law, 48, 52 Cyclic redundance check (CRC), 329 D Dangerous radio emissions, detection of, 468 Data accumulation (processing) time, 113 Data radio channel (DARC), 167 Dead zone, 160 Decay slope, 36 Decoding or demodulating units, 84 Delphi/C++ Builder, 386 Demodulated messages, recording of, 130 Demodulation process, 139 Demodulator/decoder selection, 202 Detection threshold, 107–108, 111, 373–374, 376–377, 384, 481 Difference-distance-measuring (DDM) systems, 320 Diffraction theory, 409 Diffusive dispersion (re-reflection), 165 Digital-analog tracking systems, 55 Digital cartographic information, 404
Subject Index Digital communication protocols, 163 Digital demodulators, 57 Digital enhanced cordless telecommunications (DECT), 178–179 Digital frequency synthesizers, 57 Digital radio mondiale (DRM), 161 Digital radio receiver (DDR), 56–60, 297, 356 ARK-CT1, 67–73 ARK-CT3, 75–79 ARK-KNV4 external remote-controlled converter, 79–81 ARK-PR5 argamak digital radio receiver, 81–93 general principles, 56 types of ARM receivers, 58 Digital signal processing (DSP), 57, 199, 240 Digital spectrum analysis, 63, 128 Digital video broadcasting (DVB-T), 170 Directional antenna, properties of a, 238 Direction finder accuracy, 16, 19, 21, 241, 243, 252–253, 256, 280, 311, 314–315, 320, 397, 458 disadvantages of, 255 mono-pulse, 242, 248, 273, 316, 334, 384 operation accuracy of, 245 sensitivity, 246 Direction-finding (DF) stations, 16 Direction-finding errors, 243, 268, 270, 307–309, 313 correction, 307–315 Direction-finding method, 240, 242–243, 250–252, 256, 267, 309–310, 378, 460 multi-channel, 382–384, 464 Direct sequence spread spectrum (DSSS), 176 Discrete (digital) modulation, 147–158 amplitude-shift keyed signal, 147 Discrete Fourier transformation (DFT), 197, 281 Discretization errors, 350 Discrimination coefficient, 37–38 Dispatcher panel, 208 Distant monitoring of remote premises, 125 Distant radio monitoring, 438–451 ARK-D13 system, 450 ARK-D3T system, 443 ARK-D9 system, 448 construction principles, 438 examples of, 441 DLL/ActiveX, 386 Doppler effect, 161, 240, 270 Doppler shift compensation, 55
501 Double-channel method, 436 Double-channel panoramic digital receiver, 462 Double phase telegraphy, 153 Double relative phase telegraphy (DRPT), 154 Double-sideband AM (DSB), 143 Double-sideband AM with suppressed carrier (DSBSC), 143 Double side-band modulation (DSB), 219 DRM broadcasting formats, 192 DVB-T broadcasting formats, 192 Dynamic AM (DAM), 143 E Electromagnetic accessibility (EMA), 335 Electromagnetic compatibility (EMC), 25, 136 Electromagnetic district relief, 418 Electromagnetic emissions of computing equipment, 472 Electromagnetic environment analysis application (SMO-DX), 496 Electromagnetic field strength measurement, 403–424 calculation of, 408, 414 compatibility calculation, 423 distribution estimation, 406 district topography, 409 location determination, 419 mathematical relations, 404 processing of measurements, 415 transmitters checking, 421 urban build-up, 411 vegetation influence, 412 Electron-beam tube (EBT), 266 Emission parameters, 3, 6, 9, 18, 20, 428 Enhanced data rates for global evolution (EDGE), 176 EPP protocols, 73 Equisignal method, 253–255, 263 Error correction, 164, 185, 315 Euroboard standard, 86, 88 European Broadcast Union (EBU), 166 European Telecommunication Standard Institute (ETSI), 136 F Facsimile, see Image transmission system False-detection probability, 101 False switching, 367 Fast Fourier transform (FFT), 72, 196 Fast task correction during scanning, 131 Field-effect-transistors (FET), 357 Filter fine-tuning, 69 Forward error correction (FEC), 164
502 Frequency converter, 12, 14, 17, 22, 26, 30, 65, 67, 71, 89–90, 125, 128, 334, 341, 348, 447 Frequency correction coefficients, 80 Frequency deflection, 151, 196, 198, 271 Frequency deviation, 205 estimation, 212 Frequency measurement, 195–201 FFT method, 197–198 instantaneous method, 196–197 spectrum width measurement, 199–201 Frequency mixer, 31 Frequency modulation index, 143, 146, 151 Frequency-shift keyed signal, 148 Frequency synthesizer (FS), 54, 280 Frequency-time diagram, 373 Full detection probability, 108 G G4–164 signal generator, 363 Gain coefficient, 24, 37, 405 Gain-transfer characteristic (GTC), 44 Gasoline electric generators (GEG), 369, 371 Gating, 220–221 Gaussian distribution, 101 Gaussian FSK (GFSK) signal, 148 GeminiPD+, 328 General package radio service (GPRS), 176, 331 GeoConstructor, 386 Geodesic line view, 237 Geo-information systems (GIS), 386 Gilbert transform, 197, 290–291 GISToolkit Panorama, 386 Global positioning system (GPS), 168, 352 Global system for mobile communications (GSM), 175 GWX ActiveX control, 386 Gyroscopes, 353–355 H Handheld radio direction finder ARK-RP3, 256–262 ARK-RP4, 262–263 Heterodyne, 26 High-pass filter (HPF), 68 High performance local area network (HiperLAN), 182 Homing method, 20, 395 Horn measuring antenna, 79 Hut model, 165 I IEEE standards, 192 Image transmission system, 162
Subject Index Inertial navigation system (INS), 351 Inertial sensors, 353–355 Information leakage, electromagnetic channels of, 428–430 Information security calculation, 492, 496 Information security index, 471–473, 488–490 calculation of, 473–474 Information security monitoring, 488–492 Information transmission, 1, 3, 220, 435 Input-disturbing signal level, 48 Instantaneous frequency measurement (IFM), 196 IntegraTR or T-96SR radio, 328 Interceptors, 427 Interferometer correlation, 273–281 double-channel, 293 measuring system, 281–289 N–channel correlation, 279 phase, 251, 273–278 principle, 356 single-channel measuring system, 289–295 structural diagram of, 273 Intermediate frequency, 11, 25–26, 31, 33, 55, 57, 78, 80, 89, 198, 221–242, 266–267, 270, 281, 297, 304, 337, 341, 363, 377, 379 Intermediate frequency unit (IFU), 78 Intermodulation characteristics, 46 Intermodulation component (IC), 47, 49, 51 International Consultative Committee on Radio Broadcast, 166 International frequency range distribution (IFRD), 188–191 International Organization of Radio, 166 International Telecommunication Union (ITU), 136 International teleprinter alphabet (ITA2), 163–164 Intrusion protection systems, 189 Inverse operation, 153 Inverse weighting distance method, 417–418 Ionosphere irregularities, 165 K Kaiser-Bessel window, 72 Kotelnikov theorem, 198 L Lacing, 200 LCD monitor testing mode parameters, 475 Likelihood functions, 96, 479 Limiters, 57 Linear interpolation method, 74–75
Subject Index Linear voltage regulators and stabilizers, 360 LO harmonics, 27, 32 Low frequency modulating oscillation, 142 Low-noise amplifier (LNA), 41 M Malfunctions in communication channels, 203 MapInfo, 386 Marker pegs, 243 Mark frequencies, 151 Maximal-likelihood, 99, 398–399, 480 Measurement mode, 221–231 Measuring receiver ARGAMAK-I panoramic receiver, 93 ARK-D1TP digital panoramic receiver, 73–75 Micro-cellular communication systems, 178 Microprocessor-controlled (MPC) unit, 90 Minimal shift-keying (MSK), 220, 151 MobiDARC, 168 Mobile direction finder, 249, 297, 308, 311–313, 315–316, 393, 395, 397, 402, 457, 461, 464 ARTIKUL-M1, 299–301 ARTIKUL-M4, 295–299 ARTIKUL-P, 301–306 ARTIKUL-P11, 306–307 Mobile radio monitoring stations, 289, 364–365, 404, 428 Mobile telegraph, 179 Modern radio electronic signals, 158–188 international system for signal designation, 182–188 SW range signals, 158–164 VHF range signals, 165–182 Modulating signal, 139–143, 146, 167, 199, 204–205, 209, 211, 272, 437 auto-correlation function of, 320 binary, 147–148 coded designation of, 185 filtering of, 150 sinusoidal, 140–141, 211 Modulation parameters, measurement problems of, 318 Modulation theory, 135 Modulation type, determination of, 201, 203, 205 Moore code, 164 Morse code, 182 MPEG-4, 162 Multichannel multipoint distribution systems (MMDS), 169 Multi-channel receivers
503 ARK-D11 double-channel complex, 125–126 ARK-RD8M multi-channel complex, 126–130 panoramic multi-channel receivers, 123–125 Multi-path propagation, 240 Multipath radio waves propagation, 270, 460 Multiple-pass panoramic coverage mode, 60 Multiple signal classification (MUSIC), 241 Multi-position radio monitoring, 390 N Narrow-band components, 95, 98, 101, 479 Narrow-band frequency modulation (NFM), 146, 162, 304 Narrow-band signal threshold algorithm of, 100 double-channel detection of, 117 Navigation software package, 398 Near instantaneous companded audio multiplex, 170 Neiman-Pirson criterion, 99 NICAM, see Near instantaneous companded audio multiplex Noise immunity of receiver, 58 Non-linear law, 52 Non-linear radar, 428 Normal time interval, 175–176 O On-board network of aircraft, 359 One-signal selectivity, 34, 43 Operation concealment, 396, 427 Optimal maximal-likelihood estimate, 101 Orthogonal frequency division multiplexing (OFDM), 161 P PACTOR, 164 Paging communication, see Mobile telegraph Panoramic measurement receiver, 59 Panoramic receiver, 8, 55, 58–59, 93, 257, 281, 283–286, 292, 335, 341, 407 Panoramic spectral analysis, 130–131, 133, 257, 429, 433 Panoramic technical analysis (PTA), 362 ParagonPD+ basic station, 328 Path transfer function, 28, 44, 46 Periodograms, 478–481 Phase crosstalk distortions, 54 Phase deflection amplitude, 146 Phase delays, determination of, 284 Phase detectors, 207–208
504 Phase direction finders, 250 Phase-locked-frame (PLL), 265 Phase-locked-loop methods, 205 Phase shift-keying (PSK), 152 Point-of-multiple-access, 327 Point-point type transmission, 332 Poisson event, 112 Polarization angle, 238, 245 Polarization direction finders, 250 Post-processing signal analysis, 208 Power gain factors, 39 Pre-detection and post-detection amplification, 24 Programmable operating frequency tuning (POFT), 13, 108, 115 signal detection, 115–116, 123 signal direction finding, 382–383 signal-reveal method, 116 signal transmitter, 109, 112 Pseudo-distances, 352 Pseudo-Doppler systems, 240 Pseudo-random sequence, 177 Pulse amplitude, 139 duration, 139, 473, 475, 486 repetition rate, 139 signal polarity, 167 voltage stabilizers, 360 Purely noise, 103 Q Q–factor, 24 Quadrature amplitude modulation (QAM), 143, 157 Quadrature amplitude shift-keying (QASK), 157 Quadrature modulation (QM), 220 Quadrature phase shift-keying with the shift (OQPSK), 220 Quasi-doppler direction finders, 270–272 Quasi-doppler systems, 251, 316 Quasi-optimal methods, 99, 118 Quasi-peak detector, 75 Quasi-stationary method, 396–397 R Radial communication system, 171 Radio amateurs’ zones, 171 Radio amateur standard, 163 Radio communication regulations of Russian Federation, 136 Radio data system (RDS), 167 Radio direction finders, 237, 239–251, 253, 262, 319–320
Subject Index structural diagram and characteristics, 241–242 technical parameters of, 242–250 accuracy of direction finding, 243 being-found signals, 249 cost, 250 deployment time, 249 noise immunity, 247 operating frequency range, 249 operating rate, 248 resolution, 248 sensitivity, 246 Radio direction-finding technique, 238–241 Radio electronic environment (REE), 2 Radio electronic intelligence vs. radio frequency spectrum, 318 Radio electronic means (REM), 1, 5, 9, 136 Radio emission occupied frequency band of, 199 classification, 203 detection, 435 sources (RES) drive method, 395 automatic calculation method, 398 localization methods, 395 peculiarities of multi-channel direction finding, 401 quasi-stationary method, 396 simultaneous direction finding, 402 Radio-Ethernet operating, 327 Radio frequency center, 414 Radio frequency interference, 370 Radio frequency spectrum (RF), 135 administrative division of, 137 Radio-game, 437 Radio microphone detection of, 442 detection problems, 428 identification of, 437–438, 442, 453 localization of, 437–438, 453 polarization type of, 433 Radio-monitoring devices, 11 Radio monitoring equipment characteristics, 16–21 detection systems, 16 manpack ARM equipment, 19–21 portable equipment, 18–19 radio monitoring and RES location stationary and mobile stations, 16–18 classification of, 6–9 design constraints, 8 performance, 8 design philosophy, 9–12
Subject Index technical parameters, 12–15 quality criterion selection, 12–13 Radio monitoring stations, 195, 200, 262, 319, 344, 350, 357, 362, 364, 369–370, 398 Radio monitoring system Archa stationary station, 334–338 Arena portable station, 346–347 Argument mobile station, 338–346 control arrangement in system, 326–333 combined ARK-POM, 332 low-speed radio channel, 328 mobile and deployed posts, 328 radio modems, 331 stationary posts, 326 electric power supply, 356–372 ARK-UPS12, 365 autonomous electric station usage, 369 multi-channel pulse source, 364 pulse power supply of low power, 362 radio equipment sources, 357 requirements for, 356 secondary sources, 360 location determination system, 318–320 mast devices, 347–349 navigation systems for, 349–356 features of, 350 mobile stations navigation, 355 special software support, 372–390 Structure of, 320–326 ARK-POM1, 321 ARK-POM2, 322 ARK-POM3 geographically-distributed system, 322 combined ARK-POM system, 323 Radio-monitoring unit, 59 Radio navigation, 23, 147, 237, 352 Radio network configuration, 172 Radio-operators, 158 Radio-phone communication, 171 Radio receiver parameters, 27–56 amplitude-frequency response, 28 attenuator influence, 51 blockage effect, 52 crosstalk distortions, 53 inherent noise and sensitivity, 36 intercept points, 47, 52 intermodulation noise, 43 intermodulation-free dynamic range, 50 main and spurious channels, 30 multi-signal selectivity, 43 phase noise, 54 pre-amplifiers, 39, 42
505 RR selectivity, 34 voltage standing wave ratio, 29 Radio signal converters, 92, 450, 461 demodulation, 64, 91 detection in monitored premise, 430–437 detection algorithm, 435 detection effectiveness dependence, 436 generalized structure of equipment, 432 near-field and far-field regions, 430 deviation, 167–168 emission, 110 modulation type, 208–218 AM signal, 208 automated radio signal, 219 FFSK signal, 211 FM signal, 211 PSK signal, 216 parameters, automatic measurement of, 379 processing, 343 propagation channel, 140 recording, 125, 449 superposition, 281 transition, 79 Radiotext, 167 Radio transparent radome, 340, 344, 458–459, 461 Radio wave incidence angle, 271 Radio wave propagation in forest-park zones, 412 mathematical model of, 409 Radio wave reflection, 307 Random direction-finding errors, 243–244 Rayleigh distribution, 118–120 Receiving and processing unit (RPU), 257 Red-black trees, 402 Reference-methodical documents (RMD), 471 Reference oscillator (RO), 56 Reference spatial signal (RSS), 278, 284 Refinement method, 278 Reflex receiver, 24 Regenerative and super-regenerative amplifiers, 24 Regional radio broadcast, 160 Remote radio monitoring systems, software for, 451–457 detection algorithms, 454 equipment operation, 455 radio microphone detection, 453 radio microphone localization, 455 SMO-DX application, 452 Repeater (repeater network), 171 RF amplifier (RFA), 23
506 Root-mean-square (RMS) error, 243 Root-mean-square (RMS), 36 RTTY protocol, 163, 233–235 Rubidium oscillator, 198 Russian Arm systems, 60–67 fifth-generation radio receivers, 66 first- and second-generation systems, 60 third and fourth generation, 62 Russian coaxial cables, 440 Russian minivan, 313 Russian Standard Agency, 482 S Satellite radio navigation systems (SRNS), 350 Satellite retransmitter-transponder, 154 Scanning generator, 311–312 Scanning receivers, 58, 60, 428 Scrambling, 170, 202, 437 Secondary power supply (SPS), 90 Selective micro-voltmeter, 28, 58, 93 Sensor array processing, 251 Shepard method, 417 Shift-keying characteristics, determination, 204 Signal amplitude, measurement, 294 Signal attenuation, 35–36, 40, 69, 89, 247, 409, 412, 440, 488–490, 494 Signal demodulation, 90, 206 Signal frequency estimation, 198 Signal generation unit (SGU), 291 Signal inverse shift, 33 Signal-missing probability, 105 Signal-noise ratio (SNR), 13, 37, 89, 458 Signal spectrum, measurement error of, 111 Signal transfer function variation, 52 Single-channel signal detection, 97–105, 108–117 ARM system parameters, 113 characteristics, 105 discrete Fourier transform (DFT), 97 frequency observation time, 109 registered frequencies, 112 separate frequency registration, 111 Single-channel vs. double-channel processing, 119–121 Single side band (SSB), 56, 142–143 Single sideband with suppressed carrier (SSBSC), 143 Sinusoidal carrier oscillation, 142 Sinusoidal modulating oscillation, 143, 146, 204 Sinusoidal signals, 47 Skin-effect, 361
Subject Index Slow frequency hopping (SFH), 176 SMO-KN application package, 373, 380, 387 SMO-MCRM software, 130–133 operation modes, 131 purpose and performance capabilities, 130 Smoothing window, 104 SMO-PPK (PA) software, 304, 372, 379, 407 SMO-PRIZ application, 485–487, 490–492 SMO-SECTOR program, 464 SMO-STA software, 206–208 SMO-TESTER application, 482–484, 492 SMO-THESIS, 492–496 Solar batteries, 9, 369, 372 Space frequencies, 151 Special mathematical software (SMS) program, 11 Spectral and bearing data (SBD), 374 Spectral-bearing data (SPD), 383, 388 Spectral frequency scale, 209 Spectral-power density, 200 Spectrum analyzer (SA), 28, 58, 93, 115, 196, 427–428 modification, character of, 202 transition, 25, 31, 72 Speech transmission method, 171 Spike antenna, 238–239, 241, 270 Spline-interpolation, 313, 407 Spot-jamming, 439–441, 443, 452 SPP protocols, 73 Spurious channel formation, 30 Star topology, 330, 333 Statistical theory of pattern recognition, 204 Stenographing, 131, 133 Sub-carrier Communication Allocation (SCA), 167 Superheterodyne receiver, 23, 25–27, 30, 58, 60, 70, 93 Super-regenerative receiver, 25 Suppression method, 70 Survey mode, 231–233 SXP format digital maps, 403 Synchronization connectors, 88 Synthesis theory of joint optimal algorithms, 96 System of active noisiness (SAN), 473, 490 System-wide car equipment, 343 T TCP/IP, see Transport Control Protocol/Internet Protocol T-DAB broadcasting, 168, 192 Technical channel of information leakage (TCIL)
Subject Index antenna system selection, 458 ARTIKUL-M6 mobile direction finder, 461 detection of sources by mobile station, 457–468 frequencies checking, 467 initial data frames, 464 RES detection, 459 software structure and search procedure, 462 suspicious frequency list, 465 Technical surveillance countermeasures (TSCM), 403, 427 Testing protocol, 487, 492, 495 Thermal-stable micro-power voltage regulators, 357 Thinned out antenna groups, 278 Third register, 163 Threshold displacement, 414 Time division multiple access (TDMA), 175 Time-frequency diagram (TFD), 384 Time multiplexing, 279, 290 TR-965SR radio, 328 Transport Control Protocol/Internet Protocol, 326–327, 331, 337, 372, 380, 453 Trunking (hitcher) systems, 171–172, 192, 374 analog, 173 digital, 173–174 Tuned radio receiver, 23–27 Tuner spectrum components, 32 TV channel frequencies, 190–191 U Ultra high frequency (UHF), 67, 86, 88, 143, 256
507 Universal mobile telecommunications system (UMTS), 178 V Very high frequency (VHF), 67, 89, 165–168, 170 Video signal spectrum, 170 Voice codec, 175 Voltage multiplication operation, 43 Voltage standing wave ratio (VSWR), 27, 29 W Walsh 64-bit sequence, 177 Watson-Watt direction finders, 239, 250 Wave impedance, 30, 239 Weak electromagnetic emissions, detection, 476 Weighting distance, 417 Wide-band signals, 42, 95, 126, 334 Wide frequency modulation (WFM) 146 Window functions (weighting functions), 198 Wireless local area network (WLAN), 180 Wireless metropolitan area network (WAN), 180 Wireless personal area network (WPAN), 182 World Administrative Radio Communication Conference (WARS), 136 World Technical Standard Communication Conference (WTSC), 136 Wullenweber direction finder, 239–240, 242 Z ZOOM FFT, 198
Continued from page ii
Digital Terrestrial Broadcasting Networks Beutler, Roland ISBN 978-0-387-09634-6, Vol. 23 Logic Synthesis for Compositional Microprogram Control Units Barkalov, Alexander, Titarenko, Larysa ISBN: 978-3-540-69283-6, Vol. 22 Sensors: Advancements in Modeling, Design Issues, Fabrication and Practical Applications Mukhopadhyay, Subhas Chandra; Huang, Yueh-Min (Eds.) ISBN: 978-3-540-69030-6 Smart Sensors and Sensing Technology Mukhopadhyay, Subhas Chandra; Sen Gupta, Gourab (Eds.) ISBN: 978-3-540-79589-6 Basic Principles of Fresnel Antenna Arrays Minin, Igor V., Minin, Oleg V. ISBN: 978-3-540-79558-2 Fundamental Numerical Methods for Electrical Engineering Rosloniec, Stanislaw ISBN: 978-3-540-79518-6 RFID Security and Privacy: Concepts, Protocols, and Architectures Henrici, Dirk ISBN: 978-3-540-79075-4 Advances in Mobile and Wireless Communications: Views of the 16th IST Mobile and Wireless Communication Summit Frigyes, István; Bito, Janos; Bakki, Péter (Eds.) ISBN: 978-3-540-79040-2 Informatics in Control Automation and Robotics: Selected Papers from the International Conference on Informatics in Control Automation and Robotics 2006 Andrade Cetto, J.; Ferrier, J.-L.; Pereira, J.M.C.D.; Filipe, J. (Eds.) ISBN: 978-3-540-79141-6 Bandwidth Extension of Speech Signals Iser, Bernd, Minker, Wolfgang, Schmidt, Gerhard ISBN: 978-0-387-68898-5 Proceedings of Light-Activated Tissue Regeneration and Therapy Conference Waynant, Ronald; Tata, Darrell B. (Eds.) ISBN: 978-0-387-71808-8
