Mobile Antenna Systems Handbook (Artech House Antennas and Propagation Library)

Mobile Antenna Systems Handbook Third Edition

For a listing of recent titles in the Artech House Antennas and Propagation Series, turn to the back of this book.

Mobile Antenna Systems Handbook Third Edition Kyohei Fujimoto Editor

Library of Congress Cataloging-in-Publication Data A catalog record for this book is available from the U.S. Library of Congress. British Library Cataloguing in Publication Data A catalogue record for this book is available from the British Library. ISBN-13: 978-1-59693-126-8 Cover design by Igor Valdman  2008 ARTECH HOUSE, INC. 685 Canton Street Norwood, MA 02062 All rights reserved. Printed and bound in the United States of America. No part of this book may be reproduced or utilized in any form or by any means, electronic or mechanical, including photocopying, recording, or by any information storage and retrieval system, without permission in writing from the publisher. All terms mentioned in this book that are known to be trademarks or service marks have been appropriately capitalized. Artech House cannot attest to the accuracy of this information. Use of a term in this book should not be regarded as affecting the validity of any trademark or service mark. 10 9 8 7 6 5 4 3 2 1

To the memory of Professor J. R. James, who unfortunately passed away at the planning stage of the third edition of Mobile Antenna Systems Handbook (MASH). Professor James greatly contributed to the first and second editions of MASH as a coeditor and one of the coauthors. Professor James is well known as a pioneer of printed and microstrip antennas. His death leaves us with a large scientific, technological, and personal void, but the rich heritage he has left is of inestimable benefit to us. His pioneering work stimulated the further study and development of printed antennas and microstrip antennas and encouraged the creation of various types of planar antennas, which are particularly significant in mobile systems where small, compact antennas are required.

Contents Preface to the Third Edition

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Chapter 1 Importance of Antennas in Mobile Systems and Recent Trends 1.1 Introduction 1.2 Trends 1.2.1 Mobile Systems 1.2.2 Increasing Information Flow 1.2.3 Propagation 1.3 Modern Mobile Antenna Design 1.4 Objectives of This Book References

1 1 9 13 15 15 15 19 22

Chapter 2 Essential Techniques in Mobile Antenna Systems Design 2.1 Mobile Communication Systems 2.1.1 Technologies in Mobile Communications 2.1.2 Frequencies Used in Mobile Systems 2.1.3 System Design and Antennas 2.2 Fundamentals in Land Mobile Propagation 2.2.1 Propagation Problems in Land Mobile Communications 2.2.2 Multipath Propagation Fundamentals 2.2.3 Classification of Multipath Propagation Models: NB, WB, and UWB 2.2.4 Spatio-Temporal Propagation Channel Model 2.2.5 Relation Between Space Correlation Characteristics and Space Diversity Effect 2.2.6 Propagation Modeling for OFDM 2.2.7 Propagation Studies for UWB References

25 25 25 31 33 34 34 36

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38 40 44 47 50 51

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Chapter 3 Advances in Mobile Propagation Prediction Methods 3.1 Introduction 3.2 Macrocells 3.2.1 Definition of Parameters 3.2.2 Empirical Path Loss Models 3.2.3 Physical Models 3.2.4 Comparison of Models 3.2.5 Computerized Planning Tools 3.2.6 Conclusions 3.3 Microcells 3.3.1 Dual-Slope Empirical Models 3.3.2 Physical Models 3.3.3 Nonline-of-Sight Models 3.3.4 Microcell Propagation Models: Discussion 3.3.5 Microcell Shadowing 3.3.6 Conclusions 3.4 Picocells 3.4.1 Empirical Models of Propagation Within Buildings 3.4.2 Empirical Models of Propagation into Buildings 3.4.3 Physical Models of Indoor Propagation 3.4.4 Constitutive Parameters for Physical Models 3.4.5 Propagation in Picocells: Discussion 3.4.6 Multipath Effects 3.4.7 Conclusions 3.5 Megacells 3.5.1 Shadowing and Fast Fading 3.5.2 Local Shadowing Effects 3.5.3 Empirical Narrowband Models 3.5.4 Statistical Models 3.5.5 Physical-Statistical Models for Built-Up Areas 3.5.6 Wideband Models 3.5.7 Multisatellite Correlations 3.5.8 Overall Mobile-Satellite Channel Model 3.6 The Future 3.6.1 Intelligent Antennas 3.6.2 Multidimensional Channel Models 3.6.3 High-Resolution Data 3.6.4 Analytical Formulations 3.6.5 Physical-Statistical Channel Modeling 3.6.6 Real-Time Channel Predictions 3.6.7 Overall References

55 55 55 57 58 65 76 76 77 78 79 81 86 92 93 93 93 94 97 101 105 105 106 108 108 110 111 113 115 122 131 131 133 134 134 135 135 135 136 136 136 137

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Chapter 4 Antennas for Base Stations 4.1 Basic Techniques for Base Station Antennas 4.1.1 System Requirements 4.1.2 Types of Antennas 4.1.3 Radio Zone Design 4.1.4 Diversity 4.2 Design and Practice of Japanese Systems 4.2.1 Multiband Antennas 4.2.2 Remote Beam Tilting System 4.2.3 Antennas for Radio Blind Areas 4.2.4 Antennas for CDMA Systems 4.3 Adaptive Antenna Systems 4.3.1 Personal Handy Phone System 4.3.2 W-OAM 4.3.3 i-Burst System 4.3.4 Experimental System of Adaptive Array for WCDMA 4.3.5 Experimental System of Adaptive Array for CDMA2000 1xEV-DO 4.4 Design and Practice II (European Systems) 4.4.1 Antenna Configurations 4.4.2 Antenna Solutions 4.4.3 Antenna Units 4.4.4 Antenna Development Trends References

141 141 141 143 144 146 151 151 157 158 164 170 170 172 173 175

Chapter 5 Antennas for Mobile Terminals 5.1 Basic Techniques for Mobile Terminal Antennas 5.1.1 General 5.1.2 Brief Historical Review of Design Concept 5.1.3 Modern Antenna Technology 5.2 Design and Practice of Antennas for Handsets I 5.2.1 Some Fundamental Issues 5.2.2 Various Multiband Antenna Concepts 5.2.3 Antenna Integration and Some Practical Issues 5.2.4 The Multichannel Antenna Applications 5.2.5 Human Body Interaction with Terminal Antennas and Some Measurement Methods 5.3 Design and Practice of Antennas for Handsets 5.3.1 Multiband and Broad Band Antenna Technologies 5.3.2 Diversity Antenna Technologies 5.3.3 Antenna Technologies Mitigating Human Body Effect 5.3.4 Antenna Technologies for Reducing SAR

213 213 213 215 217 219 220 226 239 245

176 177 179 187 195 203 208

257 266 268 274 287 298

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5.3.5 Technique of Omitting Balun 5.3.6 Technology of Downsizing PIFA 5.4 Evaluation of Antenna Performance 5.4.1 Measurement Method Using Optical Fiber References Radio Frequency Exposure and Compliance Standards for Mobile Communication Devices 6.1 Introduction 6.2 Physical Parameters 6.3 Types of RF Safety Standards 6.4 Exposure Standards 6.4.1 ICNIRP 6.4.2 IEEE C95.1-2005 6.4.3 Similarities and Differences Between the 1998 ICNIRP Guidelines and IEEE C95.1-2005 6.4.4 Regulations Based on Older Standards 6.5 Compliance Standards 6.5.1 Main Features of IEEE 1528-2003 (Including 1528a-2005) and IEC 62209-1 6.5.2 Other Standards Related to Mobile Communication 6.6 Discussion and Conclusions References

304 307 309 309 313

Chapter 6

330 330 333 333 339 339 341

Chapter 7 7.1 7.2

7.3 7.4

7.5

Applications of Modern EM Computational Techniques: Antennas and Humans in Personal Communications Introduction Definition of Design Parameters for Handset Antennas 7.2.1 Absorbed Power and Specific Absorption Rate 7.2.2 Directivity and Gain 7.2.3 Antenna Impedance and S 11 Finite-Difference Time-Domain Formulation Eigenfunction Expansion Method 7.4.1 EEM Implementation 7.4.2 Hybridization of the EEM and MoM Results Using EEM 7.5.1 Human Head Model 7.5.2 EM Interaction Characterizations 7.5.3 Effects of Size of the Head Model: Adult and Child 7.5.4 Comparison Between Homogeneous and Multilayered Spheres 7.5.5 Vertical Location of Antennas 7.5.6 Comparison with EEM and FDTD

321 322 322 323 325 326 328

343 343 347 347 348 348 349 351 351 352 353 353 354 358 360 361 364

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7.5.7 Anatomical Head Versus Spherical Head 7.5.8 Directional Antennas 7.5.9 High-Frequency Effect 7.6 Results Using the FDTD Method 7.6.1 Tissue Models 7.6.2 Input Impedance and the Importance of the Hand Position 7.6.3 Gain Patterns 7.6.4 Near Fields and SAR 7.7 Assessment of Dual-Antenna Handset Diversity Performance 7.7.1 Dual-Antenna Handset Geometries 7.7.2 Simulated Assessment of Diversity Performance 7.7.3 Experimental Assessment of Diversity Performance 7.7.4 Results References

368 370 372 376 376 378 383 384 389 390 390 392 394 396

Chapter 8 Digital TV Antennas for Land Vehicles 8.1 Reception Systems 8.1.1 Digital Television Services in Japan 8.1.2 Problems of Mobile Reception 8.1.3 Diversity Reception Methods 8.1.4 Demonstration 8.2 Digital Television Antennas 8.2.1 Quarter Glass Antenna for a Van 8.2.2 Thin Antenna 8.2.3 Omnidirectional Pattern Synthesis Technique for a Car 8.2.4 Antennas Currently on the Market References

399 399 399 400 400 402 405 405 407 408 410 415

Chapter 9.1 9.2 9.3

417 417 418 419 419 421 425

Chapter 10 Antennas for ITS 10.1 General 10.2 Antenna Design 10.2.1 Communication Beam Coverage 10.2.2 Antenna Fundamental Design

427 427 429 429 431

9 Antennas for the Bullet Train Introduction Train Radio Communication Systems Antenna Systems 9.3.1 LCX Cable 9.3.2 Train Antenna References

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10.2.3 Microstrip Antenna Design 10.2.4 Communication Coverage 10.2.5 Multiple Reflections 10.3 Field Strength in Communication Area 10.3.1 Multiple Reflections from Canopies 10.3.2 Mitigation Using an Absorber at the ETC Gate 10.3.3 Propagation in DSRC Coverage 10.3.4 Data Rate of DSRC 10.4 Antennas for DSRC 10.5 Applications for DSRC References Chapter 11 Antennas for Mobile Satellite Systems 11.1 Introduction 11.2 System Requirements for Vehicle Antennas 11.2.1 Mechanical Characteristics 11.2.2 Electrical Characteristics 11.2.3 Propagation Problems 11.3 Omnidirectional Antennas for Mobile Satellite Communications 11.3.1 Overview 11.3.2 Quadrifilar Helical Antenna 11.3.3 Crossed-Drooping Dipole Antenna 11.3.4 Patch Antenna 11.4 Directional Antennas for Mobile Satellite Communications 11.4.1 Antennas for INMARSAT 11.4.2 Directional Antennas in the ETS-V Program 11.4.3 Airborne Phased Array Antenna in the Domestic Satellite Phone Program 11.4.4 Directional Antennas in the MSAT Program 11.4.5 Directional Antennas in the Ku-Band CBB Program 11.5 Antenna Systems for GPS 11.5.1 General Requirements for GPS Antennas 11.5.2 Quadrifilar Helical Antennas 11.5.3 Microstrip Antennas 11.6 Multiband Antennas for Future GPS/ITS Services 11.6.1 Slot Ring Multiband Antenna for Future Dual Bands (L1 , L2 ) GPS 11.6.2 Microstrip Multiband Antennas for GPS, VICS, and DSRC 11.7 Satellite Constellation Systems and Antenna Requirements 11.7.1 Constellation Systems and Demands on Antenna Design 11.7.2 Handset Antennas for Satellite Systems References

435 441 442 443 443 444 448 450 453 453 457 459 459 461 461 461 465 467 467 467 468 469 470 470 481 489 490 495 498 498 502 504 507 507 517 523 523 526 538

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Chapter 12 UWB Antennas 12.1 UWB Systems: Introduction 12.2 Requirements for UWB Antennas 12.2.1 Basic Principle of UWB Antennas 12.2.2 Modeling and Structure of Feeding Points 12.2.3 Current Distributions of Circular Disc Monopole Antenna 12.3 Characteristics of Popular UWB Antennas 12.3.1 Three-Dimensional UWB Antennas 12.3.2 Planar UWB Antennas 12.3.3 CPW Feed 12.3.4 Multilayer Technologies 12.3.5 Band-Rejection for Coexistence with Other Wireless Systems 12.4 Wire-Structured UWB Antennas and Wire-Grid Modeling Simulation 12.4.1 High Efficiency Moment Method 12.5 UWB Antennas in Specific Wireless Environments 12.5.1 UWB Antennas Used in Unlicensed and Autonomous Wireless Environments 12.5.2 Measurements of Multipath Propagation Environments for UWB Antennas 12.5.3 Transmission Characteristics of UWB Antennas and Effects of the Human Body 12.5.4 UWB Antennas Near the Human Body 12.6 UWB Antenna Evaluation Indexes 12.7 UWB Antenna Measurements 12.7.1 Radiation Pattern Measurements 12.7.2 Impedance Measurements 12.7.3 Scale Model Measurements 12.7.4 Impedance Measurements with Two Coaxial Cables 12.8 Integrated Antenna Design Approach Based on LSI Technology 12.9 Radio Wave Resource Sharing with Technology Leadership and the Role of the Antenna References Chapter 13 Antennas for RFID 13.1 The Characteristics of an RFID System 13.1.1 What Is RFID? 13.1.2 Operating Frequencies 13.1.3 Operating Principles 13.1.4 Read Range 13.2 Reader Antennas 13.2.1 Fixed Reader 13.2.2 Mobile Reader

543 543 544 544 545 549 551 552 555 557 561 562 565 565 567 567 568 569 574 576 577 577 578 579 580 583 583 584 589 589 589 591 592 595 596 596 599

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13.3

Tag Antennas 13.3.1 Structure of a Tag Antenna 13.3.2 Impedance Matching 13.3.3 Tags on Metallic Surface 13.3.4 Bandwidth-Enhanced Tag Antennas 13.3.5 SAW Tags 13.4 Measurement of Tag Antennas 13.4.1 Measurement of the Tag Antenna Impedance 13.4.2 Read Range Measurement 13.4.3 Efficiency Measurement References

605 605 607 609 611 612 612 613 614 615 616

Chapter 14 Multiple-Input Multiple-Output (MIMO) Systems 14.1 Introduction 14.2 Diversity in Wireless Communications 14.2.1 Time Diversity 14.2.2 Frequency Diversity 14.2.3 Space Diversity 14.3 Multiantenna Systems 14.4 MIMO Systems 14.5 Channel Capacity of the MIMO Systems 14.6 Channel Known at the Transmitter 14.6.1 Water-Filling Algorithm 14.7 Channel Unknown at the Transmitter 14.7.1 Alamouti Scheme 14.8 Diversity-Multiplexing Trade-Off 14.9 MIMO Under an Electromagnetic Viewpoint 14.9.1 Case Study 1 14.9.2 Case Study 2 14.9.3 Case Study 3 14.9.4 Case Study 4 14.9.5 Case Study 5 14.10 Conclusions References

619 619 620 620 621 622 623 624 627 628 629 629 630 631 632 634 635 635 639 641 643 644

Chapter 15.1 15.2 15.3 15.4

647 647 649 650 652

15 Smart Antennas Definition Why Smart Antennas? Introduction Background

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15.5

Beam Forming 15.5.1 Minimum Mean Square Error 15.5.2 Minimum Variance Distortionless Response 15.6 Direct Data Domain Least Squares (D3LS) Approaches to Adaptive Processing Based on a Single Snapshot of Data 15.6.1 Eigenvalue Method 15.6.2 Forward Method 15.6.3 Backward Method 15.6.4 Forward-Backward Method 15.7 Simulations 15.8 Conclusion References

653 655 656 659 662 663 665 666 667 671 671

Appendix A Glossary A.1 Catalog of Antenna Types A.1.1 Linear Antennas A.1.2 Material Loading A.1.3 Planar Antenna A.1.4 Broadband and Multiband Antennas A.1.5 Balance-Unbalance Transforming A.1.6 Arrays and Diversity Systems A.1.7 Recent Innovative Concepts References A.1.8 Key to Symbols and Acronyms Used in Sections A.2 to A.3 A.2 Land Mobile Systems A.2.1 Automobiles A.2.2 Portable Equipment A.2.3 Trains A.2.4 Base Stations A.2.5 Satellite Systems A.2.6 UWB A.2.7 RFID A.3 Typical Antenna Types and Their Applications

675 675 676 678 679 680 681 681 682 682 703 704 704 711 718 719 723 727 729 732

Acronyms and Abbreviations

735

List of Contributors

739

Index

747

Preface to the Third Edition

It was a profound shock when I suddenly received an e-mail from Professor J. R. James’ son-in-law, informing me that Professor James had passed away on July 24, 2006. I had never imagined receiving such a sad e-mail at that time, as we, Professor James and I, were exchanging discussions about the planning of the third edition of our Mobile Antenna Systems Handbook (MASH). I had known he was suffering from a serious disease since December 2005; however, I was reassured somehow by receiving an e-mail from Professor James in February 2006, telling that he had undergone a successful operation and returned home from the hospital for rehabilitation. But his health condition was not so enduring, and our e-mail exchange was interrupted for a while. When I was about to send a draft outline of the book to him, the sad news came. It was a great grief that I lost my coeditor Professor James while we were planning the next edition of MASH. It was not so hard to imagine that Professor James himself regretted that he left many things unfinished, including this book, before untimely end of his life. We, the editor and the editorial staff of the publisher, continued the book project, and now are very pleased to have the third edition of MASH published. The first edition of MASH was published in 1994. The book aimed not only to describe up-to-date antenna technology, but also to provide useful data for design and to demonstrate practical antennas so that the book could assist the design and development of mobile antennas. The book was at that time recognized as the first book that provided a comprehensive treatment of antennas pertaining to mobile communications. Fortunately, the worldwide response to MASH has indeed been gratifying and the handbook was appreciated across a wide spectrum of people, who have felt the need for enlightenment and a broader perspective on some or all aspects of mobile antennas. This can be understood by noticing the remarkable progress of mobile communications in the past decade. Mobile communications have emerged as a dominant influence on antenna design and development, and the handbook was conceived to present these important developments. Almost xvii

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no one expected that the speed of change in mobile communications would be so fast and its influence on antenna design would be so profound. This situation inspired the necessity of revising MASH, and the second edition was published in 2000. The second edition extended the topics to include new materials for antennas used in advanced mobile satellite systems and mobile phone systems, which have evolved to employ downsized base stations and smaller but functional mobile terminals, while retaining as much of the previous fundamental materials as size of the book allowed. Two new chapters were added in the second edition, one covering the precise prediction method of propagation in smaller mobile cell environments, and another treating the application of computer modeling and measurement of EM interaction between the handsets and nearby human body. Since the early years of the 2000s, we have observed further notable changes in mobile systems—one is evolutions in mobile phones technology, and another is the deployment of new wireless mobile systems. With the deployment of the 3G systems in addition to the 2G systems, mobile phones operating in areas where both 2 G and 3G services are provided should have multiband antennas and yet smaller dimensions. Reductions in size of the cell areas have accelerated the downsizing of antennas and the applications of planar structures to base station antennas. Requirements for small mobile phone antennas are not only multiband performance, but also, in most cases, sizes small enough to be contained within the small mobile terminal unit. Mobile terminals have gradually evolved to incorporate functions such as cameras, Internet linking, including e-mail and various contents reception, TV reception while in motion, and so forth. Thus, they should no longer be recognized as only telephone terminals, but also as information terminals, which perform a large variety of functions, including high-speed data transmission. Some typical new mobile systems appeared recently are WLAN, wireless mobile broadband systems, and Near-Field Communication (NFC) Systems, including RFID (radio frequency identification), UWB (ultrawideband), and Bluetooth systems. Antennas used for these systems are not particular ones, but those designed to fit the systems adequately. They are generally small antennas, so they need specific design techniques based on the small antenna concept. In addition, antennas and communications engineers pay intensive attention to MIMO (multi input multioutput) systems, because of their superb performance in high data rate transmission. Considering these situations, we, Professor James and I, started discussions about the renewal of the second edition of MASH, but it was in vain, because of his unfortunate passing. However, every effort to publish the third edition continued. The contents have been updated by adding many new materials, and new chapters are arranged for UWB (Chapter 12), RFID (Chapter 3), smart antennas (Chapter 14), and MIMO (Chapter 15). Almost 70% of the contents in the second edition were revised, leaving only two chapters, Chapters 3 and 7, as they are considered still useful, although about half of the old Chapter 7 was deleted or partly moved to the new Chapter 5. In Chapter 2, discussions on the

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propagation problems in digital systems have been added. Chapter 4, describing base station antennas, is divided into two parts: one dealing with European systems, and another covering Japanese systems. Modern base station antennas are required to be downsized, to operate in multiple bands, and to have beam-shaping functions. Some systems even employ adaptive control performance. The contents of Chapter 5, which deals with mobile terminal antennas, have expanded to include new antennas, which require multiband operation, and yet having a small size so that they can be contained inside the small handset unit. Some newly developed antennas for terrestrial digital-TV reception are introduced in Chapter 8. Chapter 9 is a short chapter, in which recently introduced antennas for Shin-kansen (Bullet train in Japan) are added. Chapter 10 describes Intelligent Transportation Systems (ITS) with some revision from the second edition, including upto-date antennas used for ETC (electronic toll collection) systems and mobile terminals defined as part of the DSRC (dedicated short range communication) systems. Chapter 11 introduces some recent mobile satellite systems with discussion about progress in mobile terminals, particularly transportable ones, and the introduction of small antennas and antenna arrays. There are some size unbalances among the chapters. However, it does not mean that small chapters, for instance, Chapter 9, are less significant, but it should be understood that the systems treated are rather specific so that their description is condensed to a shorter one. The glossary in this edition has been revised extensively to include many new materials and should provide useful information that will help the design and development of antennas for future mobile systems as well as present systems. Readership of this edition of MASH remains the same as projected in the previous editions. Once again, I sincerely appreciate everyone who generously cooperated and excellently contributed to this handbook, and most especially I wish to express my sincere condolences to the family of my former coeditor and one of my best friends, the late Professor J. R. James. ACKNOWLEDGMENTS This third edition of the Mobile Antenna Systems Handbook (MASH) was published with great support by many persons and organizations. First, I, as the editor of this handbook, would like to extend sincere thanks to all of the authors who made valuable contributions to the handbook with their highly technical skills and experiences. I am also most grateful to the reviewers for their patient and skillful advice. Sincere thanks are also due to the previous authors of the first and second editions, who contributed to establish MASH as the useful book it became. Every author’s elaboration and efforts to complete the manuscripts were enormous, and it is not hard to conjecture that they must have sacrificed their busy time to write their chapters.

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We, the editor and the authors, owe many persons and organizations for their generous consideration to provide us with very precious materials and useful information for the handbook. Appreciation is owed to the editorial and production staff of Artech House, who have done excellent work to complete this third edition. They also helped me greatly in my editing work. I also acknowledge Dr. Julie Lancashire, a former commissioning editor at Artech House, who occasionally gave me advice and suggestion for my editorial work. My many thanks go to Dr. J. R. Copeland, who greatly helped my editorial work with his skill and experience pertaining to grammar and wording. In particular, he energetically checked and brushed up nonnative authors’ English, from a technical point of view as an experienced antenna researcher. (Dr. Copeland is one of the researchers who did pioneering work on the integrated antenna array for the pattern synthesis, which was developed in 1962 at the Ohio State University.) Credit must also be given to my former colleague, Professor Hiroyoshi Yamada of Niigata University, Japan, and his students. They helped me in computer work for producing figures, tables, and manuscripts. Finally, special recognition is hereby expressed to my wife Machiko Fujimoto for her constant understanding and consideration in giving up so many days of our family time. Kyohei Fujimoto Editor Fujisawa, Japan June 2008

Chapter 1 Importance of Antennas in Mobile Systems and Recent Trends Kyohei Fujimoto

1.1 INTRODUCTION The first mobile communication system was initiated in 1885 with wireless telegraph between trains and stations, and was developed by Thomas Edison [1]. Telegraph signals were conveyed through the trolley wires, which were electrostatically coupled with a metal plate installed on the ceiling of the train. Edison also experimented with communication on a vehicle in 1901 [2], using a thick cylindrical antenna placed on the roof of the vehicle. Real mobile communication services started with wireless telegraph on ships, developed in 1898 by Guglielmo Marconi, using long vertical wire antennas in various forms such as T, inverted L, and umbrella shapes. Portable equipment appeared in 1910 [3]. Both World Wars I and II provided the need for advanced antenna design and the surge of technology progress [4]: wire antennas were firmly established in the 1920s, while present-day microwave antenna design and technology were commonplace in the 1950s. In the 1960s a new antenna era emerged because of the revolutionary progress in semiconductor integrated circuits, attributed initially to the Cold War defense industry but substantially carried forward into the commercial equipment sector. Quite simply, the demand opened up designers to the possibilities of redesign, recreation, and transformation of known antenna types into less bulky, lightweight, low-cost, easy-to-manufacture radiating structures, compatible with the newly conceived integrated electronic packages. Most notable has been the creation of the printed antenna technology, which lends itself to multifunction antenna devices [5]. Planar antennas were originated from printed antennas 1

2

and have been used in various mobile systems, both in base station and mobile terminals, where small, compact, lightweight antennas are required. Some of the salient factors that have increasingly influenced antenna design in this era [6] and continue to do so today are noted in Table 1.1, which emphasizes that communications, particularly mobile communication systems, are acting as the most significant driver of antenna technology today. In addition, deployment of new wireless mobile systems and connection of mobile systems to the IP network are other important factors that significantly influence development of a variety of novel antenna systems. Many other information-relaying systems are now emerging that have much in common with mobile systems. Here ‘‘mobile systems’’ imply not only those pertaining to the communication systems, but also systems for control, sensing, identification, and so forth, including wireless mobile systems other than mobile phone systems such as wireless access systems, near field communications (NFC) systems, radio identification Table 1.1 Factors Influencing Recent Antenna Technology and Design Factor Spectral congestion and utilization Remarkable growth in mobile/ personal communication systems Increase in high rate information and data transmission Deployment of new wireless mobile systems Growth in mobile satellite communications Link with IP network and optical fiber networks Intelligent traffic information, control, safety and management systems Development and application of new materials Development of precise computer modeling for analysis, design, and measurement Public awareness of electromagnetic radiation and safety Increase in mobile phone functions development of wearable communications

Trends Wider bandwidth operation, improved performance, interference rejection, use of millimeter and submillimeter antennas New small compact higher performing antennas for cellular and other mobile wireless systems ‘‘Smart’’ antennas, adaptive antennas, and MIMO systems Appropriate design of antennas for each system Higher performance space-borne antennas offering multifunction operation, and small high performance mobile terminal antennas Small high-performance antennas for mobile terminals Specifically designed antennas for both vehicles and roadside infrastructures Redesign of existing, and creation of new, small functional, highperformance antennas, application of metamaterials to antenna structure Strengthens design methods to create higher performing antennas, development of precise EM simulation method to deal with small, complicated antenna structure Preference for lower transmitted power antenna and reduction of EMI and SAR level Specifically designed antennas for each purpose Investigation of signal transmission through human body and development of implant antenna

3

(RFID) systems, and others. Table 1.2 describes some applications of mobile systems in various fields and requirements for antennas used in these systems. These systems generally have the massive customer base and resourcing associated with mobile systems, and they demand much ingenuity from the antenna designers. This brief historical perspective indicates how the creation and development of antennas have accelerated rapidly in response to worldwide demands that are raised by the growth and deployment of newly developed wireless mobile systems. The worldwide impact of mobile communications and related systems on antennas during the past decade is significant in many ways. • •

•

It has arisen mainly from the commercial sector. The period of accelerated antenna design activity is already at a record length, showing no sign of diminishing, and is associated with massive and increasing resourcing by the public. The infusion of mobile communications into all remote corners of the world community has brought about both sociological changes [7] and an increased public awareness of antennas in their everyday environment. The latter awareness of electro-

Table 1.2 Some Applications Related to Mobile Communications Demanding Innovative Antenna Design Application Animal tracking Product information and display explanation Broadcasting reception Traffic information, control, safety and management systems Security systems Environmental monitoring and navigation

Information and data transmission system at home/office E-commerce Entertainment Route guidance

Requirements Inconspicuous robust body-mounted antennas for satellite or terrestrial monitoring Electrically small antenna for IC card RF tag, barcode, etc., in near field data transmission systems Particularly designed antenna for AM, FM, or TV reception Vehicle-mounted and roadside-installed antenna for dedicated short range communications Inconspicuous antennas for smart cards, door operation, personal identification, personal key Remote recording of terrestrial environment and weather data Inconspicuous small antennas for both Global Positioning Satellite (GPS) handsets and vehicle-mounted terrestrial navigation systems Short-range communication (data and video) and control systems Small antenna mounted on the handset unit for banking, ticketing, e-commercing Small antenna mounted on a small portable unit for game play, movie and music delivery services reception Inconspicuous antenna in guidance systems for aged persons and weak-eye-sighted persons

4

•

•

magnetic radiation has already influenced base station and handset antenna design specifications. Deployment of mobile wireless systems has accelerated the rising data rate of signal transmission while in moving status as well as the need for antennas, which can deal with high data rate transmission systems. Meanwhile, very short range wireless systems, which generally deal with control, sensing, identification, and so forth, other than communications, have emerged into public environments. They require small antennas, dimensions of which depend on the requirements for systems to be installed. In some systems, very small electrically small antennas (ESA) are used.

It is appreciated that future forecasts are seldom very accurate, and the influence of mobile systems on current antenna designs was certainly not foreseen. As such, the authors of this work will not attempt predictions far into the future; they will instead examine the many facets of contemporary mobile systems and how they in turn influence current antenna design and manufacture. There is much that can still be learned from the immediate views and aspirations of this new vibrant communications industry as a whole. This has been a decade when the sales of mobile communications equipment have increased explosively and outstripped prediction, as the cellular handsets received overwhelming acceptance as an important personal necessity for social activity, daily life, enjoying entertainment, and so forth. However, the market for new subscribers of mobile phone systems has nearly reached saturation in some areas, particularly where mobile services were first introduced in the 1980s and the 1990s. Presently, the number of mobile phone subscribers continues to increase, to a large part because of the rapid increase of mobile phone systems in developing countries such as China, India, and some countries in Africa. Figure 1.1 shows the trend in number of cellular system subscribers in the world until 2005, and after 2005 with an extended line based on the prediction.

Figure 1.1 Trend in increase in number of mobile phone subscribers.

5

An outline of some of the milestones, both achieved and planned, in relation to the evolution of global mobile systems is given in Figure 1.2. The early analog systems in the 1980s—now referred to as ‘‘first generation (1G)’’ systems—laid the foundation for the subscriber market by demonstrating the benefit of mobile communications. In comparison with present-day equipment, the early systems were relatively simple, but large, and were developed to suit local requirements. The second generation (2G) digital systems deployed in early 1990s have shown not only the advantages of digital over analog processing with respect to increasing the available capacity and services provided, but also the way forward to achieve some global standardization. There was much discussion of the Global Communication Village concept comprised of large-scale terrestrial megacells, which are resolved into nested geometrical regions of macro, micro, pico, and femto cells, which are pertinent to in-building communication. Such aspirations led to the concept of complete network configurations of integrated mobile communication systems. The very successful Global System for Mobile Communications (GSM) has been instrumental in this and has paved the way for the third generation (3G) European Universal Mobile Telecommunication Systems (UMTS), which occupy bands up to 2.2 GHz. The global standardization was, after all, settled to specify five systems

Figure 1.2 Evolution of mobile communication systems showing some widely used systems and future concept.

6

by the International Telecommunication Union Radiocommunication Sector (ITU-R). Two of these systems, CDMA2000 and wideband code division multiple access (WCDMA), referred to as the 3G systems, have so far been implemented and operated worldwide. It was proposed [8] that global integration would take place under the International Mobile Telecommunications–2000 (IMT-2000) standards. The anticipated lower cost per bit capacity demands for multimedia, including the Internet, could be realized by: • • • •

Employing the CDMA technology [9, 10]; Signal compression techniques particularly concentrating on speech coding [11] and video coding; ‘‘Smart’’ adaptive antenna array methodologies [12]; Advanced VLSI technology, by which functional circuits can be packaged into a tiny chip with high density, precisely processed circuit structure.

A more quantifiable understanding of the propagation phenomena taking place in various propagation environments has also contributed to constructing modern digital mobile systems. Attention should also be paid to the significance of wireless mobile systems, which include systems that operate in very short range and function as other than voice communications. An example is the wireless local area network (WLAN), which replaces the wired LAN with the radio linked LAN. Without wire connection, the radio link connects computers to other computers through access points, and even voice communication is possible, if it is required. The wireless systems are classified by their communication range: • • • •

Wireless personal area network (WPAN) for the very short range of less than 10m or so; WLAN for up to 100-m range; Wireless metropolitan area network (WMAN) for the long or wide ranges, like in city areas, with the range of 10 km or so; Wireless wide area network (WWAN) for the suburban areas, with the range of up to 50 km.

Antennas for these systems differ depending on the individual system requirements. For WPAN systems, antennas are generally simple and small in size. For long range systems, antennas used are generally ordinary ones employed in the conventional communication systems. If systems need high quality and high data rate transmission, complicated or sophisticated antennas like smart array and multi-input multioutput (MIMO) systems are taken as the appropriate candidates. Another significant mobile system is the Intelligent Transport System (ITS), which is an advanced transportation system aimed at the realization of safety, smoothness, and economy in vehicular traffic, the preservation of environmental ecology, and the

7

improvement of traffic and road management. The core systems of ITS are communications and computer systems. The communication system in ITS is referred to as the dedicated short range communication (DSRC) system, which is classified in two ways: communication between vehicles and roadside (CVR) and communication of vehicles to vehicles (CVV). The DSRC, as applied to electronic toll collection (ETC) systems, is the toll gate system, where the toll collection is automatically processed when a vehicle passes through the gate, so that the vehicle does not need to stop at the gate. The toll collection is done electronically by the communication system between vehicles and the toll-gate facility. The DSRC system uses 5.8 GHz and amplitude shift keying (ASK) or phase shift keying (PSK) in the data transmission system. At the gate, a phased array is installed which produces an elliptical spot area on the road, where vehicles communicate with the ETC facilities. Antennas for vehicles are rather simple ones such as small microstrip antennas, or planar or printed antennas, which are usually placed on the dashboard in the car. Like many emerging areas in electronics, the new equipment concepts have been made possible by revolutionary semiconductor chip products that are now easily exploitable (the electronics packed into mobile phone handsets are indeed an impressive sight). The latest trend is application of VLSI chips to functional circuits in mobile phones. This brings enhancement of functions without increasing appreciably the volume of the mobile phone unit. In turn, it influences antenna design. The smaller the mobile phone unit becomes, the greater will be the influence of the components located near the antenna on the antenna performance. Typically such materials are a speaker, a camera, a liquid crystal display (LCD), and other electronic components. The antenna design, then, should adopt a concept of integration in which surrounding hardware and electronic components are included into the antenna structure. A battery is one of the essential components in mobile phones, since the capacity determines the transmitting power and the operating time between charging, which is desired to be as long as possible. It should have higher efficiency so that the transmitting power can be kept as high as possible and users can use the mobile phone for a longer time without frequent charging. It should also be small, compact, and lightweight. In turn, if the antenna gain can be made higher, the battery capacity can be made smaller, because the transmitting power is allowed to be lower. Similar to other components, the battery should also be included in the antenna design, as the antenna current may flow on the surface of the battery unit. The challenge for the mobile antenna designer goes further because there is now an awareness that clever design of the antenna can give added value by embodying additional system function such as multiband operation, diversity in both transmitting and reception, reduction of multipath fading and interference, or adaptive control to the environmental conditions. The most serious problem in the antenna design is to make the antenna structure small and compact enough to be contained within the small mobile phone unit. Mobile antenna design is no longer confined to small, lightweight, omnidirectional radiators on a well-defined flat ground plane, but is rather the creation of a sophisticated electromagnetic configuration that plays a significant role in signal processing, while

8

operating in a generally ill-defined time varying environment. The antenna should be recognized as an integral part of the overall system as described pictorially in Figure 1.3. The nature of the mobile system itself greatly influences the ultimate antenna design, and several distinctions can be made between land, maritime, aeronautical, and satellite mobile systems, as well as the type of mobile platforms such as vehicles, ships, aircraft, and portable equipment. Such factors as frequency effective use techniques, the type of information, modulation/demodulation schemes, data rate of signal transmission, and functions of mobile terminals are of serious concerns to antenna designers. The influence of a handset operator and the human hazard problems, which may arise from radiation of the handset transmitter, are another serious problem for antenna designers. The practical measure to evaluate the human hazard is the specific absorption rate (SAR) value, and it should be kept as small as possible, especially against human brains. Various trials to reduce the SAR value against the human body have been undertaken so far. One way is to constitute an antenna system in a balanced structure and feed it with a balanced line. By this means, current flows on the ground plane inside the handset unit can be significantly reduced, and as a consequence, the variation of the currents due to the human body will

Figure 1.3 Antenna system as an integral part of mobile system.

9

be made very small so that the antenna performance can be kept within only slight variation. The simplest way is to place antenna element on the mobile terminal as far apart from the human head as possible. Location of the antenna is restricted in the limited volume inside the handset unit, especially for a built-in antenna. Table 1.3 summarizes this requirement and its implication on antenna design. 1.2 TRENDS Mobile systems are presently being advanced toward fourth generation (4G) systems. There are five major trends in modern mobile systems: 1. 2. 3. 4. 5.

Progress of personalization; Advancement of globalization; Increase of multimedia services; Deployment of multidimensional network; Sophistication of mobile systems by implementing software processing.

The typical trends in modern mobile systems are listed in Figure 1.4, in which related demands and antenna structure are illustrated, and these are discussed in the following sections. Table 1.3 Mobile Antenna System Design Requirement Requirement Antenna as a system Designed to accommodate propagation effects Transmission of high data rate signals Compatible with environmental conditions Downsizing Integration of antenna with platform and nearby materials Latest manufacturing technology User-friendly and reliable performance EMC constraints Multimedia applications

Implication Not as an isolated receive/transmit terminal Diversity, rake reception, and introduction of adaptive antenna Smart antennas, MIMO systems, and antennas for SDMA Pattern characteristics to match zone requirements and reduction of environmental effects, including human body effect Small size, compact, light weight, built-in structure, and integrated antenna To include proximity effects in antenna design Exploitation of new composite materials, integrated electronic technology, and high density compact circuit structure Minimum complexity in human interface; high reliability of mechanical design Reduction of EMI and SAR level Enhancement of antenna performance; wideband, multiband, MIMO system

10

Figure 1.4 Trend in mobile communication and antenna structure.

Personalization Remarkable personalization has been seen in recent mobile terminals. This has been spurred not only by equipment downsizing, but also by the enhancement of functions of mobile terminals, especially of mobile phone systems. Modern mobile phones are equipped with functions to obtain various content, including functions of personal entertainment such as games, movies, TV broadcasting, and music, in addition to personal telephone use. Some mobile phones have the capability of ticketing, banking, navigation with GPS, e-mailing, and connection to the Internet for receiving information services. These mobile phones should be recognized as being no longer merely ‘‘telephones,’’ but as sophisticated information terminals. The downsizing of mobile terminals has also given impetus to the personalization of mobile systems because the smaller terminals are more convenient to carry and easier to operate. There was a time when mobile phone manufacturers were competing on downsizing dimensions and reducing weight and volume of mobile phones. Downsizing gave rise to severe problems for antenna designers: the requirement of smaller antennas for downsized terminals without degradation of the antenna performance, and conversely

11

with enhancement of antenna functions, and realization of wideband and multiband operation. Mobile systems other than mobile phones, which function in control, identification, and message transmission/reception mostly within very short ranges, have also accelerated personalization, because they are usually used for individual personal purposes. The typical systems are NFC systems, including RFID systems and ultrawideband (UWB) systems. Antennas used for these systems are not necessarily designed for the particular objectives, but may be ordinary small, compact, and lightweight antennas. They need not necessarily be high gain. On the contrary, mobile broadband systems such as WMAN and worldwide interoperability for microwave access (WiMAX), which deal with high data rate signals, employ generally functional antennas such as adaptive arrays and MIMO systems. Globalization Globalization of communication systems, including mobile systems, has progressed with satellite systems, which are classified by their orbits: low Earth orbit (LEO), medium Earth orbit (MEO), and geostationary orbit (GEO). However, the global communication services do not depend on satellite systems alone, but also on wired systems like Internet Protocol (IP)-based networks, which have worldwide linkage and also connections to wireless mobile networks. In addition, there are also wireless systems in which mobile terminals can roam from country to country, where the same network services are available. The typical systems are GSM systems, which have deployed their networks worldwide, and some 3G systems, including WCDMA and CDMA2000 systems. Dual and tripleband antennas are mounted on mobile terminals for these systems. Some mobile phones install a pentaband antenna and can operate in areas where both 2G and 3G services in different frequency bands are available, in addition to receiving GPS information at the same time. Multimedia Services In addition to nonvoice services, as was mentioned previously, digital technology has gradually increased the transmission rate from the order of kilobits per second (Kbps) to megabits per second (Mbps). The data rate of the 3G system in Japan initially started at 384 Kbps, but in the current 3.5G systems, it has been increased to 14.4 Mbps at maximum by introducing a high speed downlink packet access (HSDPA) system. Now, the trend is to increase the data rate to the order of 100 Mbps in high speed mobile systems and 1 Gbps in lower speed or nomadic status in 3.9G systems, which are the systems aimed at gradually transferring from the 3.5G systems to the 4G systems. In the 4G systems, the highest data rate is expected to be 4 Gbps in the nomadic status.

12

Apart from the mobile phone systems, mobile wireless systems such as WiMAX, which generally assume data transmission backed up by the Internet connection, can treat with high data rate up to several tens of megabits per second in high speed moving status, say more than 100 km/hr. They use sophisticated antenna systems such as adaptive array and MIMO systems. Multimedia services in the ITS are also inevitable for providing drivers and traffic managers with information related to road, traffic, and navigation, to ensure safety and comfortable driving. Multidimensional Networks There is a growing trend for mobile systems to be integrated into other systems so multidimensional services have been made available. A typical example is the combination of communication systems and broadcasting systems; one is the TV broadcasting through communication satellite systems and another is the terrestrial digital TV broadcasting for mobile terminals, including mobile phone handsets. The latter is referred to as ‘‘onesegment broadcasting,’’ because one segment of the 13 orthogonal frequency division multiplexing (OFDM) segments is exclusively designed for broadcasting to mobile terminals. Furthermore, a system referred to as fixed mobile convergence (FMC) has been in practical use recently. The mobile network is combined with a wired network at a home or office environment; thus, users use a mobile phone just as a wireline subscriber’s phone at home, while alternatively using it as an ordinary mobile phone in the outdoor environment. As such, communication systems tend to be integrated into multidimensional networks that embrace multi-informational media, multitransmission media, and multilayered networks. Information media are composed of both voice and nonvoice systems including digital voice, sound, still and moving images, and computer data. Transmission media include both wire and wireless lines, radio, and optical links. Land, maritime, aeronautical, and satellite systems will be integrated into complex multilayered networks, thus allowing seamless communication worldwide regardless of time and space. Wireless communication systems in ITS are combined with IP networks, which will be upgraded to the Internet Protocol version 6 (IPv6) to deal with almost unlimited amounts of information and so that multifunctional services for multiclient situations will become available. Ubiquitous environments will be constituted in the near future. The latest approach is expected to apply the NFC systems, which can function for data transmission, control, identification, and sensing at indefinite numbers of places in very short range regardless of time. Also expected by the deployment of NFC is flexibly and unlimitedly building up networks of home electronic appliances, computers, LAN terminals, and even mobile terminals. Mobile terminals play a great part for such systems. Antennas used for these systems are generally very small, compact, and lightweight.

13

The demand for more intelligent antennas that are integrated with signal processing, adaptive control, and software will continue unabated. Applications of higher frequencies such as microwave and millimeter wave, even terahertz regions, will be explored further to develop novel antenna systems. Software Implementation One of the ways to enhance antenna performance as well as system performance is through the implementation of software. An example is adaptive control of the antenna performance to meet the environmental conditions for obtaining the optimum antenna performance— for example, adaptive control of an antenna circuit when the antenna performance is degraded by the proximity effect like that from the human body. The antenna circuit, for instance, the matching circuit, is adaptively controlled to attain the maximum S/N at the receiver output. This sort of adaptive control is useful for an antenna systems located in an environment where both interference signals and the desired signal exist in the field. In these systems, the software needed is generally simple. Software-defined mobile systems will appear in the near future. One potential system is the software installed mobile terminal, which can roam and operate in areas of different systems by software-defined switching of the operating frequency, data transmission rate, modulation/demodulation schemes, and so on, to match the parameters of different system. The reconfigurable system is a similar potential system, which enables mobile terminals to update the system performance to operate, for example, with adaptability to the different quality of service by the downloaded software without change in the hardware. In these systems, the antenna systems are mostly wideband or multiband in order to correspond to the different uses. 1.2.1 Mobile Systems The digital technique, particularly with regard to progress in VLSI technology, has contributed to remarkable advancement of mobile systems. Analog components are now in the minority, although they are still important. The new techniques have enabled an order of magnitude increase in processing power to be packaged in a much reduced volume of equipment space. The latest mobile phones are equipped with various functions in addition to voice communication facility, such as a camera, Internet connection, movie display, sound delivery reception, and so forth. Such multifunctional mobile terminals have been realized by installing advanced software and hardware. Regarding the software size, the object size has increased up to a few hundred megabytes in 3G systems, whereas it was only on the order of kilobytes in 1G systems. Working memory for temporary storage has also increased proportionately. Software program size also depends on the development of the advanced application processor, which can perform multimedia processing along with the baseband processor [13]. Advancement of these software and hardware implemen-

14

tations has been supported significantly by VLSI technology, by which miniaturized threedimensional circuits, constructed in a tiny package, have become available. This has, in turn, presented a great challenge to antenna designers. It was possibly said that until 10 years ago the typical land mobile communication system was the mobile phone system, which had overwhelmed other mobile systems. However, mobile wireless systems have now come to the fore and have become a competitor to the mobile phone systems in terms of data transmission, since they can presently deal with much higher rate than that of the mobile phone system. The mobile phone systems, initially designed for voice transmission only, have now progressed to treat higher data rate transmission than before and further to raise the data rate that eventually will surpass the mobile wireless systems. Another important trend in mobile systems is application of satellite systems to small mobile terminals on the Earth to treat multifunctions such as multimedia communication, very high rate data (e.g., in order of Mbps), and high-quality information transmission. For this purpose, the satellite systems install a large deployable antenna and the high quality of services (QoS) management system on the satellite. Mobile terminals on the Earth can be either a type of handset or a small portable personal computer (PC) and use small planar antennas. Table 1.4 itemizes the trends in mobile systems for the last century, subdivided into six generations. In the early days the services were provided by single channel systems covering unspecified areas on the Earth and oceans. Subsequently, antennas have evolved into numerous composite and integrated configurations and arrays with sophisticated adaptive control and signal processing facilities. In urban areas, multipath fading has Table 1.4 Trends in Mobile Communication Concepts Year

1900–

Environment Over Earth/sea Propagation Radiation Antenna Size Function System

1950–

1970–

Urban, highway, rail

Reflection/ Multipath diffraction Omnidirectional Shaped pattern Single element Composite Phased array Electrically small

Low profile

Telegraph

Telephone

Service area Wide coverage Channel Single Problems

1990–

2000–

Indoors/tunnels Mobile satellites Delay spread

2010– Home/office

Built-in

Circular polarization Sig. processing Adaptive array Reconfigurable MIMO antenna Downsizing

Diversity Analog

Digital

Adaptive control, RAKE Data transmission, multimedia, high data rate Macrozone Microzone Picozone Femtozone Multizone Multiple access Intermodulation, multipath fading, EM noise, delay, proximity effects

15

become a serious problem, while zone arrangement and multichannel access networks improve the efficiency of frequency spectrum management. The rapid growth in personal mobile terminals has stimulated the creation of new types of small antennas, and deployment of newly developed mobile wireless systems spurred development of adaptive arrays and MIMO systems. 1.2.2 Increasing Information Flow Mobile phone services have been expanded to include nonvoice message transmission, and the latest mobile phones are designed for the transmission and reception of a variety of information media, including sound, graphics, characters, computer data, and video images. The latest mobile phone handsets can display graphics, characters, and both still and moving video in color with high quality on the LCD panel. In addition, services such as delivery of music, playing of games, broadcasting of radio and/or TV, mobile banking and commerce, and so on, have become available. In another way, mobile terminals can function in control and telemetering by attaching small equipment to handsets for that purpose. A typical example is a mobile terminal, to which RFID equipment is attached. By using this mobile terminal (actually the RFID), a registered person entering a room is identified at the entrance and can unlock the door. Another example is in production management with the help of an RFID mounted mobile terminal at the manufacturing plant. The RFID on the mobile terminal can retrieve the product data stored in the small chip attached to the product and send it to the data center, where the collected data are processed for production management functions. The RFID mounted mobile handsets are also useful for control and operation of electronic appliances at home. IN the not too distant future, the use of RFID will evolve to constitute ubiquitous control and communication environments at home, office, and other sites of confined spaces. 1.2.3 Propagation With the enormous demand for mobile systems, there is a trend that radio environments will become smaller zones such as micro-, pico-, and femto-cells, and antenna height will be lowered to cover less visible lower levels. Much research is being directed toward methods of precisely locating the sites of a call within a small cell region. The use of indoor mobile systems has been increasing and the problems of small cell propagation are well appreciated. 1.3 MODERN MOBILE ANTENNA DESIGN The progress in mobile antenna design is listed in Table 1.5, which shows the related technical issues over the six generations that were cited in Table 1.4. As the system

16

Table 1.5 Progress in Antennas for Mobile System Year

1900–

1950–

1970–

1990–

2000–

2010–

Frequency

10 kHz
h 0 for h b ≤ h 0

(3.34)

76

冦

0.7

k f = −4 +

冉 冉

冊 冊

fc −1 925

f 1.5 c − 1 925

for medium-sized city and suburban centers with medium tree density (3.35) for metropolitan centers

For approximate work, the following parameter values can be used: h0 =

再

3n floors 3n floors + 3

for flat roofs for pitched roofs

w = 20m to 50m; d m =

(3.36)

w ; ␾ = 90° 2

where n floors is the number of floors in the building. The model is applicable for 800 MHz ≤ f c ≤ 2,000 MHz, 4m ≤ h b ≤ 50m, 1m ≤ h m ≤ 3m, and 0.02 km ≤ R ≤ 5 km. An alternative approach is to replace the L msd term by L n (t ) from the flat edge model. This would enable the path loss exponent to vary according to the number of buildings and to be uniformly valid for h b ≤ h 0 . Note, however, that for very low base station antennas, other propagation mechanisms, such as diffraction around vertical building edges and multiple reflections from building walls, are likely to be significant (see Section 3.3). 3.2.4 Comparison of Models The path loss predictions of all of the models described in this chapter are compared in Table 3.2, which shows the exponents of power variation predicted by each model. Thus a ‘‘−2’’ in the path loss exponent column means that the model predicts that received power is inversely proportional to the square of range. In some cases it is difficult to express the variation in this form, but otherwise it is useful as a means of comparison. 3.2.5 Computerized Planning Tools The methods described in this chapter are most often implemented for practical planning within computer software. The development of such software has been motivated and enabled by a number of factors: • • •

The enormous increase in the need to plan cellular systems accurately and quickly. The development of fast, affordable computing resources. The development of geographical information systems, with index data on terrain, clutter, and land usage in an easily accessible and manipulable form.

77

Table 3.2 Comparison of Macrocell Propagation Models

Model Free space (Sec. 2.2.2) Plane earth (Sec. 2.2.3) Egli (Sec. 3.2.2.1)

Path Loss Exponent (−)

hb

hm

fc

−2 −4 −4

0 2 2

0 2 1 (h m < 10) 2 (h m > 10) See (3.5)

−2 0 −2 ≈ −2.6

2 (h 0 − h m )−2 (h 0 − h m )−2 (h 0 − h m )−2 ≈1 ≈1

10−fc /400 ≈ −2 −1 ≈ −1 −1.05 See (3.28)

Okumura/Hata (Sec. 3.2.2.2) −4.5 + 0.66 COST231-Hata (Sec. 3.2.2.3) log h b Ibrahim (Sec. 3.2.2.4) −4 Allsebrook (Sec. 3.2.3.1) −4 Ikegami (Sec. 3.2.3.2) −1 Flat edge (Sec. 3.2.3.4) From −2 to −4 Walfisch-Bertoni (Sec. 3.2.3.5) −3.8 COST-231 Walfisch-Ikegami (Sec. 3.2.3.6) −3.8

1.38 + 0.66 log r 2 2 0 ≈2 ≈ 1.8 ≈ 1.8

Such techniques have been implemented in a wide range of commercially available and company-specific planning tools. Although most of these are based on combined empirical and simple physical models, it is anticipated that there will be progressive evolution in the future toward more physical or physical-statistical methods as computing resources continue to cheapen, as clutter data improve in resolution and cost, and as research into numerically efficient path loss predication algorithms develops. 3.2.6 Conclusions Propagation path loss modeling is the fundamental method of predicting the range of a mobile radio system. The accuracy of the path loss predictions is crucial in determining whether a particular system design will be viable. In macrocells, empirical models have been used with great success, but deterministic physical models are being increasingly investigated as a means of improving accuracy, based on the use of multiple rooftop diffraction as the key propagation mechanism. This accuracy comes at the expense of increased input data requirements and computational complexity. Another generation of models is expected to appear that combines sound physical principles with statistical parameters, which can economically be obtained in order to provide the optimum balance between accuracy and complexity. The path loss for macrocells may be taken, very roughly, to be given by: PR 1 h h2 = = k m4 b PT L r fc

(3.37)

78

It should be emphasized that this expression, and all of the models described in this chapter, account only for the effects of typical clutter on flat or gently rolling terrain. When the terrain variations are sufficient to cause extra obstruction loss, then the models must be supplemented by explicit calculations of terrain diffraction loss [1]. 3.3 MICROCELLS The deployment of microcells is motivated by a desire to reduce cell sizes in areas where large numbers of users require access to the system. Serving these users with limited radio spectrum requires that frequencies be reused over very short distances, with each cell containing only a reduced number of users. This could, in principle, be achieved with base station antennas at the same heights as in macrocells, but this would increase the costs and planning difficulties substantially. In a microcell, the base station antenna is typically at about the same height as lamp posts in a street (3–6m above ground level), although the antenna is more often mounted onto the side of a building (Figure 3.12). Coverage, typically over a few hundred meters, is then determined mostly by the specific locations and electrical characteristics of the surrounding buildings, with cell shapes being far from circular. Pattern shaping of the base station antenna can yield benefits in controlling interference, but is not the dominant factor in determining the cell shape. The dominant propagation mechanisms are then free-space propagation, plus multiple reflection and scattering within the cell’s desired coverage area, together with diffraction around the vertical edges of buildings and over rooftops, which becomes significant when determining interference

Figure 3.12 A microcell in a built-up area.

79

between cochannel cells. Microcells thus make increased use of the potential of the environment surrounding the base station to carefully control the coverage area and hence to manage the interference between sites. More general information on microcell systems is available [20–22].

3.3.1 Dual-Slope Empirical Models To model the path loss in microcells, empirical models of the type described earlier, could, in principle, be used. However, measurements (see, e.g., [23]) indicate that a simple power law path loss model cannot usually be used to fit measurements with good accuracy. A better empirical model in this case is a dual-slope model. Two separate path loss exponents are used to characterize the propagation, together with a breakpoint distance of a few hundred meters between them where propagation changes from one regime to the other. In this case the path loss is modeled as:

冦冉 冊 k

1 = L

for r ≤ r b

r n1 k

r rb

n2

(3.38)

for r > r b rnb 1

or, in decibels:

L=

10n 1 log r + L 1 r 10n 2 log + 10n 1 log r b + L 1 rb

冦

for r ≤ r b for r > r b

(3.39)

where L 1 is the reference path loss at r = 1m, r b is the breakpoint distance, n 1 is the path loss exponent for r ≤ r b , and n 2 is the path loss exponent for r > r b . To avoid the sharp transition between the two regions of a dual-slope model, the model can also be formulated as follows, according to an approach proposed by [24]: 1 = L

k

r n1

冉 冊 r 1+ rb

n2 − n1

(3.40)

80

This can be considered in two regions: for r > r b , L r 1

k

. Hence the path loss exponent is again n 1 for short distances and n 2 for larger n r 2 rb distances. The model is conveniently expressed in decibels as:

冉冊

冉 冊

L = L 1 + 10n 1 log r + 10(n 2 − n 1 ) log10 1 +

r rb

(3.41)

where L ref is a reference value for the loss at 1m. Figure 3.13 compares (3.39) and (3.41). Typical values for the path loss exponents are found by measurement to be around n 1 = 2 and n 2 = 4, with breakpoint distances between 200 and 500m, but it should be emphasized that these values vary greatly among individual measurements. (See, for example, [23, 25, 26].) To plan the locations of microcells effectively, it is important to ensure the coverage areas of the cochannel cells do not overlap within the breakpoint distance. The rapid

Figure 3.13 Dual-slope empirical loss models. Here n 1 = 2, n 2 = 4, r b = 100m, and L 1 = 20 dB.

81

reduction of signal level beyond the breakpoint then produces a large carrier-to-interference ratio, which can be exploited to maximize system capacity. 3.3.2 Physical Models In creating physical models for microcell propagation, it is useful to distinguish line-ofsight (LOS) and nonline-of-sight (NLOS) situations. We will see that it is possible to make some reasonable generalizations about the LOS cases, while the NLOS cases require more site-specific information. See Figures 3.14 and 3.15 for an example of practical measurements, in which it is clear that the obstructed path suffers far greater variability at a given range than the others. Such effects must be accounted for explicitly in the models. 3.3.2.1 Line-of-Sight Models (Two-Ray Models) In a line-of-sight situation, at least one direct ray and one reflected ray will usually exist (Figure 3.16). This leads to an approach similar to that followed in the derivation of the

Figure 3.14 Measurements of path loss from a suburban microcell. Routes A, B, and C are radial streets, often with a line-of-sight present, whereas route D is a transverse street, with most locations obstructed. Note how the measurements on route D vary over almost 45 dB, despite the range being almost constant at around 30m. The frequency is 900 MHz.

82

Figure 3.15 Measurement routes corresponding to Figure 3.14.

plane earth loss (two-wave theory) in Chapter 2, except that it is no longer appropriate to assume that the direct and reflected path lengths are necessarily similar, or that the reflection coefficient necessarily has a magnitude of unity. The loss is then:

冉 冊|

1 ␭ = L 4␲

2

e −jkr 1 e −jkr 2 +R r1 r2

|

2

(3.42)

where R is the Fresnel reflection coefficient for the relevant polarization. In the horizontally polarized case the reflection coefficient is very close to −1, so the path loss exponent tends toward 4 at long distances as in the plane earth loss. For the vertically polarized case, the path loss exponent is essentially 2 at all distances, but the large fluctuations present at short ranges disappear at longer distances. Hence both cases produce two regimes of propagation. Because the reflection coefficient for vertical polarization is approximately +1 for large distances, the distance at which the rays are in antiphase is closely approximated by the distance at which r 2 = (r 1 + ␭ /2), and this gives the position of the last dip in the

83

Figure 3.16 Two-ray model of line-of-sight propagation.

vertically polarized signal. This is exactly the definition of the first Fresnel zone. For high frequencies, the distance at which the first Fresnel zone first touches the ground is given approximately by: rb =

4h b h m ␭

(3.43)

It has been suggested that this forms a physical method for calculating the breakpoint distance for use in empirical models such as in (3.41) [27]. The two-ray model (Figure 3.17) forms a useful idealization for microcells operated in fairly open, uncluttered situations, such as on long straight motorways where a lineof-sight path is always present and little scattering from other clutter occurs.

3.3.2.2 Street Canyon Models Although a line-of-sight path frequently exists within microcells, such cells are most usually situated within built-up areas. The buildings around the mobile can all interact with the transmitted signal to modify the simple two-ray regime described earlier. A representative case is illustrated in Figure 3.18. This case assumes that the mobile and base station are both located in a long straight street, lined on both sides by buildings with plane walls. Models that use this canonical geometry are called street canyon models.

84

Figure 3.17 Predictions from the two-ray model. Here h b = 6m, h m = 1.5m, f c = 900 MHz, and the constitutive parameters of the ground are ⑀ r = 15 and ␴ = 0.005 Sm−1.

Figure 3.18 Street canyon model of line-of-sight microcellular propagation.

85

Six possible ray paths are also illustrated. Many more are possible, but these tend to include reflections from more than two surfaces. These are typically attenuated to a much greater extent, so the main signal contributions are accounted for by those illustrated. The characteristics of this approach are illustrated by reference to a four-ray model, which considers all three of the singly reflected paths from the walls and the ground. The structure follows, but the reflections from the vertical building walls involve the Fresnel reflection coefficients for the opposite polarization to the ground. A typical result is shown in Figure 3.19. In comparison to Figure 3.17, the multipath fading is more rapid, and there are fewer differences between vertical and horizontal polarization. Eventually the vertically polarized component diminishes with an average path loss exponent of 4, while the horizontally polarized case tends to 2. However, real streets are rarely straight for long enough to observe this distance range. Figure 3.20 shows the variation of fields predicted by this model as the base station antenna height is varied at a particular range. Neither polarization shows any definite advantage in increasing the antenna height, and the particular positions of the nulls in this figure are strongly dependent on the range. In general, for line-of-sight microcells, the base station height has only a weak effect on the cell range. There is some effect due to the obstructing effect of clutter (in this case vehicles, street furniture and pedestrians), but we will see in later sections that increasing the base station height does have a

Figure 3.19 Predictions from four-ray model. Here h b = 6m, h m = 1.5m, w = 20m, d m = 10m, d b = 5m, f c = 900 MHz, and the constitutive parameters of the ground and buildings are ⑀ r = 15 and ␴ = 0.005 Sm−1.

86

Figure 3.20 Base station antenna height variation according to the four-ray model. Here r = 50m, other parameters are as given in Figure 3.19.

significant effect on interference distance. Thus the antenna should be maintained as low as possible, consistent with providing a line-of-sight to locations to be covered. 3.3.3 Nonline-of-Sight Models 3.3.3.1 Propagation Mechanisms When the line-of-sight path in a microcell is blocked, signal energy can propagate from the base to the mobile via several alternative mechanisms: • • •

Diffraction over building rooftops; Diffraction around vertical building edges; Reflection and scattering from walls and the ground.

These mechanisms are described further in [28]. At relatively small distances from the base station and low base antenna heights, the angle through which the signal must diffract over rooftops in order to propagate is large and the diffraction loss is correspondingly big. At such distances, propagation is dominated by the other two mechanisms described above, where the balance between the diffraction and reflection depends on the specific

87

geometry of the buildings. For instance, Figure 3.21 shows the motion of a mobile across the shadow boundary created by a vertical building edge. Because this building is in an isolated situation, the only possible source of energy in the shadow region is via diffraction and the energy will drop off very rapidly with increasing distance. This contrasts with the case illustrated in Figure 3.22, where the building is now surrounded by others that act as reflecting surfaces. The reflected ray is then likely to be much stronger than the diffracted ray, so that the signal remains strong over much larger distances. At larger distances still, particularly those involved in interference between cochannel microcells, the rooftop diffracted signal (Figure 3.23) again begins to dominate due to the large number of diffractions and reflections required for propagation over long distances. See, for example, Figure 3.24, which shows the plan view of buildings arranged in a regular ‘‘Manhattan grid’’ structure. In this figure, the short paths A and B involve only a single reflection/diffraction and are likely to be dominant sources of signal energy. By contrast, the long path C is likely to be very weak as four individual reflection losses are involved, and the rooftop-diffracted path D is then likely to dominate. This variation in propagation mechanism with distance is another source of the two slopes in the empirical models of Section 3.3.1. System range is greatest along the street containing the base site. When the mobile turns a corner off of this street, the signal drops rapidly, often by 20–30 dB. The resultant

Figure 3.21 Street geometry where diffraction dominates.

Figure 3.22 Street geometry where reflection dominates.

88

Figure 3.23 Rooftop diffraction as an interference mechanism.

Figure 3.24 Variation of propagation mechanisms with distance for nonline-of-sight microcells.

coverage area is therefore broadly ‘‘diamond’’ shaped, as illustrated in Figure 3.25, although the precise shape will depend very much on the building geometry. The curved ‘‘sides’’ of the diamonds in Figure 3.25, which have been confirmed by measurement, have been shown to indicate that the dominant mechanism of propagation into cross streets is diffraction rather than reflection [29]. The variation of the microcell shape with base antenna height in a Manhattan grid structure has been investigated in detail [30] using the multiple diffraction integral in [16] and it is shown that there is a smooth transition from a diamond shape to nearly circular as the antenna height increases. It has also been shown [31] that the characteristic diamond cell shape is obtained even when considering only the vertical corner diffraction plus

89

Figure 3.25 Microcellular propagation along city streets; closed circle, position of the base site; dashed line, equal path loss contours.

reflections from building walls. This work also showed that the distance at which the transition between the various mechanisms occurs depends strongly on the distance between the base station and the nearest street corners. For low antenna heights, the strong scattering present in microcells prevents the efficient use of sectorization since the free-space antenna radiation pattern is destroyed. Efficient frequency reuse can still be provided, however, by taking advantage of the building geometry. In regular street grid structures cochannel microcells should be separated diagonally across the street directions, and with sufficient spacing to ensure that cells do not overlap within their break point distance in order to maintain high signal-tointerference levels. In more typical environments, where the buildings are not regular in size, more advanced planning techniques must be applied, particularly when frequencies are shared between microcell and macrocell layers. See, for example, [32]. 3.3.3.2 Recursive Model This model is intermediate between an empirical model and a physical model [33]. It uses the concepts of GTD/UTD in that effective sources are introduced for nonline-of-

90

sight propagation at the street intersections where diffraction and reflection points are likely to exist. The model breaks the path between the base station and the mobile down into a number of segments, interconnected by nodes. The nodes may be placed either just at the street intersections or else at regular intervals along the path, allowing streets that are not linear to be handled as a set of piecewise linear segments. An illusory distance for each ray path considered is calculated according to the following recursive expressions: k j = k j − 1 + d j − 1 × q (␪ j − 1 )

(3.44)

d j = k j × r j−1 + d j−1 subject to the initial values: k0 = 1

(3.45)

d0 = 0 where d n is the illusory distance calculated for the number of straight line segments along the ray path, n (n = 3 in Figure 3.26) and r j is the physical distance [meters] for the line segment following the jth node. The result is reciprocal. The angle through which the path turns at node j is ␪ j (degrees). As this angle increases, the illusory distance is increased according to the following function:

Figure 3.26 Example geometry for recursive microcell model.

91

q (␪ j ) =

冉 冊 ␪ j q 90 90

v

(3.46)

where q 90 and ␯ are parameters of the model, with q 90 = 0.5 and ␯ = 1.5 suggested in [33]. The path loss is then calculated as:

冋 冉 ∑ 冊册

4␲ d n L = 20 log D ␭

n

j=1

r j−1

(3.47)

where: r D (r ) = r b 1

冦

for r > r b

(3.48)

for r ≤ r b

This expression, (3.47), creates a dual-slope behavior with a path loss exponent of 2 for distances less than the breakpoint r b and 4 for greater distances. The overall model is simple to apply and accounts for the key microcell propagation effects, namely, dualslope path loss exponents and street corner attenuation, with an angle dependence that incorporates effects encountered with real street layouts.

3.3.3.3 Site-Specific Ray Models Prediction of the detailed characteristics of microcells requires a site-specific prediction based on detailed knowledge of the built geometry. Electromagnetic analysis of such situations is commonly based on use of the geometrical theory of diffraction (GTD) and its extensions, as described in Section 3.5.3. These models are capable of very high accuracy, but their practical application has been limited by the cost of obtaining sufficiently detailed data and the required computation time. More recently progress in satellite remote-sensing has reduced the cost of the necessary data while advanced ray-tracing techniques and cheap computational resources have advanced the art to the point where ray-tracing models are entirely feasible. In practice, however, many operators consider the costs involved with operating such prediction tools to be prohibitive and prefer to deploy their microcells based on the knowledge of experienced planning engineers together with site-specific measurements. Example predictions from a GTD-based model for a real building geometry are shown in Figure 3.27. This includes contributions from a very large number of multiply reflected and diffracted rays.

92

Figure 3.27 Predictions for an urban microcell, based on ray tracing and the geometrical theory of diffraction. The shading indicates the received signal strength at a mobile with a quarter-wave monopole antenna (in dBm), with a transmit power of 1W at 900 MHz.

3.3.4 Microcell Propagation Models: Discussion For the practical application of microcell propagation models, there is an important tradeoff between the accuracy of the prediction and the speed with which the prediction can be made. Microcells often have to be deployed very quickly, with little engineering effort, often by people who are not necessarily radio experts. Rules of thumb and very rapid statistical planning tools are very important. Also, even with a very high resolution topographic database, propagation may often be dominated by items of street furniture (signs, lampposts, and so on) and by details of the antenna siting and its interaction with the objects on which it is mounted, which no database could hope to have available. These features may also change rapidly with time, as is certainly the case when dealing with the effects of traffic. For example, when a double-decker bus passes close by a microcell antenna, the coverage area of the microcell may change dramatically for a short time. Either these items must be entered by hand or, more likely, systems of the future will have to be capable of adapting their characteristics to suit the environment which they find, by taking measurements from the active network and responding accordingly.

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These factors will dramatically change the way in which propagation models are applied, from being processes that are run at the start of a system deployment, and then used to create a fixed set of predictions and recommendations for deployment, to realtime processes that operate within the base station, with assistance from the mobiles, which are optimized on an ongoing basis and are used by the system to assess the likely impact of changes to system parameters such as transmit powers, antenna patterns, and channel assignments. 3.3.5 Microcell Shadowing The lognormal distribution is applied to shadow fading in microcells, just as for macrocells [1]. Some measurements have suggested that the location variability increases with range, typically in the range of 6–10 dB [34]. To account for microcell shadowing crosscorrelation, the shadowing can be separated into two parts, one of which is caused by obstructions very local to the mobile and therefore common to all paths, and another which is specific to the transport of energy from the mobile to a particular base station [35]. 3.3.6 Conclusions Propagation in microcells can be modeled using either empirical or physical models, as was the case for the macrocells in Section 3.2. In either case, however, the clutter surrounding the base station has a significant impact on the cell shape and this must be accounted for to avoid serious prediction errors. In particular, a simple path loss exponent model is inadequate and dual-slope behavior must be accounted for. This clutter also creates difficulties when deploying antennas, because the clutter disrupts the free-space antenna radiation pattern. Nevertheless, the enormous potential offered by microcells in creating high-capacity cellular systems makes them increasingly attractive methods of providing outdoor coverage in areas with high user densities. 3.4 PICOCELLS When a base station antenna is located inside a building, a picocell is formed (Figure 3.28). Such cells are increasingly used in cellular telephony for high-traffic areas such as railway stations, office buildings, and airports. Additionally, the high data rates required by wireless local-area networks restrict cell sizes to picocells and impose a further requirement to predict the wideband nature of the picocell environment. The subject of picocell propagation is also relevant to determining propagation into buildings from both macrocellular and microcellular systems, which could either act as a source of interference to the indoor cells or as a means of providing a greater depth of coverage without capacity

94

Figure 3.28 Picocells.

enhancement. This section describes both empirical and physical models of picocellular propagation. 3.4.1 Empirical Models of Propagation Within Buildings 3.4.1.1 Wall and Floor Factor Models Two distinct approaches have been taken here. The first is to model propagation by a path loss law, just as in macrocellular systems, determining the parameters from measurements. This approach, however, tends to lead to large errors in the indoor case because of the large variability in propagation mechanisms among different building types and among different paths within a single building. The same is true if dual-slope models, similar to those used in microcells, are applied. A more successful approach [36] is to characterize indoor path loss by a fixed path loss exponent of 2, just as in free space, plus additional loss factors relating to the number of floors (n f ) and walls (n w ) intersected by the straight-line distance, r, between the terminals. Thus: L = L 1 + 20 log r + n f a f + n w a w

(3.49)

where a f and a w are the attenuation factors (in decibels) per floor and per wall, respectively; L 1 is the loss at r = 1m. No values for these factors were reported in [36]. An example

95

prediction using this model is shown in Figure 3.29 for a series of offices leading off a corridor, with the base station inside one of the offices. Contours are marked with the path loss (−dB). A similar approach is taken by an ITU-R model [37], except that only the floor loss is accounted for explicitly, while the loss between points on the same floor is included implicitly by changing the path loss exponent. The basic variation with frequency is assumed to be the same as in free space, resulting in the following total path loss model (in decibels): L T = 20 log f c + 10n log r + L f (n f ) − 28

(3.50)

where n is the path loss exponent, given by Table 3.3, and L f (n f ) is the floor penetration loss, which varies with the number of penetrated floors, n f according to Table 3.4. 3.4.1.2 COST 231 Multiwall Model This model of propagation within buildings [9] incorporates a linear component of loss, proportional to the number of walls penetrated, plus a more complex term, which depends on the number of floors penetrated, producing a loss that increases more slowly as additional floors after the first are added.

Figure 3.29 Example picocellular path loss prediction.

96

Table 3.3 Path Loss Exponents, n, for the ITU-R Model (3.50)

Frequency 900 MHz 1.2–1.3 GHz 1.8–2.0 GHz 4 GHz 60 GHz

Residential

Environment Office

Commercial

— — 2.8 — —

3.3 3.2 3.0 2.8 2.2

2.0 2.2 2.2 2.2 1.7

The 60 GHz figures apply only within a single room for distances less than around 100m, since no wall transmission loss or gaseous absorption is included. Table 3.4 Floor Penetration Factors, L f (n f ) [dB] for the ITU-R Model (3.50) Frequency (GHz)

Residential

Environment Office

Commercial

4n f

9 (1 floor) 19 (2 floors) 24 (3 floors) 15 + 4 (n f − 1)

6 + 3 (n f − 1)

0.9

1.8–2.0

Note that the penetration loss may be overestimated for large numbers of floors, for reasons described in Section 3.4.3.1.

W

LT = LF + Lc +

∑ n wi L wi

冋 +n f

nf + 2 −b nf + 1

册L

f

(3.51)

i=1

where L F is the free-space loss for the straight-line (direct) path between the transmitter and receiver, n wi is the number of walls crossed by the direct path of type i, W is the number of wall types, L wi is the penetration loss for a wall of type i, n f is the number of floors crossed by the path, b and L c are empirically derived constants, and L f is the loss per floor. Recommended values for 1,800 MHz are L w = 3.4 dB for light walls, 6.9 dB for heavy walls, L f = 18.3 dB, and b = 0.46. The floor loss [that is, the last term in (3.51)] is shown in Figure 3.30. Notice that the additional loss per floor decreases with increasing number of floors. 3.4.1.3 Ericsson Model In this model, intended for use around 900 MHz, the path loss, including shadowing, is considered to be random, variable, and uniformly distributed between limits that vary with distance as indicated in Table 3.5 [38]. The path loss exponent increases from 2 to

97

Figure 3.30 Floor loss for COST231 multiwall model. Table 3.5 Ericsson Indoor Propagation Model Distance (m)

Lower Limit of Path Loss (dB)

Upper Limit of Path Loss (dB)

1 < r < 10 10 ≤ r < 20 20 ≤ r < 40 40 ≤ r

30 + 20 log r 20 + 30 log r −19 + 60 log r −115 + 120 log r

30 + 40 log r 40 + 30 log r 1 + 60 log r −95 + 120 log r

12 as the distance increases, indicating a very rapid decrease of signal strength with distance. A typical prediction from the model is shown in Figure 3.31. The model may be extended for use at 1,800 MHz by the addition of an 8.5-dB extra path loss at all distances. 3.4.2 Empirical Models of Propagation into Buildings 3.4.2.1 Introduction There are two major motivations for examining signal penetration into buildings. First, because most cellular users spend most of their time inside buildings, the level of service

98

Figure 3.31 Prediction from Ericsson in-building path loss model (900 MHz).

they perceive will depend heavily on the signal strengths provided inside the buildings (the depth of coverage). When sufficient capacity exists within the macrocells and the microcells of the network, this indoor coverage is then provided by the degree of penetration into the buildings. When, by contrast, it is necessary to serve very high densities of users within a building (for example, in heavily populated office buildings, railway stations, and airports), the indoor coverage must then be provided by dedicated picocells. It is inefficient to allocate distinct frequencies to these, so it is necessary to reuse frequencies already allocated to macrocells and microcells, via clear knowledge of the extent to which the two cell types will interfere within the building. 3.4.2.2 COST231 Line-of-Sight Model In cases where a line-of-sight path exists between a building face and the external antenna, the following semiempirical model has been suggested [9], with geometry defined by Figure 3.32. In this figure r e is the straight path length between the external antenna and a reference point on the building wall; since the model will often be applied at short ranges, it is important to account for the true path length in three dimensions, rather than the path length along the ground. The loss predicted by the model varies significantly as rp the angle of incidence, ␪ = cos−1 , is varied. re

冉冊

99

Figure 3.32 Geometry for COST231 line-of-sight building penetration model.

The total path loss is then predicted using: L T = L F + L e + L g (1 − cos ␪ )2 + max(L 1 , L 2 )

(3.52)

where L F is the free-space loss for the total path length (r i + r e ), L e is the path loss through the external wall at normal incidence (␪ = 0°), L g is the additional external wall loss incurred at grazing incidence (␪ = 90°), and: L 1 = n w L i ; L 2 = ␣ (r i − 2)(1 − cos ␪ )2

(3.53)

where n w is the number of walls crossed by the internal path r i , L i is the loss per internal wall, and ␣ is a specific attenuation (dBm−1), which applies for unobstructed internal paths. All distances are in meters. The model is valid at distances up to 500m and the parameter values in Table 3.6 are recommended for use in the 900- to 1,800-MHz frequency range. These have been found to give good agreement with measurements in real buildings, and implicitly include the effects of typical furniture arrangements. Table 3.6 Parameters for COST231 Line-of-Sight Model Parameter L e or L i (dB m−1 )

L g (dB) ␣ (dB m−1 )

Material

Approximate Value

Wooden walls Concrete with nonmetallized windows Concrete without windows Unspecified Unspecified

4 7 10–20 20 0.6

100

3.4.2.3 Floor Gain Models In most macrocell cases, no line-of-sight path exists between the base station and the face of the building. Empirical models of this situation are then most usually based on comparing the path loss encountered in the street outside the building (L out in Figure 3.33) to the path loss within the building at various floor levels. (L n , where n is the floor number defined in Figure 3.33). It is then possible to define a penetration loss as: L p = L n − L out

(3.54)

Interestingly, the penetration loss has been found to decrease with frequency in some studies; typical values for the ground floor penetration loss L 0 are 14.2, 13.4, and 12.8 dB measured at 900, 1,800, and 2,300 MHz, respectively [39]. This does not necessarily indicate that the actual wall attenuations follow this trend, since the penetration loss defined this way makes no attempt to isolate effects due to individual waves. The loss decreases with height from the ground floor upward at a rate of around 2 dB per floor and then starts to increase again with height above around the 9th [40] or 15th [41] floor. The precise variation is likely to be very dependent on the specific geometry of the surrounding buildings. 3.4.2.4 COST231 Nonline-of-Sight Model This model relates the loss inside a building from an external transmitter to the loss measured outside the building, on the side nearest to the wall of interest, at 2m above ground level. The loss is given by:

Figure 3.33 Geometry for building penetration in non-line-of-sight conditions.

101

L T = L out + L e + L ge + max(L 1 , L 3 ) − G fh

(3.55)

where L 3 = ␣ r i , and r i , L e , ␣ , and L 1 are as defined in the COST231 line-of-sight model (Section 3.4.2.2), and the floor height gain, G fh , is given by: G fh =

再

nG n hG h

(3.56)

where h is the floor height above the outdoor reference height (m) and n is the floor number, as defined in Figure 3.33. Shadowing is predicted to be lognormal with location variability of 4–6 dB. Other values are as shown in Table 3.7. Both the line-of-sight and nonline-of-sight models of COST 231 rely on the dominant contribution penetrating through a single external wall. A more accurate estimation may be obtained by summing the power from components through all of the walls.

3.4.3 Physical Models of Indoor Propagation As with microcellular predictions, ray tracing and the geometrical theory of diffraction have been applied to deterministic prediction of indoor propagation (see, for example, [42]). This can be used for site-specific predictions, provided that sufficient detail of the building geometry and materials is available. More advanced electromagnetic prediction techniques, such as the finite-difference time-domain (FDTD) approach are also useful in some cases. Such models also yield wideband information and the statistics of multipath propagation directly. As with physical models of microcellular propagation, however, limitations are associated with using physical models for practical picocell predictions due to the difficulty of obtaining and using sufficiently accurate physical data. These problems are particularly significant for picocells, where the influence of furniture and of the movement of people can have a significant (and time-varying) effect on coverage. Some basic physical models can, however, be used to yield insight into the fundamental processes affecting building propagation. Table 3.7 Parameters for COST231 Nonline-of-Sight Model Parameter L ge (dB) at 900 MHz L ge (dB) at 1800 MHz G n (dB per floor) at 900 or 1800 MHz

Approximate Value 4 6 1.5–2 normal buildings 4–7 for floor heights above 4m

102

3.4.3.1 Propagation Between Floors Figure 3.34 shows four distinct paths between a transmitter and receiver situated on different floors of the same building. Path 0 is the direct path, which encounters attenuation due to the building floors. Models such as those described in Section 3.4.1 implicitly assume that this path is the dominant source of signal power, although the wall and floor loss factors applied can be modified to account for the average effect of other paths. Paths 1 and 2 encounter diffraction in propagating out of, and back in through, the windows of the building, but are unobstructed in propagating between the floors. Finally, path 3 is also diffracted through the windows of the building, although this is through a smaller angle than path 2. It is reflected from the wall of a nearby building before diffracting back into the original building. To analyze the field strength due to paths 2 and 3, the geometry is approximated by the double wedge geometry in Figure 3.35, representing the building edges at the points where the rays enter and leave the building. The propagation is then analyzed using the geometrical theory of diffraction. The source is a point source and therefore radiates spherical waves. The field incident on wedge 1 is therefore: E1 =

√

Z0

PT 4␲ r 21

=

1 2r 1

√

Z 0PT ␲

(3.57)

where P T is the effective isotropic radiated power from the source. The diffraction process at wedge 1 then yields a field incident on wedge 2, which is approximated using GTD as:

Figure 3.34 Alternative paths for propagation between floors.

103

Figure 3.35 Double wedge geometry.

E2 = E1 × D1 ×

√

r1 r 2 (r 1 + r 2 + r 3 )

(3.58)

where the square-root factor is the spreading factor for spherical wave incidence on a straight wedge (see [43], p. 768, for example). Similarly, the field at the field point is: E3 = E2 × D2 ×

√

r1 + r2 r 3 (r 1 + r 2 + r 3 )

(3.59)

Hence the power available at an isotropic receive antenna is: Pr = P T

冉 冊

␭ 2 E 23 ␭ 2 D 21 D 22 × = PT 4␲ Z 0 4␲ r 1 r 2 r 3 (r 1 + r 2 + r 3 )

(3.60)

This result can be applied to both paths 1 and 2 by substitution of the appropriate distances. Path 3 also follows in the same way, but (3.60) is multiplied by the reflection coefficient of the nearby building. The sum of the power from the various contributions is shown in Figure 3.36. It is clear that two regimes are present; for small spacing between the transmitter and receiver, the signal drops rapidly as the multiple floor losses on path 0 accumulate. Eventually the diffracted paths (1 and 2) outside the building dominate, and these diminish far less

104

Figure 3.36 Variation of path loss with number of floors; here the floor height is 4m, building width is 30m, distance to the adjacent building is 30m, and the frequency is 900 MHz.

quickly with distance. When a reflecting adjacent building is present, the diffraction losses associated with this path are less and this provides a significant increase in the field strength for large separations. 3.4.3.2 Propagation on Single Floors When the transmitter and receiver are mounted on the same floor of a building, the dominant mode of propagation is line-of-sight, as shown in Figure 3.37. However, the floor and ceiling-mounted objects will result in the Fresnel zone around the direct ray becoming obstructed at large distances, and this will give rise to additional loss due to

Figure 3.37 Propagation between antennas on a single floor.

105

diffraction. The effective path loss exponent will then be increased and the signal strength will fall off very rapidly with distance. The point at which this occurs depends on the specific geometry, with the maximum unobstructed range being obtained when the antennas are mounted at the midpoint of the gap between the highest floor-mounted obstructions and the lowest point of the ceiling-mounted obstructions [44].

3.4.4 Constitutive Parameters for Physical Models All physical models require both the geometry and constitutive parameters of the buildings as input. Because walls and floors in buildings are inhomogeneous, predictions should, in principle, account for effects such as the reinforcement of concrete walls using steel, the periodic layered structures in cavity walls, plus other similar effects for which it is extremely difficult to obtain detailed data. However, useful information on both transmission and reflection properties can be obtained using the Fresnel reflection coefficients by assuming that walls and floors are plane and infinite, and by properly accounting for refraction at both of the interfaces between each wall/floor and free space. Representative values of the complex permittivity at various frequencies are given in Table 3.8 [37].

3.4.5 Propagation in Picocells: Discussion Propagation effects in picocells are even more geometry dependent than in microcells, placing even greater burdens on the quality of data and computational requirements if deterministic physical models are to produce useful predictions. In the near future, practical picocell system design is more likely to rely on empirical models and engineering experience. In the longer term, however, combinations of physical models with statistics are expected to yield significant benefits.

Table 3.8 Complex Permittivity of Typical Construction Materials

Concrete Lightweight concrete Floorboard (synthetic resin) Plaster board Ceiling board (rock wool) Glass Fiberglass

1 GHz

57.5 GHz

78.5 GHz

95.9 GHz

7.0– j 0.85 2.0– j 0.50 — — 1.2– j 0.01 7.0– j 0.10 1.2– j 0.10

6.50– j 0.43 — 3.91– j 0.33 2.25– j 0.03 1.59– j 0.01 6.81– j 0.17 —

— — 3.64– j 0.37 2.37– j 0.10 1.56– j 0.02 — —

6.20– j 0.34 — 3.16– j 0.39 2.25– j 0.06 1.56– j 0.04 — —

106

3.4.6 Multipath Effects In macrocells, it is usual to assume that waves arrive with uniform probability from all horizontal angles, leading to the classical Doppler spectrum [1]. By contrast, a more reasonable assumption for the indoor environment, particular when propagation occurs between floors, is that waves arrive with uniform probability from all angles. The resulting Doppler spectrum is then relatively uniform so, for simulation purposes, it is reasonable to assume a flat Doppler spectrum, given by:

S( f ) =

冦

1 2f m 0

| f | ≤ fm

(3.61)

f > fm

where f m is the maximum Doppler frequency. With regard to the RMS delay spread of the channel, values encountered in most cases are very much lower than those found in either micro- or macrocells. The variability around the median value is large, however (although there is a strong correlation with the path loss [45]), and there are occasionally cases where the delay spread is very much larger than the median. To provide reasonably realistic simulations both situations must be considered. Tables 3.9 and 3.10 give suitable channels for an indoor office scenario and for an outdoor to indoor scenario respectively, intended for evaluation purposes at around 2 GHz [46]. Values for the RMS delay spread for indoor-indoor environments are also shown in Table 3.11; case A represents low, but frequently occurring values; case B represents median values; and case C gives extreme values that occur only rarely [37]. The very high cases can occur particularly if there are strong reflections from buildings situated significant distances from the building under test. More details of the statistics and structure of the indoor wideband channel are available in references such as [47–50]. In particular, [50] proposes that the power-delay Table 3.9 Indoor Office Test Environment Wideband Channel Parameters Median Channel ␶ RMS = 35 ns

Bad Channel ␶ RMS = 100 ns

Relative Delay, ␶ (ns)

Relative Mean Power (dB)

Relative Delay, ␶ (ns)

Relative Mean Power (dB)

0 50 110 170 290 310

0 −3.0 −10.0 −18.0 −26.0 −32.0

0 100 200 300 500 700

0 −3.6 −7.2 −10.8 −18.0 −25.2

Doppler spectrum for all taps is flat.

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Table 3.10 Outdoor-to-Indoor Test Environment Wideband Channel Parameters Median Channel ␶ RMS = 45 ns

Bad Channel ␶ RMS = 750 ns

Relative Delay, ␶ (ns)

Relative Mean Power (dB)

Relative Delay, ␶ (ns)

Relative Mean Power (dB)

0 110 190 410 — —

0 −9.7 −19.2 −22.8 — —

0 200 800 1200 2300 3700

0 −0.9 −4.9 −8.0 −7.8 −23.9

Doppler spectrum for all taps is classical.

Table 3.11 RMS Delay Spread in Nanoseconds in Indoor-to-Indoor Environments Environment Indoor residential Indoor office Indoor commercial

Case A

Case B

Case C

20 35 55

70 100 150

150 460 500

profile tends to follow a doubly exponential distribution (Figure 3.38), where the peaks of the individual exponentials can be reasonably accurately predicted from ray-tracing models, but where the associated weaker signals result from rough scattering and fine detail that cannot easily be predicted from a deterministic physical model. It has also been

Figure 3.38 Doubly exponential power-delay profile in indoor channels [50].

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observed that the number of the multipath components follows a Gaussian distribution, with a mean value that increases with antenna separation [51]. 3.4.7 Conclusions Picocell propagation is affected by a wide range of mechanisms, operating on a complex, three-dimensional environment, the details of which are rarely available for propagation predictions. Some simple models can give useful estimates of in-building propagation, however, and further progress in these areas is strongly motivated by the growing importance of in-building communication, particularly for very high data rates. The use of appropriate distributed antenna structures helps considerably in providing controlled coverage around buildings, and it is expected that the provision of intelligence within such units will allow systems to be installed without the need for detailed propagation predictions (see Section 3.6). 3.5 MEGACELLS Mobile systems designed to provide truly global coverage using constellations of lowand medium-earth orbit satellites are now in operation. These form megacells, consisting of the footprint from clusters of spotbeams from each satellite, which move rapidly across the earth’s surface. Signals are typically received by the mobile at very high elevation angles, so that only environmental features that are very close to the mobile contribute significantly to the propagation process (Figure 3.39). Atmospheric effects are also significant in systems operated at SHF and EHF [1]. Megacell propagation prediction techniques must also combine predictions of fast (multipath) fading and of shadowing effects, since these tend to occur on similar distance scales and cannot therefore be easily separated. The predictions tend to be highly statistical in nature, since coverage across very wide areas must be included, while still accounting for the large variations due to the local environment. Mobile satellite systems are usually classified according to their orbit type: low earth orbits (LEO) involve satellites at an altitude of 500–2,000 km and require a relatively large number of satellites to provide coverage of the whole of the earth (for example, 66 at 780 km in the case of the Iridium system). Medium earth orbits (MEO) involve altitudes of around 5,000–12,000 km and involve fewer, slower moving satellites for whole-earth coverage (for example, 12 at 10,370 km in the case of the Odyssey system). Geostationary satellites (GEO), at the special height of around 36,000 km, have the benefit of requiring only three satellites for whole-earth coverage and require almost no tracking of satellite direction. The large altitudes in GEO systems lead to a very large free-space loss component. For example, the free-space loss for a geostationary satellite at 1.5 GHz would be around 186 dB at zenith. The transmit powers needed at both the satellite and mobile to

109

Figure 3.39 Megacell propagation geometry.

overcome this loss are excessive, so LEO and MEO systems are far more attractive for mobile communications, while GEO is more usually applied to fixed satellite links. For further details, see [52, 53]. In all orbits other than GEO, the satellite position changes relative to a point on the earth, so the free-space loss for a particular mobile position becomes a function of time. In the extreme case of a LEO system, an overhead pass might last only a few minutes, so the path loss will change rapidly between its maximum and minimum values (see, for example, Figure 3.40). Similarly, the motion of the satellite relative to the location of the mobile user will create significant Doppler shift, which will change rapidly from positive to negative as the satellite passes overhead. In the case illustrated in Figure 3.40, the maximum Doppler shift would be around ±37 kHz. This shift can be compensated for by retuning either the transmitter or receiver, and should not be confused with the Doppler spread, which arises from motion of the mobile relative to sources of multipath, which is of a much smaller size but which cannot be compensated for. The main local sources of propagation impairments in mobile-satellite propagation are trees, buildings, and terrain. These interact with wave propagation via the following mechanisms: reflection, scattering, diffraction, and multipath.

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Figure 3.40 Free-space loss for a circular LEO satellite orbit at an altitude of 778 km at 1.625 GHz. This range of values would be encountered over a period of around 7.5 minutes.

3.5.1 Shadowing and Fast Fading In mobile satellite systems, the elevation angle from the mobile to the satellite is much larger than for terrestrial systems, with minimum elevation angles in the range of 8–25°. Shadowing effects therefore tend to result mainly from the clutter in the immediate vicinity of the mobile. For example, when the mobile is operated in a built-up area, only the building closest to the mobile in the direction of the satellite is usually significant. As the mobile moves along the street, the building contributing to this process changes rapidly, so the shadowing attenuation may also change at a relatively high rate. By contrast, terrestrial macrocellular systems involve elevation angles on the order of 1° or less, so a large number of buildings along the path are significant (see Section 3.2). As a consequence of this effect, rapid and frequent transitions may occur between line-of-sight and nonline-of-sight situations in the satellite-mobile case, causing a variation in the statistics of fast fading that is closely associated with the shadowing process. See Figure 3.41 for a typical measured example of the variation of signal level with location in a suburban area; note that the fade depths are much greater during the obstructed periods. It is therefore most convenient to treat the shadowing and narrowband fast fading for satellite-mobile systems as a single, closely coupled process, in which the parameters of the fading (such as the Rice k -factor and local mean signal power) are time varying. In subsequent sections we examine the basic mechanisms of channel variation in mobile

111

Figure 3.41 Example channel variations measured in a suburban area at 1.5 GHz.

satellite systems, and then describe a number of models which can be used to predict these effects. 3.5.2 Local Shadowing Effects Roadside buildings are essentially total absorbers at mobile-satellite frequencies, so they can be regarded as diffracting knife edges. They can be considered to block the signal significantly when at least 0.6 of the first Fresnel zone around the direct ray from the satellite to the mobile is blocked (see Figure 3.42). Thus, shadowing may actually be less at higher frequencies due to the narrower Fresnel zones for a given configuration. Once

Figure 3.42 Shadowing by (a) buildings and (b) trees.

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shadowing has occurred, building attenuation may be estimated via the single knife edge diffraction formula given in Section 3.5.2. Tree shadowing also occurs primarily when the tree is contained within the 0.6 × first Fresnel zone region. In this case, however, the tree is not a complete absorber, so propagation occurs through the tree as well as around it. The attenuation varies strongly with frequency and path length. The simplest approach is to find the path length through the tree and calculate the attenuation based on an exponential attenuation coefficient in dB/m. Single-tree attenuation coefficients have been measured at 869 MHz, and the largest values were found to vary between 1 and 2.3 dB/m, with a mean value of 1.7 dB/m [54].

3.5.2.1 Local Multipath Effects As well as the existence of a direct path from the satellite to the mobile, reflection and scattering processes lead to other viable wave paths. These multiple paths interfere with each other, leading to rapid fading effects as the mobile’s position varies. The multipath may result from adjacent buildings, trees, or the ground. The level of the multipath is usually rather lower than that of the direct path, but it may still lead to significant fading effects, similar to those described in Chapter 2. Note that multiple scattering paths, such as those that reach the mobile via points x and y in Figure 3.43 are attenuated by two reflection coefficients and are therefore unlikely to be significant.

Figure 3.43 Multipath propagation.

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The multipath can also lead to wideband fading if the differential path lengths are sufficiently large. In practical systems, however, the dominant source of wideband effects may be the use of multiple satellites to provide path diversity (see Section 3.5.7). 3.5.3 Empirical Narrowband Models Models for narrowband propagation in mobile satellite systems are different from those used in terrestrial systems in two key ways. First, they include the excess path loss and shadowing effects as dynamic processes, along with the fast fading. Second, they rarely use direct deterministic calculation of physical effects, since this is not practical for predicting satellite coverage of areas exceeding tens of thousands of square kilometers. Instead they use statistical methods, although these may be based on either empirical or physical descriptions of the channel. Empirical models, particularly the empirical roadside shadowing (ERS) model, have been constructed for mobile satellite systems operated in areas characterized mainly by roadside trees. The ERS model is expressed as [55]: L (P , ␪ ) = −(3.44 + 0.0975␪ − 0.002␪ 2) ln P + (−0.443␪ + 34.76)

(3.62)

where L (P , ␪ ) is the fade depth exceeded for P percent of the distance traveled (decibels), at an elevation angle (degrees) to the satellite, where for elevation angles from 7° to 20°, (3.62) is used with ␪ = 20 deg. If the vehicle travels at constant speed, P is also the percentage of the time for which the fade exceeds L . This model applies only to propagation at L-band (1.5 GHz), elevation angles from 20° to 60°, and at fade exceedance percentages from 1% to 20%. The result is shown in Figure 3.44. The ERS model can then be extended up to 20 GHz using the following frequencyscaling function:

冋 冉√

L ( f 2 ) = L ( f 1 ) exp 1.5

冊册

1 1 − f 1 √f 2

(3.63)

where L ( f 2 ) and L ( f 1 ) are the attenuations in decibels at frequencies f 1 = 1.5 GHz and f 2 , with 0.8 GHz ≤ f 2 ≤ 20 GHz. This function is shown in Figure 3.45. The ERS model has also been extended to larger time percentages from the original distributions at L-band. The model extension is given by: L (P , ␪ ) =

冉冊

80 L (20%, ␪ ) × ln ln 4 P

(3.64)

for 80% ≥ P > 20%. Predictions from the extended model at 20 GHz for elevation angles from 20° to 60° are shown in Figure 3.46.

114

Figure 3.44 Predictions from empirical roadside shadowing model. The curves represent, from left, increasing elevation angle in steps of 10° from 10–60°.

Figure 3.45 Empirical roadside shadowing model frequency-scaling function.

115

Figure 3.46 Extended empirical roadside shadowing model at 20 GHz.

Foliage effects can also be modeled empirically using a model that was developed from mobile measurements taken in Austin, Texas, in 1995 [56]. The trees had no leaves in February and in May were in full foliage. A least-squares fit to the equal probability levels of the attenuations yielded the following relationship: L foliage = a + bL cno_foliage

(3.65)

where the constants are a = 0.351, b = 6.8253, and c = 0.5776 and where the model applies in the range 1 ≤ L no_foliage ≤ 15 dB and 8 ≤ L foliage ≤ 32 dB. 3.5.4 Statistical Models Statistical models give an explicit representation of the channel statistics in terms of parametric distributions, which are a mixture of Rice, Rayleigh, and lognormal components. Such models use statistical theory to derive a reasonable analytical form for the distribution of the narrowband fading signal and then use measurements to find appropriate values of the parameters in the distribution. These models all have in common an assumption that the total narrowband fading signal in mobile-satellite environments can be

116

decomposed into two parts: a coherent part, usually associated with the direct path between the satellite and mobile, and a diffuse part arising from a large number of multipath components of differing phases. The magnitude of the latter part is assumed to have a Rayleigh distribution. Thus, the multiplicative complex channel, ␣ , corresponding to all such models can be expressed as:

␣ = A c s c e j␾ + rs d e j(␪ + ␾ )

(3.66)

where A c is the coherent part, s c and s d are the shadowing components associated with the coherent and diffuse parts, respectively, and r has complex Gaussian distribution (that is, its magnitude is Rayleigh distributed). The simplest model of this form is the Rice distribution [1], which assumes that both components of the signal have constant mean power. More recent work such as that in [57–59] has generalized this model to account for the rapidly changing conditions associated with attenuation and shadowing of both the coherent and diffuse components that arise from mobile motion. The models are summarized in Table 3.12. Note that, as with terrestrial shadowing, the distribution of the mean power arising from s c and s d is widely assumed to be lognormal. These models can all be implemented within the structure shown in Figure 3.47. If the parameters of these models are appropriately chosen, they can provide a good fit to measured distributions over a wide range of environmental and operating conditions, although the Loo model is only really applicable to moderate rural situations. The Hwang model has been shown [59] to include the Rice, Loo, and Corazza models as special cases. Note, however, that Table 3.12 reveals that a further generalization of the Hwang model to allow correlation in the range [0,1] would allow still further generality. 3.5.4.1 Loo Model This model [57] is specifically designed to account for shadowing due to roadside trees. It is assumed that the total signal is composed of two parts; a line-of-sight component Table 3.12 Parametric Mobile-Satellite Channel Models

Model

Coherent Part

Correlation Between Coherent and Diffuse Parts

Rice [1] Loo [57] Corazza and Vatalaro [58] Hwang et al. [59]

Constant Lognormal Lognormal Lognormal

Zero Variable Unity Zero

Diffuse Part Constant mean power Rayleigh Constant mean power Rayleigh Lognormal-Rayleigh Lognormal-Rayleigh

117

Figure 3.47 Generative structure for analytical-statistical narrowband channel model. WGN is white Gaussian noise.

that is lognormally distributed due to the tree attenuation, plus a multipath component that has a Rayleigh distribution. Thus the total complex fading signal ␣ is given by:

␣ = de j␾ 0 + se j␾

(3.67)

where d is the lognormally distributed line-of-sight amplitude, s is the Rayleigh distributed multipath amplitude, and ␾ 0 and ␾ 0 are uniformly distributed phases. The pdf of the fading amplitude, r = | ␣ |, is too complicated to evaluate analytically, but it can be approximated by the Rayleigh distribution for small values and by the lognormal distribution for large values:

p (r ) =

冦

r

␴ 2m

冉 冊 冋

exp −

r2

2␴ 2m

1 (20 log r − ␮ )2 exp − 2␴ 0 20 log r √2␲␴ 0

for r Ⰶ ␴ m

册

(3.68) for r Ⰷ ␴ m

118

where ␴ m is the standard deviation of either the real or imaginary component of the multipath part, ␴ 0 is the standard deviation of 20 log d (dB), and is the mean of 20 log d (dB). An example prediction is shown in Figure 3.48 where the parameters of the Loo model have been chosen to fit the results from the ERS model at an elevation angle of 45 deg. Expressions for the level crossing rate and average fade duration are also given in [57] and it is found that these depend on the correlation between d and s, with the highest crossing rates occurring for low values of the correlation. 3.5.4.2 Corazza Model This model [58] can be seen as a development of the Loo approach, where both the direct path and multipath are lognormally shadowed, so that the channel amplitude is given by:

␣ = S (e j␾ 0 + se j␾ )

(3.69)

where S is the lognormal shadowing and other parameters follow the definitions of (3.67). The parameters of the model are then the Rice factor, k, and the lognormal mean ␮

Figure 3.48 Comparison of Loo model (dotted line) and ERS model (solid line) at 1.5 GHz and 45° elevation. Parameters are ␴ m = 0.3, ␴ 0 = 5 dB, and ␮ = 0.1 dB.

119

decibels and standard deviation ␴ L decibels. A wide range of environments can be modeled by appropriate choice of these parameters. The following empirical formulations for the parameters were extracted from measurements at the L-band in a rural tree-shadowed environment: k = 2.731 − 0.1074␪ + 2.774 × 10−3␪ 2

␮ = −2.331 + 0.1142␪ − 1.939 × 10−3␪ 2 + 1.094 × 10−3␪ 3

(3.70)

␴ L = 4.5 − 0.05␪ 3.5.4.3 Lutz Model In this model [60], the statistics of line-of-sight and nonline-of-sight states are modeled by two distinct states, which is particularly appropriate for modeling in urban or suburban areas where there is a large difference between shadowed and unshadowed statistics. The parameters associated with each state and the transition probabilities for evolution between states are empirically derived. These models permit time-series behavior to be examined and may be generalized to more states to permit a smoother representation of the transitions between LOS and NLOS conditions [61], or to characterize multiple satellite propagation [62]. For example, modeling in an environment that includes building blockage, tree shadowing, and line-of-sight conditions can be modeled via a three-state approach. In [60], the line-of-sight condition is represented by a ‘‘good’’ state, and the nonline-of-sight condition by a ‘‘bad’’ state (Figure 3.49). In the good state, the signal amplitude is assumed to be Rice distributed, with a k -factor that depends on the satellite

Figure 3.49 Two-state Lutz model of narrowband mobile-satellite fading. WGN is white Gaussian noise.

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elevation angle and the carrier frequency, so that the pdf of the signal amplitude is given by p good (r ) = p Rice (r ), where p Rice is a given in (10.30). The noncoherent (multipath) contribution has a classical Doppler spectrum. In the bad state, the fading statistics of the signal amplitude, r = | ␣ |, are assumed to be Rayleigh, but with a mean power S 0 = ␴ 2, which varies with time, so the pdf of r is specified as the conditional distribution p Rayl (r | S 0 ), and S 0 varies slowly with a lognormal distribution, p LN (S 0 ), with mean ␮ decibels and standard deviation ␴ L decibels, representing the varying effects of shadowing within the nonline-of-sight situation. The overall pdf in the bad state is then found by integrating the Rayleigh distribution over all possible values of S 0 , so that: ∞

冕

p bad (r ) =

p Rayl (r | S 0 ) p LN (S 0 )dS 0

(3.71)

0

The proportion of time for which the channel is in the bad state is the time-share of shadowing, A, so that the overall pdf of the signal amplitude is: p r (r ) = (1 − A) p good (r ) + Ap bad (r )

(3.72)

Transitions between states are described by a first-order Markov chain. This is a state-transition system, in which the transition from one state to another depends only on the current state, rather than on any more distant history of the system. These transitions are represented by the state transition diagram in Figure 3.50, which is characterized by a set of state-transition probabilities. The transition probabilities are: • • • •

Probability Probability Probability Probability

of of of of

transition transition transition transition

from from from from

good state to good state: p gg ; good state to bad state: p gb ; bad state to bad state: p bb ; bad state to good state: p bg .

Figure 3.50 Markov model of channel state.

121

For a digital communication system, each state transition is taken to represent the transmission of one symbol. The transition probabilities can then be found in terms of the mean number of symbol durations spent in each state: • •

p gb = 1/D g , where D g is the mean number of symbol durations in the good state; p bg = 1/D b , where D b is the mean number of symbol durations in the bad state. The sum of the probabilities leading from any state must sum to 1, so: p gg = 1 − p gb and p bb = 1 − p bg

(3.73)

Finally, the time-share of shadowing (the proportion of symbols in the bad state) is: A=

Db Db + Dg

(3.74)

The model can be used to calculate the probability of staying in one state for more than n symbols by combining the relevant transition probabilities. Thus: Probability of staying in ‘‘good’’ state for more than n symbols = p g (> n ) = p ngg

(3.75)

Probability of staying in ‘‘bad’’ state for more than n symbols = p b (> n ) = p nbb An example of the signal variations produced using the Lutz model is shown in Figure 3.51, where the two states are clearly evident in the signal variations. Typical parameters, taken from measurements at 1.5 GHz, are shown in Figure 3.52 (highway environment) and Figure 3.53 (city environment). This two-state model of the satellite-mobile channel is very useful for analyzing and simulating the performance of satellite-mobile systems. It is inaccurate in representing the second-order statistics of shadowing, however, since it assumes that the transition from LOS to NLOS situations is instantaneous. In practice this is not true, due to the smooth transitions introduced by diffraction and reflection effects, particularly at lower frequencies. One way of overcoming this is to introduce extra states, which represent intermediate levels of shadowing with smaller Rice k -factors than the LOS ‘‘good’’ state. It has been shown [61] that at least three states are needed in order to accurately model these transitions even at 20 GHz and the expectation is that even more would be needed for accurate modeling at L-band. Alternatively, the sharp transitions between states can be smoothed by filtering. In other work [63], the statistics of the fading process vary according to the

122

Figure 3.51 Example time-series output (signal level and shadowing state) from Lutz model. Parameters used are D g = 24m, D b = 33m, ␮ = −10.6 dB, ␴ L = 2.6 dB, k = 6 dB, f c = 1,500 MHz, and v = 50 km h−1.

environment, with several environment types (urban, open, suburban) distinguished as model states. 3.5.5 Physical-Statistical Models for Built-Up Areas Deterministic physical models of mobile-satellite propagation have been created and have produced good agreement with measurements over limited areas [64]. For practical predictions, however, an element of statistics has to be introduced. One approach is to use physical models to examine typical signal variations in environments of various categories and use these to generalize over wider areas. A more appropriate approach is to use physical-statistical models, which derive fading distributions directly from distributions of physical parameters using simple electromagnetic theory. This class of models is described in some detail here, as the modeling approach has been examined only slightly in the open literature. In modeling any propagation parameter, the aims of modeling are broadly similar. The key point is to predict a particular parameter with maximum accuracy, consistent with minimum cost in terms of the quantity and expense of the input data and in terms of the computational effort required to produce the prediction.

123

Figure 3.52 Highway parameters for Lutz model; x-axis is elevation angle.

Figure 3.53 City parameters for Lutz model; x-axis is elevation angle.

124

For empirical models, the input knowledge consists almost entirely of previous measurements that have been made in environments judged to be representative of practical systems. An approximation to these data, usually consisting of a curve fit to the measurements, is used for predictions. The input data are then fairly simple, consisting primarily of operating frequency, elevation angle, range, and a qualitative description of the environment (for example, rural or urban). Such models are simple to compute and have good accuracy within the parameter ranges spanned by the original measurements. However, because the models lack a physical basis, they are usually very poor at extrapolating outside of these parameter ranges. There is a classification problem involved in describing the environment, since an environment judged to be urban in some countries may be little more than a small town elsewhere. Additionally, the use of a curve-fitting approach implies that the real data will generally be considerably scattered around the predicted values and this represents a lower limit on the prediction accuracy. For example, predictions of loss are subject to an error resulting from the effects of shadowing and fading. The input knowledge used in physical models, by contrast, consists of electromagnetic theory combined with engineering expertise that is used to make reasonable assumptions about which propagation modes are significant in a given situation. Provided that the correct modes are identified, the theoretical approach is capable of making very accurate predictions of a wide range of parameters in a deterministic manner. The output that can be given is point-by-point rather than an average value, so the model can apply to very wide ranges of system and environment parameters, certainly well beyond the range within which measurements have been made. To make such predictions, however, the models may require very precise and detailed input data concerning the geometrical and electrical properties of the environment. This may be expensive or even impossible to obtain with sufficient accuracy. Also, the computations required for a full theoretical calculation may be very complex, so extra assumptions often have to be made for simplification, leading to compromised accuracy. Physical-statistical modeling is a hybrid approach that builds on the advantages of both empirical and physical models while avoiding many of their disadvantages. As in the physical model case, the input knowledge consists of electromagnetic theory and sound physical understanding. However, this knowledge is then used to analyze a statistical input data set, yielding a distribution of the output predictions. The outputs can still effectively be point by point, although the predictions are no longer linked to specific locations. For example, a physical-statistical model can predict the distribution of shadowing, avoiding the errors inherent in the empirical approach, although it does not predict what the shadowing value will be at a particular location. This information is usually adequate for the system designer. Physical-statistical models therefore require only simple input data such as input distribution parameters (such as mean building height, building height variance). The environment description is entirely objective, avoiding problems of subjective classification, and is capable of high statistical accuracy. The models are based on sound physical principles, so they are applicable over very wide parameter ranges.

125

Finally, by precalculating the effect of specific input distributions, the required computational effort can be very small. One example of a physical-statistical model has been used to predict the attenuation statistics of roadside tree shadowing, using only physical parameters as input [65]. This modeled the trees as consisting of a uniform slab whose height and width were uniformly distributed random variables. For a given direction from mobile to satellite, the mean and standard deviation of the path length through the block was calculated and used with an empirical model of specific attenuation through trees to calculate the mean and standard deviation of the tree attenuation. These values are then taken as the mean and standard deviation of the lognormal distribution in the Lutz structure (Figure 3.49). In Sections 3.5.5.2 and 3.5.5.3, two physical-statistical models for megacells operated in built-up areas are described [66, 67]. First, the basic physical parameters used in both these models are introduced. 3.5.5.1 Building Height Distribution The geometry of the situation to be analyzed is illustrated in Figure 3.54. It describes a situation in which a mobile is situated on a long straight street with the direct ray from the satellite impinging on the mobile from an arbitrary direction. The street is lined on both sides with buildings having heights that vary randomly. In the models to be presented, the statistics of building height in typical built-up areas will be used as input data. A suitable form was sought by comparing with geographical data for the city of Westminster and for the city of Guildford, United Kingdom (Figure

Figure 3.54 Geometry for mobile-satellite propagation in built-up areas.

126

3.55). The probability density functions that were selected to fit the data are the lognormal and Rayleigh distributions with parameters the mean value ␮ and the standard deviation ␴ b . The pdf for the lognormal distribution is: ln2(h b /␮ )

1 p b (h b ) = ⭈e h b √2␲ ␴ b

2␴ 2b

(3.76)

and for the Rayleigh distribution is: p b (h b ) =

hb

␴ 2b

h 2b 2

⭈ e 2␴ b

(3.77)

The best fit parameters for the distributions are shown in Table 3.13. The lognormal distribution is clearly a better fit to the data, but the Rayleigh distribution has the benefit of greater analytical simplicity.

Figure 3.55 Building height distribution for (a) city of Westminster and (b) city of Guildford.

Table 3.13 Best Fit Parameters for the Theoretical pdfs Log-Normal pdf City Westminster Guildford

Rayleigh pdf

Mean (␮ )

Standard Deviation (␴ b )

Standard Deviation (␴ b )

20.6 7.1

0.44 0.27

17.6 6.4

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3.5.5.2 Time-Share of Shadowing This model [66] estimates the time-share of shadowing [A in (3.72)] for the Lutz model [60] using physical-statistical principles. The direct ray is judged to be shadowed when the building height h b exceeds some threshold height h T relative to the direct ray height h s at that point. A can then be expressed in terms of the probability density function of the building height, p b (h b ) as: ∞

P s = Pr (h b > h T ) =

冕

p b (h b )dh b

(3.78)

hT

Assuming that the building heights follow the Rayleigh distribution, (3.77), this yields: ∞

A=

冕

hb

␴ 2b

冉 冊

exp −

h 2b

2␴ 2b

冉 冊

dh b = exp −

h 2T

2␴ 2b

(3.79)

hT

The simplest definition of h T is obtained by considering shadowing to occur exactly when the direct ray is geometrically blocked by the building face (a more sophisticated approach would account for the size of the Fresnel zone at that point). Simple trigonometry applied to this yields the following expression for h T : d m tan ␾ sin ␪ hT = hr = (w − d m ) tan ␾ hm + sin ␪ hm +

for 0 < ␪ ≤ ␲ (3.80) for −␲ < ␪ ≤ 0

Figure 3.56 compares this model with measurements of A versus elevation angle in city and suburban environments at L -band, taken from [60, 68, 69]. The model parameters are ␴ b = 15, w = 35, d m = w/2, h m = 1.5, and ␪ = 90°. 3.5.5.3 Time-Series Model Although the approach in the last section allowed one of the parameters in the Lutz model to be predicted using physical-statistical methods, the remainder of the parameters still has to be determined empirically. All of the statistical models described earlier assume that the lognormal distribution was valid for predicting the shadowing distribution. However, Chapter 9 shows that this distribution comes as a result of a large number of individual effects acting together on the signal. In the case being treated here, only a single building

128

Figure 3.56 Comparison of theoretical and empirical results for time-share of shadowing. Measurements include L- and S-band examples.

is involved, so the lognormal approximation is questionable. The model described here avoids this assumption and directly predicts the statistics of attenuation [67]. The power received in this case is predicted as a continuous quantity, avoiding the unrealistic discretization of state-based models such as [60]. The total received power for a mobile in a built-up area consists of the direct diffracted field associated with the diffraction of the direct path around a series of roadside buildings, plus a multipath component whose power is set by computing reflections from the buildings on the opposite sides of the street and from the ground. The direct field is given by: u o (P ) = u o (P )

再

j 1− 2

冦

∑

冋冉 冕

v x 2m

N

m=1

v x 1 = −∞, v x 2 = vy 1 =

√

e

−∞

√

−

i␲ 2 v 2 x dv

冊 冉冕

vy 2m

x

⭈

−

e

vy 1m

2(d 1 + d 2 ) (x 22 − x m ) ␭ d 1d 2

k (d 1 + d 2 ) ( y 2i − y m ), i = 1, 2 ␲ d 1d 2

i␲ 2 v 2 y dv

冊册冎

y

(3.81)

(3.82)

129

where x 22 defines the building height, y 21 defines the position of the left building edge, y 22 defines the position of the right building edge, d is the distance from the satellite to the building, and d 2 is the distance from the building to the mobile. This result is similar to the Fresnel integral formulation of single knife edge diffraction used earlier, but accounts for diffraction from around the vertical edges of buildings as well as over the rooftops. For the direct-diffracted field, d 2 ≡ d m and x 3 = h m . For the wall reflected path, (3.81) is again used, but applied to the image of the source, so d 2 = 2w − d m . For the groundreflected field x 3 = −h m . The satellite elevation angle and azimuth angles, and, respectively, are introduced using the following geometrical relationships, with R being the direct distance from mobile-satellite: y sat =

R cos ␾

√1 + tan

2

␪

x sat = R sin ␾ + h m d1 =

√R

2

2

cos ␾ −

(3.83) y 2sat

− d2

The formulation above is then used with a series of roadside buildings, randomly generated according to the lognormal distribution of (3.76), with parameters applicable to the environment under study, including gaps between the buildings to represent some open areas. Figure 3.57 illustrates measured and simulated time-series data for a suburban

Figure 3.57 Comparison of theoretical output with real outdoor measurement data from a satellite.

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environment at 18.6 GHz with the same sampling interval and with a 90° azimuth angle and a 35° elevation angle. The other model parameters were w = 16m, d m = 9.5m, open area = 35% of total distance, ␮ = 7.3m, and ␴ b = 0.26. For the building and ground reflections, the conductivity was set to 0.2 and 1.7 Sm−1 and the relative permittivity was set to 4.1 and 12, respectively. Qualitatively, the characteristics of the signal variation are similar, although the statistical nature of the prediction implies that the model should not be expected to match the predictions at any particular locations. Figures 3.58 and 3.59 illustrate comparison results for the first-order and secondorder statistics for the same satellite measurements as in Figure 3.57 and also for helicopter

Figure 3.58 Comparison of first-order statistics with real outdoor (a) satellite and (b) helicopter measurements.

Figure 3.59 Comparison of second-order statistics with real outdoor (a) satellite and (b) helicopter measurements.

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measurements [61]. The frequency for the helicopter measurements was 1.2 GHz, with a 90° azimuth angle and a 60° elevation angle. 3.5.6 Wideband Models Wideband measurement and modeling has received relatively little attention for LMS systems, partly because the delay spreads encountered are far smaller than in most terrestrial systems, rendering the channel essentially narrowband for most first-generation LMS systems. However, future systems will offer multimedia services requiring very large channel bandwidths and will use spread spectrum techniques to provide high reliability and capacity, increasing the significance of wideband effects. We saw previously that shadowing effects in mobile-satellite systems are dominated by the clutter in the immediate vicinity of the mobile. This is also true when considering wideband scattering, where the significant scatterers tend to contribute only relatively small excess path delays. The resulting wideband channel has been found to have an RMS delay spread of around 200 ns. The wideband channel model is then very similar to the terrestrial case, except that, in general, the tap gain processes may each be represented as instances of narrowband models such as those described in Sections 3.5.3, 3.5.4, and 3.5.5. Additionally, the first arriving tap will usually have a significant coherent component and hence a relatively high Rice k -factor compared to the later echoes. Several measurements have been conducted [69, 70]. The latter work has led to results for the variation of delay spread at the L- and S-bands in five different environmental categories at elevation angles from 15° to 18°. A tapped delay-line model has been created based on the analysis of a representative fraction of the data [71]. A more sophisticated structure has also been proposed [69] in which the earliest arriving echoes, which arise from scatterers within 200-m excess delay (600 ns) of the mobile, are assumed to have exponentially decreasing power with increasing delay. In this region the number of echoes is assumed to be Poisson distributed and the delays are exponentially distributed. Occasional echoes also occur with longer delays, with the delays being uniformly distributed up to a maximum of around 15 ␮ s. 3.5.7 Multisatellite Correlations When considering the effects of multiple-satellite diversity (or for more accuracy in the single-satellite case) it is crucial to consider the effects of the correlation between the fading encountered for satellites at different elevation and azimuth angles. Two satellites at the same elevation angle may exhibit very different outage probabilities, when, for example, one is viewed perpendicular to the direction of a building-lined street, while the other is viewed down the street and therefore much more likely to be unshadowed. One method of accounting for these effects is to generalize the Lutz model (Section 3.5.4.3) to four states, representing all the ‘‘good’’ and ‘‘bad’’ combinations of the two

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mobile-satellite channels, as shown in Figure 3.60 [62]. The transition probabilities can again be found using information from measurements and models of time-share of shadowing; it is also necessary, however, to have knowledge of the correlation between the shadowing states of the two channels. When two satellites are present, their locations are defined by two pairs of elevation and azimuth angles (␪ 1 , ␾ 1 ) and (␪ 2 , ␾ 2 ). The shadowing state of satellite i is defined as: Si =

再

0 1

if h b ≥ h r (bad channel state) if h b < h r (good channel state)

(3.84)

where (␪ i , ␾ i ) is used to find the corresponding value for h r . The correlation between these shadowing states is then defined by:

␳=

E [(S 1 − S 1 ) ⭈ (S 2 − S 2 )] 1 = ⭈ [E (S 1 S 2 ) − S 1 ⭈ S 2 ] ␴1 ⭈ ␴2 ␴1 ⭈ ␴2

(3.85)

where E [S ] = S . If ␳ is close to unity, the satellites suffer simultaneous shadowing always, and satellite diversity does not increase system availability significantly. If ␳ is close to zero, however, the satellites suffer simultaneous shadowing only rarely, and the system is available far more of the time than would be the case with a single satellite. Still better is the situation where ␳ is close to −1, and the availability of the satellites complements each other perfectly.

Figure 3.60 Four-state model of satellite diversity.

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Measurements of the correlation are available [73] that have used the results of circular flight-path measurements to derive a correlation function, which is then used to derive appropriate transition probabilities for the model. Other approaches are to derive the correlations using a fish-eye lens photographs [74] or physical expressions [72]. In all cases, the correlation encountered in built-up areas diminishes rapidly with increasing azimuth angle between the satellites, being sufficiently small for appreciable diversity gain above around 30‚ difference (see Figure 3.61). Negative correlations are also possible when the environment has a particular geometrical structure. Parameters such as the timeshare of shadowing can also be extracted from fish-eye lens photographs and a very close correspondence between measurements and predicted parameters has been found [75]. 3.5.8 Overall Mobile-Satellite Channel Model The overall mobile-satellite channel model is shown in Figure 3.62. There are two parts to the model: a satellite process, which includes effects between the satellite and the earth’s surface, and a terrestrial process, which accounts for all the effects in the vicinity of the mobile. The satellite process includes a delay of ␶ sg , arising from the total propagation path length, a total path loss (excluding shadowing) A, which includes a free-space and atmospheric loss component, and a Doppler shift through a frequency of f sg = ␻ sg /2␲ , which arises from the relative speed of the satellite and a point on the ground adjacent to the mobile.

Figure 3.61 Example prediction of correlation coefficient from [72].

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Figure 3.62 Complete mobile-satellite channel model.

The terrestrial process is modeled as a time-variant transversal filter (tapped delay line) representation of the wideband channel, with tap-gain processes r 1 (t ), r 2 (t ), . . . , r n (t ). Each of these processes may be modeled using any of the narrowband models described earlier in this chapter, so essentially the same parametric representation applies. Note that there are two quite distinct sources of Doppler in the megacell channel: One is a Doppler shift arising from satellite motion relative to the ground, whereas the other is a Doppler spread arising from motion of the mobile relative to the scatterers in the immediate vicinity. It is assumed in this structure that all waves arriving at the mobile are subject to the same Doppler shift, which is a good approximation for scatterers in the near vicinity of the mobile. 3.6 THE FUTURE This brief chapter provides a speculative and highly personal view of the likely developments in future understanding and treatment of the wireless communication channel. Although the fundamental properties of radio waves that arise from Maxwell’s equations have been well understood for almost a century, the application of these properties to practical wireless systems is an evolving field. The points raised in this chapter should indicate some of the key future challenges. 3.6.1 Intelligent Antennas To meet the changing needs of various wireless systems, antenna structures developed from the generic families of antennas described in the rest of this book are being developed on an ongoing basis. A more radical shift, however, is emerging from the possibilities of merging the functionality of digital signal processing with multiple antenna elements.

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This creates intelligent antennas, whose characteristics are varied over time to optimize the antenna characteristics with respect to specific system goals, such as coverage, capacity, or quality. Intelligent antennas include the diversity and adaptive antenna concepts as special cases, but they can also merge the concepts of equalizers to overcome wideband effects, adaptive matching to improve power delivery, and interference cancellation to enhance system capacity. Initially these concepts are being applied primarily at base stations, where the extra processing and relatively large numbers of antenna elements are relatively easily accommodated, but developments in compact multielement antenna structures and available signal processing power will enable intelligent antennas to be a standard feature of future mobile terminals as well. Ultimately, an understanding of how these structures operate in conjunction with the characteristics of the propagation channel will enable increases in user densities and channel data rates that are many orders of magnitude beyond those possible with today’s technology. 3.6.2 Multidimensional Channel Models For the most part, this chapter examines the variations of the channel in space, time, and polarization as separate topics. However, future systems will increasingly require knowledge of the joint behavior of channels with respect to these variables in order to optimize their performance, particularly to support large numbers of users with high data rates. A clear example is the joint angle-of-arrival/time-of-arrival scattering maps described in [1]. This will require the creation of new models that account for the correlations between various channel parameters in detail, and which permit the design, characterization, and verification of advanced processing systems such as time-space beamforming or multipleuse detectors. 3.6.3 High-Resolution Data The demand for high-resolution data for use as input to propagation predictions has increased substantially in recent years. This has coincided with the expansion in the use of small macrocells and microcells, which require a greater level of detail for useful propagation predictions. The data available include terrain, building height data, vector building data, land usage categories, and meteorological information to increasingly fine resolution and high accuracy. Nevertheless, the costs of such data are still prohibitively high for many applications when compared with making measurements. These costs are expected to fall over time as the demand increases further and the technology for acquiring such data becomes cheaper. 3.6.4 Analytical Formulations There is an ongoing need for the solution of basic electromagnetic problems in order to permit propagation models to be based on sound physical principles. These are essential

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in order to gain value from the high-resolution input data described above. Particular needs are in the consideration of three-dimensional problems involving diffraction and rough scattering from multiple obstacles. Such formulations will permit rapid evaluation compared to brute-force numerical techniques and allow predictions to make the best possible use of any available data. 3.6.5 Physical-Statistical Channel Modeling Section 3.5.5 described a physical-statistical propagation prediction methodology within the context of mobile satellite systems. These methods are also attractive in terrestrial applications, particularly when full deterministic input data may be too expensive for a given application, or when the channel is randomly time-varying due to the motion of vehicles, people, or other scatterers. It is expected that physical-statistical models will increasingly be seen as an appropriate compromise between the accuracy and applicability of deterministic physical models and the coarse but rapid results produced by empirical models. 3.6.6 Real-Time Channel Predictions In the past, system designers made predictions of propagation parameters and system performance as part of the initial design process for a new system. However, this approach requires that considerable fade margins be included, leading to system operation that is far from optimum at any particular point in time. Significant performance gains are possible if the parameters of the wireless system (such as power level, modulation and coding rates, antenna patterns, channel reuse) are permitted to evolve over time to meet changing constraints resulting from the behavior of the users and the propagation channel. This can best be done by allowing the system to maintain an evolving model of the propagation characteristics with which it can test the likely best choice of parameters. Such models are likely to take a very different form from those currently used in nonreal-time situations, because they will necessarily need to run much faster and also to integrate both theoretical predictions and measurements. 3.6.7 Overall The fast-moving developments in the field of wireless communications imply that the state of the art may not be described by this chapter by the time it is read. It is hoped, however, that by focusing on fundamental physical mechanisms and a wide range of wireless system types, the book has been able to give an indication of methodologies for analyzing and understanding antennas and propagation that will be useful for many years to come. For more information on the latest developments in the areas listed above the

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reader is invited and encouraged to visit the following web site: http://www.ee.surrey.ac.uk/ CCSR/AP/.

REFERENCES [1] Saunders, S. R., Antennas and Propagation for Wireless Communication Systems, New York: John Wiley, 1999. [2] Parsons, J. D., The Mobile Radio Propagation Channel, New York: John Wiley, 1992. [3] Egli, J. J., ‘‘Radio Propagation above 40MC over Irregular Terrain,’’ Proc. IRE, pp. 1383–1391, 1957. [4] Okumura, Y., et al., ‘‘Field Strength and Its Variability in VHF and UHF Land Mobile Radio Service,’’ Rev. Electr. Commun. Lab., Vol. 16, pp. 825–873, 1968. [5] Hata, M., ‘‘Empirical Formula for Propagation Loss in Land Mobile Radio Services,’’ IEEE Trans. Vehicular Technology, Vol. VT-29, pp. 317–325, 1980. [6] International Telecommunication Union, ‘‘ITU-R Recommendation P. 529-2: Prediction Methods for the Terrestrial Land Mobile Service in the VHF and UFH Bands,’’ Geneva, 1997. [7] COST-207 Management Committee, Digital Land Mobile Radio Communications, Final Report, Commission of the European Communities, Luxembourg, 1989. [8] Kozono, S., and K. Watanabe, ‘‘Influence of Environmental Buildings on UHF Land Mobile Radio Propagation,’’ IEEE Trans. Comm., Vol. 25, No. 10, pp. 1133–1143, 1977. [9] COST 231 Committee, Digital Mobile Radio: COST 231 View on the Evolution Towards 3rd Generation Systems, Final Report, European Commission-COST Telecommunications, Brussels, Belgium, 1998. [10] Ibrahim, M. F., and J. D. Parsons, ‘‘Signal Strength Prediction in Built-Up Areas,’’ Proc. IEE, Vol. 130F, Vol. 5, pp. 377–384, 1983. [11] Edwards, R., and J. Durkin, ‘‘Computer Prediction of Service Areas for VHF Mobile Radio Metworks,’’ Proc. IEE, Vol. 116, No. 9, pp. 1493–1500, 1969. [12] Huish, P. W., and E. Gurdenli, ‘‘Radio Channel Measurements and Prediction for Future Mobile Radio Systems,’’ British Telecom Tech. J., Vol. 6, No. 1, pp. 43–53, 1988. [13] Allsebrook, K., and J. D. Parsons, ‘‘Mobile Radio Propagation in British Cities at Frequencies in the VHF and UHF Bands,’’ IEEE Trans. Vehicular Technology, Vol. VT-26, No. 4, pp. 95–102, 1977. [14] Delisle, G. Y., et al., ‘‘Propagation Loss Prediction: A Comparative Study with Application to the Mobile Radio Channel,’’ IEEE Trans. Vehicular Technology, Vol. 26, No. 4, pp. 295–308, 1985. [15] Ikegami, F., T. Takeuchi, and S. Yoshida, ‘‘Theoretical Prediction of Mean Field Strength for Urban Mobile Radio,’’ IEEE Trans. Ant. Prop., Vol. 39, No. 3, pp. 299–302, 1991. [16] Vogler, L. E., ‘‘An Attenuation Function for Multiple Knife-Edge Diffraction,’’ Radio Science, Vol. 17, No. 6, pp. 1541–1546, 1982. [17] Saunders, S. R., and F. R. Bonar, ‘‘Prediction of Mobile Radio Wave Propagation over Buildings of Irregular Heights and Spacings,’’ IEEE Trans. Antennas and Propagation, Vol. 42, No. 2, pp. 137–144, February 1994. [18] Saunders, S. R., and F. R. Bonar, ‘‘Explicit Multiple Building Diffraction Attenuation Function for Mobile Radio Wave Propagation,’’ Electron. Lett., Vol. 27, No. 14, pp. 1276–1277, July 1991. [19] Walfisch, J., and H. L. Bertoni, ‘‘A Theoretical Model of UHF Propagation in Urban Environments,’’ IEEE Trans. Antennas & Propagation, Vol. AP-36, No. 12, pp. 1788–1796, December 1988. [20] Greenstein, L. J., et al., ‘‘Microcells in Personal Communication Systems,’’ IEEE Communications Magazine, Vol. 30, No. 12, pp. 76–88, 1992. [21] Sarnecki, J., et al., ‘‘Microcell Design Principles,’’ IEEE Communications Magazine, Vol. 31, No. 4, pp. 76–82, 1993. [22] Madfors, M., et al., ‘‘High Capacity with Limited Spectrum in Cellular Systems,’’ IEEE Communications Magazine, Vol. 35, No. 8, pp. 38–46, 1997.

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[23] Green, E., ‘‘Radio Link Design for Microcellular Systems,’’ British Telecom Tech. J., Vol. 8, No. 1, pp. 85–96, 1990. [24] Harley, P., ‘‘Short Distance Attenuation Measurements at 900 MHz and 1. 8 GHz Using Low Antenna Heights for Microcells,’’ IEEE J. Selected Areas in Communinications, Vol. 7, No. 1, pp. 5–11, 1989. [25] Chia, S. T. S., ‘‘Radiowave Propagation and Handover Criteria for Microcells,’’ British Telecom Tech. J., Vol. 8, No. 4, pp. 50–61, 1990. [26] Bultitude, R. J. C., and D. A. Hughes, ‘‘Propagation Loss at 1. 8GHz on Microcellular Mobile Radio Channels,’’ Proc. 7th IEEE Int. Symp. Personal, Indoor and Mobile Radio Communications, PIMRC ’96, Taipei, Taiwan, October 1996, pp. 786–790. [27] Xia, H. H., et al., ‘‘Radio Propagation Characteristics for Line-of-Sight Microcellular and Personal Communications,’’ IEEE Trans. Antennas and Propagation, Vol. 41, No. 10, pp. 1439–1447, 1993. [28] Dersch, U., and E. Zollinger, ‘‘Propagation Mechanisms in Microcell and Indoor Environments,’’ IEEE Trans. Vehicular Technology, Vol. 43, No. 4, pp. 1058–1066, 1994. [29] Goldsmith, A. J., and L. J. Goldsmith, ‘‘A Measurement-Based Model for Predicting Coverage Areas of Urban Microcells,’’ IEEE J. Selected Areas in Communications, Vol. 11, No. 7, pp. 1013–1023, 1993. [30] Maciel, L. R., and H. L. Bertoni, ‘‘Cell Shape for Microcellular Systems in Residential and Commercial Environments,’’ IEEE Trans. Vehicular Technology, Vol. 43, No. 2, pp. 270–278, 1994. [31] Erceg, V., A. J. Rustako, and P. S. Roman, ‘‘Diffraction Around Corners and Its Effects on the Microcell Coverage Area in Urban and Suburban Environments at 900-MHz, 2-GHz, and 6-GHz,’’ IEEE Trans. Vehicular Technology, Vol. 43, No. 3, Pt. 2, pp. 762–766, 1994. [32] Dehghan, S., and R. Steele, ‘‘Small Cell City,’’ IEEE Communications Magazine, Vol. 35, No. 8, pp. 52–59, Aug. 1997. [33] Berg, J. E., ‘‘A Recursive Method for Street Microcell Path Loss Calculations,’’ PIMRC ’95, Vol. 1, pp. 140–143, 1995. [34] Feuerstein, M. J., et al., ‘‘Path Loss, Delay Spread and Outage Models as Functions of Antenna Height for Microcellular System Design,’’ IEEE Trans. Vehicular Technology, Vol. 43, No. 3, pp. 487–498, 1994. [35] Arnold, H. W., D. C. Cox, and R. R. Murray, ‘‘Macroscopic Diversity Performance Measured in the 800 MHz Portable Radio Communications Environment,’’ IEEE Trans. Antennas and Propagation, Vol. 36, No. 2, pp. 277–280, 1988. [36] Keenan, J. M., and A. J. Motley, ‘‘Radio Coverage in Buildings,’’ British Telecom Tech. J., Vol. 8, No. 1, pp. 19–24, Jan. 1990. [37] International Telecommunication Union, ‘‘ITU-R Recommendation P. 1238: Propagation Data and Prediction Models for the Planning of Indoor Radiocommunication Systems and Radio Local Area Networks in the Frequency Range 900 MHz to 100 GHz,’’ Geneva, 1997. [38] Akerberg, D., ‘‘Properties of a TDMA Picocellular Office Communication System,’’ IEEE Global Telecommunications Conference Globecom ’88, Hollywood, pp. 1343–1349, 1988. [39] Turkmani, A. M. D., and A. F. Toledo, ‘‘Propagation Into and Within Buildings at 900, 1800 and 2300 MHz,’’ IEEE Vehicular Technology Conference, 1992. [40] Turkmani, A. M. D., J. D. Parsons, and D. G. Lewis, ‘‘Radio Propagation into Buildings at 441, 900 and 1400 MHz,’’ Proc. 4th Int. Conf. on Land Mobile Radio, 1987. [41] Walker, E. H., ‘‘Penetration of Radio Signals into Buildings in Cellular Radio Environments,’’ IEEE Vehicular Technology Society Conference, 1992. [42] Catedra, M. F., et al., ‘‘Efficient Ray-Tracing Techniques for Three-Dimensional Analyses of Propagation in Mobile Communications: Application to Picocell and Microcell Scenarios,’’ IEEE Antennas and Propagation Magazine, Vol. 40, No. 2, pp. 15–28, 1998. [43] Balanis, C. A., Advanced Engineering Electromagnetics, John Wiley & Sons, New York, 1989. [44] Honcharenko, W., et al., ‘‘Mechanisms Governing UHF Propagation on Single Floors in Modern Office Buildings,’’ IEEE Trans. Vehicular Technology, Vol. 41, No. 4, pp. 496–504, 1992. [45] Hashemi, H., ‘‘Impulse Response Modeling of Indoor Radio Propagation Channels,’’ IEEE J. Selected Areas in Communications, Vol. 11, No. 7, 1993.

139

[46] European Telecommunication Standards Institute, ‘‘Selection Procedures for the Choice of Radio Transmission Technologies of the Universal Mobile Telecommunications System (UMTS),’’ DTR/SMG-50402, 1997. [47] Hashemi, H., ‘‘The Indoor Radio Propagation Channel,’’ Proc. IEEE, Vol. 81, No. 7, pp. 943–967, 1993. [48] Hashemi, H., and D. Tholl, ‘‘Statistical Modeling and Simulation of the RMS Delay Spread of the Indoor Radio Propagation Channel,’’ IEEE Trans. Vehicular Technology, Vol. 43, No. 1, pp. 110–120, 1994. [49] Rappaport, T. S., S. Y. Seidel, and K. Takamizawa, ‘‘Statistical Channel Impulse Response Models for Factory and Open Plan Building Radio Communication System Design,’’ IEEE Trans. Communications, Vol. 39, No. 5, pp. 794–806, 1991. [50] Saleh, A. A. M., et al., ‘‘Distributed Antennas for Indoor Communication,’’ IEEE Trans. Communications, Vol. COM-35, No. 11, pp. 1245–1251, 1987. [51] Hashemi, H., ‘‘Impulse Response Modeling of Indoor Radio Propagation Channels,’’ IEEE J. Selected Areas in Communications, Vol. 11, No. 7, 1993. [52] Evans, B. G., ed., Satellite Communication Systems, 3rd ed., IEE, London, 1999. [53] Pattan, B., Satellite-Based Cellular Communications, McGraw-Hill, New York, 1998. [54] Vogel, W. J., and J. Goldhirsh, ‘‘Tree Attenuation at 869 MHz Derived from Remotely Piloted Aircraft Measurements,’’ IEEE Trans. Antennas and Propagation, Vol. AP-34, No. 12, pp. 1460–1464, 1986. [55] International Telecommunication Union, ‘‘ITU-R Recommendation P. 681-3: Propagation Data Required for the Design of Earth-Space Land Mobile Telecommunication Systems,’’ Geneva, 1997. [56] Goldhirsh, J., and W. J. Vogel, ‘‘Propagation Effects for Land Mobile Satellite Systems: Overview of Experimental and Modeling Results,’’ NASA Reference Publication 1274, Feb. 1992. [57] Loo, C., ‘‘A Statistical Model for a Land Mobile Satellite Link,’’ IEEE Trans. Vehicular Technology, Vol. VT-34, No. 3 , pp. 122–127, 1985. [58] Corazza, G. E., and F. Vatalaro, ‘‘A Statistical-Model for Land Mobile Satellite Channels and Its Application to Nongeostationary Orbit Systems,’’ IEEE Trans. Vehicular Technology, Vol. 43, No. 3, Pt. 2, pp. 738–742, 1994. [59] Hwang, S., et al., ‘‘A Channel Model for Nongeostationary Orbiting Satellite System,’’ 47th IEEE International Vehicular Technology Conference, Phoenix, Arizona, May 5–7, 1997. [60] Lutz, E., et al., ‘‘The Land Mobile Satellite Communication Channel-Recording, Statistics and Channel Model,’’ IEEE Trans. Vehicular Technology, Vol. 40, No. 2, pp. 375–385, May 1991. [61] Ahmed, B., et al., ‘‘Simulation of 20 GHz Narrowband Mobile Propagation Data Using N-state Markov Channel Modeling Approach,’’ 10th International Conference on Antennas and Propagation, Edinburgh, Apr. 14–17, 1997, pp. 2.48–2.53. [62] Lutz, E., ‘‘A Markov Model for Correlated Land Mobile Satellite Channels,’’ Int. J. Satellite Communications, Vol. 14, 1996. [63] Vucetic, B., and J. Du, ‘‘Channel Model and Simulation in Satellite Mobile Communication Systems,’’ IEEE Trans. Vehicular Technology, Vol. 10, pp. 1209–1218, 1992. [64] van Dooren, G. A. J., et al., ‘‘Electromagnetic Field Strength Prediction in an Urban Environment: A Useful Tool for the Planning of LMSS,’’ IMSC ’93, June 16–18, 1993, pp. 343–348. [65] Barts, R. M., and W. L. Stutzman, ‘‘Modeling and Measurement of Mobile Satellite Propagation,’’ IEEE Trans. Antennas and Propagation, Vol. 40, No. 4, pp. 375–382, April 1992. [66] Saunders, S. R., and B. G. Evans, ‘‘A Physical Model of Shadowing Probability for Land Mobile Satellite Propagation,’’ Electron. Lett., Vol. 32, No. 17, pp. 1548–1589, Aug. 15, 1996. [67] Tzaras, C., B. G. Evans, and S. R. Saunders, ‘‘A Physical-Statistical Analysis of the Land Mobile Satellite Channel,’’ Electron. Lett., Vol. 34, No. 13, pp. 1355–1357, 1998. [68] Parks, M. A. N., B. G. Evans, and G. Butt, ‘‘High Elevation Angle Propagation Results Applied to a Statistical Model and an Enhanced Empirical Model,’’ Electron. Lett., Vol. 29, No. 19, pp. 1723–1725, Sep. 1993. [69] Jahn, A., H. Bischl, and G. Heiss, ‘‘Channel Characterization for Spread-Spectrum Satellite-Communications,’’ IEEE Fourth Int. Symp. Spread Spectrum Techniques & Applications, Vols. 1–3, Chap. 261, pp. 1221–1226, 1996.

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[70] Parks, M. A. N., et al., ‘‘Simultaneous Wideband Propagation Measurements for Mobile Satellite Communications Systems at L- and S-Bands,’’ Proc. 16th Int. Communications Systems Conference (ICSSC96), Washington, D.C., Feb. 25–29, 1996, pp. 929–936. [71] Parks, M. A. N., S. R. Saunders, and B. G. Evans, ‘‘Wideband Characterisation and Modeling of the Mobile Satellite Propagation Channel at L and S Bands,’’ 10th Int. Conference on Antennas and Propagation, Edinburgh, Apr. 14–17, 1997, pp. 2.39–2.43. [72] Tzaras, C., S. R. Saunders, and B. G. Evans, ‘‘A Physical-Statistical Propagation Model for Diversity in Mobile Satellite PCN,’’ 48th IEEE Int. Vehicular Technology Conference, Ottawa, Canada, May 18–21, 1998, pp. 525–529. [73] Bischl, H., M. Werner, and E. Lutz, ‘‘Elevation-Dependent Channel Model and Satellite Diversity for NGSO S-PCNS,’’ Proc. IEEE 46th Vehicular Technology Conference, Vols. 1–3, Chap. 385, pp. 1038–1042, 1996. [74] Meenan, C., et al., ‘‘Availability of First Generation Satellite Personal Communication Network Service in Urban Environments,’’ Proc. IEEE Int. Vehicular Technology Conference, 1998, pp. 1471–1475. [75] Lin, H. P., R. Akturan, and W. J. Vogel, ‘‘Photogrammetric Prediction of Mobile Satellite Fading in Roadside Tree-Shadowed Environment,’’ Electron. Lett., Vol. 34, No. 15, 1998, pp. 1524–1525.

Chapter 4 Antennas for Base Stations Yoshihide Yamada, Yoshio Ebine, Masayuki Nakano, Anders Dernaryd, Bjorn Johannisson, and Martin Johansson

4.1 BASIC TECHNIQUES FOR BASE STATION ANTENNAS 4.1.1 System Requirements [1] The role of antennas in mobile communication systems is to establish a radio transmission line between radio stations, at least one of which is moving. There are two types of mobile communication systems: one where a transmitter and receiver communicate directly, and the other where they communicate through a base station. It is the latter type that has advanced around the world in recent years. Examples include automobile telephone systems, portable telephone systems, and multichannel access (MCA) systems for private use. Automobile telephone and portable telephone systems adopt a cellular structure, and the relation between system requirements and the necessary antenna technology is illustrated in Figure 4.1. In order for the base station to communicate with the mobile station located in the service area, radio wave energy must be radiated uniformly inside the area. Moreover, antenna gain should be as high as possible. Since the width of the service area is already specified, antenna gain cannot be increased by narrowing the beam in the horizontal plane. Therefore, it is necessary to narrow the antenna beam in the vertical plane to increase gain; a vertically arranged linear array antenna is effective to achieve this. Normal cellular systems use antennas with a gain from 7 to 15 dBd as the base station antennas. Antenna heights of these antennas are from 3m to 5m. So, slender mechanical designs are required for the ease of installations and lightning wind pressures. 141

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Figure 4.1 System requirements and antenna technologies.

In order for the base station antenna to communicate with many mobile stations simultaneously, multiple channels must be handled. This requires wide-frequency characteristics and a function for branching and/or combining the channels. For example, the base station of a Japanese cellular system using the 800-MHz band uses one antenna for both transmitting and receiving. The required bandwidth of the antenna is more than 7% where the specified voltage standing wave ratio (VSWR) is less than 1.5. Moreover, if the antenna is shared by several systems, wider antenna frequency bandwidth is required. In Table 4.1, assigned frequency bands for mobile communication systems in the Japanese Radio Regulation allocations are shown [2]. Frequency bands are grouped into 800, 1,500, and 2,000-MHz bands. Multifrequency band antennas are requested. Historical trends of base station antennas and typical multifrequency antenna elements are described in Section 4.1.2. Due to the rapid growth of demand, the lack of communication channels has become a serious problem for metropolitan areas in the United States, Europe, and Japan, and Table 4.1 Frequency Bands for Mobile Communications in Japan Frequency Band (MHz)

Assigned Frequency (MHz)

800 1,500 2,000

810–850, 860–901, 915–950 1,429–1,516 1,710–2,025, 2,110–2,200, 2,500–2,690

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technologies for effective frequency reuse are strongly needed. Although the cellular system has an advantage in terms of reusing frequency, its efficiency significantly depends on the radiation pattern of the base station antenna. Technologies for main beam tilting and beam shaping have been developed and effectively contribute to frequency reuse. The technologies are described in detail in Section 4.1.3. One of the most common features of mobile communication is that the base station and the mobile station do not fall within line of sight of each other. Moreover, the mobile station must move within a complex propagation environment. As a result, fading occurs constantly at the base and mobile stations, and the receiving levels may fluctuate by 10 dB or more. If system design takes the minimum receiving level into account, the load on the devices is excessive and system cost becomes too high. One technology for overcoming fading is diversity reception, which has been studied since the 1960s. Its effectiveness has been confirmed both experimentally and theoretically. In Section 4.1.4, diversity technologies are described. 4.1.2 Types of Antennas 4.1.2.1 Historical Trends of Base Station Antennas The historical trends of base station antennas in Japan are shown in Figure 4.2. The first base station antenna is the omnidirectional type. Four radiating elements are arranged around the vertical axis and are combined in order to achieve the omnidirectional radiation pattern. The antenna height is 5,700 mm and the diameter of the cylinder is 300 mm. Antenna gain is 15 dBd. In the high capacity system, 120 beamwidth in the horizontal plane is achieved for 3-sector zone use. As for radiating elements, printed dipole antennas are employed. Excitation coefficients of radiating elements are designed in order to achieve low sidelobe characteristics. Moreover, the beam tilting angle can be settled by selecting beam tilting panels in the beam tilting box. In digital and analog systems, radiation elements are replaced by the dual frequency band elements. In the 3G system, triple frequency band elements are employed. 4.1.2.2 Types of Radiation Elements Typical radiation element antennas used in array antennas of base stations are shown in Figure 4.3(a–c). Figure 4.3(a) is a fundamental printed dipole antenna. The height of this antenna is about a half wavelength. This antenna has the omnidirectional radiation pattern in the horizontal plane and the letter 8 radiation pattern in the vertical plane. As for bandwidth of this antenna, about 20% relative bandwidth of VSWR less than 2 is achieved. Figure 4.3(b) is a dual frequency band antenna. A parasitic element is electromagnetically coupled to the printed dipole antenna and achieves high frequency operation. Figure 4.3(c) is a triple frequency band antenna. The printed dipole antenna is composed to operate in

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Figure 4.2 Historical trends in base station antenna technologies.

dual frequency band and a parasitic antenna operates in the third frequency band. When these antennas are used as element antennas of an array antenna of a base station, a small reflector is attached behind these elements in order to achieve the sector beam. 4.1.3 Radio Zone Design 4.1.3.1 Sector Zone [3] In Figure 4.4, beam widths of base station antennas for a 6-sector zone and a 3-sector zone, respectively, are shown. Ideal beam shapes are shown in shadow areas. In actual radiation patterns, radiation levels are decreased about 3 dB at the zone edges. In Figure 4.5(a, b), subscriber capacities carried by up- and downlinks in 3- and 6-sector zones are shown, respectively. In the case of the 3-sector zone, subscriber capacity becomes the maximum value at the beam width of about 60 degrees. Sixty degrees is one-half of the illuminating area of 120 degrees. This is owing to the maximum antenna gain of the base station antenna. In the case of the 6-sector zone, subscriber capacity becomes the maximum

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Figure 4.3 (a–c) Typical radiation elements of base station antennas.

value at the beam width of about 30 degrees. This case is also owing to the maximum antenna gain of the base station antenna. 4.1.3.2 Beam Tilt [4] The principal idea of the beam tilt-down technique is to tilt the main beam in order to suppress the direction level toward the reuse cell and to increase (C/N)ANT. In this case, the carrier level also decreases in the zone edge. However, the interference level decreases more than the carrier level, so the total (C/N)ANT increases. This is an advantage from the viewpoint of system design, and this technique is used in most cellular systems in the world. Figure 4.6 shows the comparison between antennas with and without beam tilt, which verify the effectiveness of the beam tilt technique [5]. It can be easily understood from this figure that the distance from the base station inside which the interference level exceeds the threshold level of the system can be significantly reduced. In Japan, beam tilt is achieved electrically by adjusting the excitation coefficient of the array, while in Europe it is mainly achieved mechanically.

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Figure 4.4 (a, b) Beam width in sector zones.

Figure 4.5 (a, b) Subscriber capacity carried by links.

4.1.4 Diversity 4.1.4.1 Effect of Reception Diversity [6] The effect of reception diversity of the base station was first reported in 1965 by [7]. They showed that fading reduction could be achieved by placing two antennas approximately 10 wavelengths apart in the horizontal plane. Figure 4.7 shows the cumulative probability

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Figure 4.6 Beam tilt effect to reduce the frequency reuse distance.

Figure 4.7 Diversity effect versus correlation coefficient.

of received level using either one isolated antenna or two antennas with a correlation coefficient of 0, 0.5 and 0.8, respectively [8]. From this it can be understood that the received level at a probability of 1% with the diversity antenna is larger by 8 dB than that with a single antenna. Although two or more ports are necessary to carry out reception diversity, it significantly reduces fading. As a result, the transmitting power of the mobile

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station is reduced, and the quality of the transmission is ensured. This is a great advantage from the total system point of view. Reception diversity in the base station has been in commercial use in AMPS since 1982 in the United States and in NTT’s Large-Capacity System since 1985 in Japan. Since the theory of diversity reception is discussed in detail in Chapter 2 of this book, as well as in other publications listed in the reference section, only those items pertaining to the design of base stations are described in this section. 4.1.4.2 Configuration of Base Station Diversity Antennas [9] Many versions of base station antennas are in use in commercial systems in the United States, England, and Japan [10]. In each base station antenna, sector beam antennas with a 3-dB beam width of 120 degrees are used. Diversity antennas in the United States and Japan are arranged at angular increments of 120 degrees; antennas in England are composed of 6-sector beam antennas with a 3-dB beam width of 60 degrees. At the border line of the sector zone, the port with the higher receiving level is chosen. Typical diversity antenna configurations mostly used in the commercial systems are space diversity and polarization diversity. In the space diversity, two base station antennas are placed in the horizontal plane with a spacing d. Figure 4.8(a, b) shows the relationship between antenna spacing and correlation coefficient inn urban and suburban areas, respectively [11]. Antenna heights are 120m (䊐) and 5m (•) in (a) and 65m in (b). It can be understood from this figure that the antenna spacing should be larger than five wavelengths in urban areas to achieve a correlation coefficient of less than 0.6, while more than 20 wavelengths are required in suburban areas. It is also understood that the correlation coefficient increases with the antenna height. In this figure, the calculated values of the correlation coefficient are shown in curved lines. Here, Sh is the standard deviation of the incoming wave profile. By comparing measured and calculated values,

Figure 4.8 (a, b) Relation between correlation coefficient and antenna spacing.

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the values of Sh are supposed to be about 1 to 2 degrees. Next, Figure 4.9 shows the polarization diversity antenna configuration [12]. This antenna has vertical and horizontal radiating elements separately. Array elements are grouped into subarrays and feed phases to subarrays are differentiated in order to achieve electrical beam tilt. 4.1.4.3 Polarization Diversity Effect [13] PDC (TDMA) System Base Evaluation Based on the concept above, applicability of polarization diversity in the cellular phone system was reexamined. A number of tests were conducted [14–17] to confirm the effect

Figure 4.9 Configuration of the polarization diversity antenna.

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of polarization diversity, typical results of which will be described here. A handy phone was held by a test-person and attached to his ear at 60 degrees from zenith to transmit signals in the field. He was instructed to walk around a circle for a while at each location to vary the antenna orientation. For measuring the received signals, the Random Field Measurement Method [18] was employed. The received signals from both ports at the base station diversity antennas were switched at a speed of 300 Hz and fed to a receiver to record the fading signals virtually at the same time. Figure 4.10(a) shows the cumulative distribution of received level at a base station using space diversity, while the hand-held phone transmitted signals from a line-of-sight location in the base station coverage. The two solid lines on the left-hand side show the received levels at the two separate vertical ports, and the dotted line shows the cumulative distribution of selection-combine diversity. Figure 4.10(b) is the case for line-of-sight location with polarization diversity reception. The two solid lines show the distribution of output levels of vertical and horizontal ports, and the dotted line again shows that of selection-combine diversity. As can be seen, the polarization diversity gain is 12.0 dB at the 1% level of cumulative distribution whereas that of space diversity is 6.2 dB, suggesting the effectiveness of polarization diversity. Similar results were obtained for the case where the mobile station transmitted signals from the out-of-sight area. The results are shown in Figure 4.10(c, d). Although the effect is not so significant as that of line-of sight case, diversity gain still is greater for polarization diversity than space diversity.

Figure 4.10 (a–d) Cumulative distribution of received level.

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Evaluation of IS-95(CDMA) System Performance [19] In order to verify the effectiveness of the new polarization diversity system, a series of measurements was performed in a suburban area before the CDMA system was put into commercial service. Comparisons were made between the polarization diversity and the space diversity when the mobile station used a vertical dipole and a tilted dipole for the transmitter. The received level at a base station without diversity was used as a reference for evaluation. In the CDMA system, when the received signal level at the base station in the uplink is higher than the specified value, the transmitting power of mobile station is reduced by the base station signal. Therefore, if the transmitting power of the mobile station, which employs the diversity system, is lower than that without the diversity system, the system is judged to have some diversity gain. In the test environment, there existed only one base station and also only one mobile station, thus no interference problem was observed. Hence, the reduced value of the transmit power could be taken as a quality index of CDMA performance, instead of Eb /N0 which should be used for a case where a large number of mobile stations exists [20]. Figure 4.11(a, b) show the reduced values of mobile transmitting power in the case of space diversity (SD) and polarization diversity (PD) condition. The reduced values indicate a decreased level in the case of diversity conditions compared to the nondiversity case. In Figure 4.11(a), a vertical dipole was used at a test mobile. Measurements were taken with a test mobile in the field traveling in the longitudinal and transversal directions to the base station receive antenna at a driving speed of 40 and 6 km/hr, respectively. Measurements were taken both in the bore-sight area of the main beam and in the 3-dB-down beam edge area of the base station receive antenna in order to take care of the base station antenna performance effect. As seen in the figure, slightly better performance is observed in the case of space diversity than polarization diversity. In Figure 4.11(b), similar measurements were taken in the case of the mobile antenna tilted by 60 degrees from zenith. This situation is the situation of the average cellular user’s manner in using a phone. The mobile transmit power was significantly more reduced (i.e., 3 to 8 dB) for the polarization diversity than for the space diversity, indicating better receive signal performance. 4.2 DESIGN AND PRACTICE OF JAPANESE SYSTEMS 4.2.1 Multiband Antennas 120° Beam Width Antennas The frequency bands used for cellular radio systems in Japan are 0.9, 1.5, and 2.0 GHz. The base station antennas are required to work at multifrequency bands in order to reduce the installation space and cost. At the first stage, the dual band antenna has been realized.

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Figure 4.11 (a, b) Reduced values of CDMA mobile transmitting power in diversity conditions.

Two types of radiation elements are developed. The first type is composed of the radiation element and the parasitic element [21]. The other type is composed of two radiation elements corresponding to the dual frequency bands [22]. Figure 4.12 shows the antenna structure for the multi (triple) frequency bands. A printed dipole antenna (black region) is used for the 0.9-GHz band. This antenna is also utilized in the 1.5-GHz band. The impedance matching is attained by adjusting the strip line placed on the back side [23]. For the 2.0-GHz band, the parasitic element is attached in front of the printed dipole antenna [24]. Figure 4.13 shows the return loss with respect to the frequency. I can be observed in this figure that the required return loss of less than −14 dB is satisfied. Figure 4.14 shows the beam width with respect to the frequency. Almost 120° beam width for the frequency range of 0.8-GHz band to the 2.0-GHz band is observed. Figure 4.15 shows

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Figure 4.12 Configuration of the multiband antenna.

Figure 4.13 Return loss of the multiband antenna.

another antenna structure that has the multifrequency elements in one plane. The 1.5-GHz band element is constituted as the inner part of the 0.8-GHz printed dipole antenna. The 2.0-GHz band element is printed on the same plane as the 0.8/1.5-GHz elements [25]. Different Beam Width Antennas Figure 4.16 shows the multifrequency band antenna having beam width of 60° (2 GHz) and 120° (0.9/1.5 GHz) in the horizontal plane [26]. The 0.9/1.5-GHz antenna is composed

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Figure 4.14 Beam width of the multiband antenna.

Figure 4.15 Configuration of another multiband antenna.

on the center plate. The 0.9-GHz element is fed and the 1.5-GHz element is parasitic (#1). The feed point is placed at the bottom of the plate. For the 2-GHz antenna, parallel two elements are introduced. Each element is composed of one fed element and two parasitic elements (#2 and #3). All elements share one reflector for radiation. The edge of the reflector is bent at right angle, extending with the length of 20 mm. The distance from the reflector to the antenna of 0.9/1.5-GHz bands is 70 mm and that of the antenna of the 2.0-GHz band is 75 mm. The width of the reflector is 140 mm. In Figure 4.17(a–c),

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Figure 4.16 Multiband antenna having different beam widths.

Figure 4.17 (a–c) Radiation patterns in the horizontal plane.

radiation patterns in the horizontal plane are shown. At 0.9- and 1.5-GHz bands, the beam widths of 120° (for the 3-sector radio zone) are achieved. At the 2.0-GHz band, the beam width of 60° (for the 6-sector radio zone) is achieved. Effects of parasitic elements #2 and #3 are ensured. Next, the return loss characteristics in the 0.9- and 1.5-GHz bands are shown in Figure 4.18. Return losses less than −15 dB are achieved in both bands. In Figure 4.19, return losses of less than −15 dB is achieved at 2.0-GHz bands. The parasitic elements (#3) play an important roll in adjusting impedance characteristics.

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Figure 4.18 Return loss in 0.9- and 1.5-GHz bands.

Figure 4.19 Return loss in the 2.0-GHz band.

In Figure 4.20, another antenna configuration having a different beam width is shown [27]. Figure 4.20(a) shows the fundamental unit. For the 2.0-GHz band, two printeddipole arrays with reflectors are arranged side by side. Each array produces 60° beam. In this configuration, two 60° beams are produced in 60° separation. At the joint portion of the 2.0-GHz arrays, the 0.9/1.5-GHz bands antenna is located. Hence, two reflectors are used as the reflector of this antenna. The 0.9/1.5-GHz band antenna produces one

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Figure 4.20 (a, b) Another antenna configuration having different beam width.

beam having 120° beam. In Figure 4.20(b), the commercial antenna configuration is shown. Two sets of the fundamental unit are employed. In Figure 4.21(a, b) radiation patterns in the horizontal plane are shown. At the 0.9/1.5-GHz bands, excellent 120° beams are achieved. At the 2.0-GHz band, two 60° beams are achieved. 4.2.2 Remote Beam Tilting System It is well known that beam tilting in the vertical plane for base station antennas can reduce interference [28]. Figure 4.22 illustrates a block diagram of a remote beam tilting system for the beam tilting, which includes 12 accommodation boxes. There are seven phase shifters and a motor in the accommodation box. The beam tilting angle is determined by adjusting the phase shifters to array elements. Adjustment control is achieved remotely in order to accomplish the beam tilt quickly. Command of beam tilting is set up at the remote control station (RCS), from which the command signals are transmitted to the

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Figure 4.21 (a, b) Radiation patterns in the horizontal plane.

control unit (CONT) by the public telephone line. The CONT relays the base band signal to the signal relay unit (SRU). Moreover, the CONT supplies electric power to the motor drive unit (MDU). In the accommodation box, seven phase shifters are operated at once by the MDU. The remote beam tilting system in IMT-2000 for the 2.0-GHz band can tilt the beam direction from 0° to 10° with 0.5° step. Recently the new mechanical phase shifters have been introduced for base stations. Avoiding intermodulation is considered an important electric performance. So, configurations including metallic contact parts are avoided in the feed line construction. In Figure 4.23(a, b), configuration of the rotational phase shifter is shown. It is composed of three arc-shaped strip lines constituted on the dielectric substrate, the three arms feed line and the feed line also shown in Figure 4.23(a). Metallic contact between arc lines, arms and feed line are avoided and capacitive couplings are designed to work as electrical phase shifters. At the arc lines, the maximum line length is determined by the required beam tilting angle (maximum feed phase). Figure 4.23(b) shows a practical model. Another type of phase shifter shown in Figure 4.24(a, b) has been developed and practically used. The phase shifter is composed of two meander lines loaded the dielectric plates of high permittivity (⑀ r = 50 ∼ 100) as shown in Figure 4.24(a). Two dielectric plates are composed to put meander lines between plates. These dielectric plates are attached to the metal ground plane and are moved simultaneously. In accordance with the movement of the two dielectric plates, the equivalent permittivity of meander lines is varied. So, phases of output ports #1 and #2 are varied [29]. Figure 4.24(b) shows a practical model. 4.2.3 Antennas for Radio Blind Areas Radio blind areas in the cellular systems can be classified into two groups: open areas and closed areas. The open radio blind areas are such areas where radio waves greatly

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Figure 4.22 Remote beam tilting system.

attenuate due to the shadowing effect by mountains, buildings, and so forth. The closed blind areas are such areas where radio waves can hardly reach because of closed environments such as inside a tunnel, inside an underground street, or inside a building, and so forth. In the open radio blind areas, the booster systems, which amplify weak signals in relay systems, have been introduced to provide signals of proper level for the open radio blind areas so that the blind areas can be included in the service areas. Mutual coupling between the antenna to the base station access (BS-ANT) and the antenna to the mobile station access (MS-ANT) should be as low as possible to prevent the loop oscillation. Mutual coupling needs to be −90 dB when the propagation margin is −10 dB and the amplifier gain coefficient is about 80 dB. So, the front-back ratio (F/B) and the frontside ratio (F/S) of both BS-ANT and MS-ANT are required to be as small as possible and they are usually designed between 25 and 30 dB. A typical BS-ANT (MS-ANT) is shown in Figure 4.25. The antenna is composed of 16 patch antenna elements. Array element excitations have a binominal distribution. The antenna gain of 17 dBi is achieved. In order to reduce mutual coupling, the edge currents of the array are reduced by using chokes [30]. Mutual coupling characteristics

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Figure 4.23 (a, b) Configuration of the rotational phase shifter.

Figure 4.24 (a, b) Configuration of the sliding phase shifter.

are shown in Figure 4.26, where d denotes antenna spacing in the back-to-back arrangement. When the antenna spacing d is greater than 3␭ , mutual coupling can be reduced to be as low as −100 dB at the 2-GHz band. As for excitation levels of array elements, the elements #6, #7, #10, and #11 are excited with an amplitude of 1.0; the elements #2, #3, #5, #8, #9, #12, #14, and #15 are excited with an amplitude of 0.33; and the elements #1, #4, #13, and #16 are excited with an amplitude of 0.11.

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Figure 4.25 Configuration of booster antenna.

Figure 4.26 Mutual coupling of booster antenna.

In the closed radio blind areas, radio-on-fiber system (ROB)—which is a system where radio signals are transmitted without converting the RF frequency through the optical fiber system—has been employed. Radio signals transmitted through the optical fiber system are divided into antennas, which are distributed in the blind areas so that signals can be transmitted to all the blind areas. The ROB systems have been employed in both the IMT-2000 system (the 2.0-GHz band) and PDC systems (the 0.9-GHz band and the 1.5-GHz band) in Japan [31]. The antennas used in the long tunnel are required to have a bidirectional pattern in the horizontal plane, high antenna gain, and low profile for its installation on the side wall. Figure 4.27 shows the low profile notch antenna, which consists of anti-phase feed with the parasitic elements on the metal ground plane. The antenna gain is around 10 dBi at the 2.0-GHz band. At the 0.9/1.5-GHz band, gain is 3∼5 dBi [32].

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Figure 4.27 Low profile antenna for long tunnel use.

The antenna configuration used in the underground market is shown in Figure 4.28. Antennas for cellular systems (0.9, 1.5, and 2.0 GHz) and the pager system are shown. As for the 2.0-GHz band antenna, the top-loaded monopole antenna with the double ground plane is employed in order to improve the impedance characteristic [33]. As for the 0.9/1.5-GHz band antenna, the semi-cylindrical monopole antenna where the diameter

Figure 4.28 Multiband antenna for underground market use.

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is 1/6 wavelength and the height is 1/4 wavelength at the 0.9-GHz band is employed [34]. Figure 4.29 shows V.S.W.R. of semi-cylindrical monopole antennas. Extremely wideband characteristic was obtained in less than V.S.W.R. = 1.5. The antenna configuration used in buildings is shown in Figure 4.30. For the 0.9-GHz band antenna, the disk loaded monopole antenna with the matching post is employed [35]. For the dual band antenna using the 1.5/2.0-GHz band, the two-layered disk loaded and loop antenna is designed [36]. In Figure 4.31 detailed configuration of the 1.5/2.0-GHz band is shown. The two-layered disk loaded and loop antenna, which consists of an upper loop and a bottom rectangle with a center feed probe and matching posts, is mounted on the ground metal plane. The square loop is connected to the bottom patch by the short pin. The bottom rectangular patch was also considered as a low profile disk loaded monopole antenna. Figure 4.32 shows the antenna set in the ceiling of the indoor building in service [37]. This base station antenna uses bidirectivity to obtain a compact and extremely efficient transmission area. It is suitable for indoor installation in locations extending over a relatively long distance from side to side, such as train station complexes and department stores. Figure 4.33 shows configurations of antenna elements that are contained in the radome. The inverted F antenna element is used for vertical polarization and printed dipole antenna element is used for horizontal polarization. These antennas have features such as extremely small coupling between polarization ports, which are capable of broadband and orthogonal polarized-wave and are configured very compactly. Figure 4.33(a) illustrates this configuration and Figure 4.33(b) shows an inverted F antenna element. Figure 4.34 shows radiation patterns in the horizontal plane for horizontal and vertical polarizations. In both polarizations, a bidirectional pattern is achieved.

Figure 4.29 V.S.W.R. of semi-cylindrical monopole antenna.

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Figure 4.30 Configuration of an antenna for in-building use.

4.2.4 Antennas for CDMA Systems 4.2.4.1 Space Diversity and Polarization Diversity [38] In most mobile communications systems, vertical polarization has been used because of the easy implementation of vertical antennas at mobiles and base stations. For the reverse link (the link from mobile to base station), diversity reception at the base station is the prerequisite for enhancing the reception performance to compensate for the inferior mobile transmission capability. Among the various types of diversity schemes, space diversity has been traditionally used. Space Diversity [38] Space diversity needs at least a pair of independent receiving antennas for each sector at a base station for receiving signals via different paths from a mobile. The antennas have to be separated from each other by at least 10 wavelengths (about 3m for the 800-MHz system) for securing low correlation between the received signals to obtain sufficient diversity gain. For a base station of the 3-sector configuration, at least six receiving

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Figure 4.31 Configuration of the two-layered antenna.

Figure 4.32 Antennas set on the ceiling of a building.

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Figure 4.33 (a, b) Configuration of the indoor antenna.

Figure 4.34 (a, b) Radiation patterns of the indoor antenna.

antennas and three transmitting antennas are required. However, it is getting very difficult to acquire cell sites in large cities like Tokyo to install such a number of antennas on top of buildings. Shared use of an antenna element for reception and transmission is a mandatory necessity for reducing the number of antennas. The approach to accommodate 2-sector elements inside a cylindrical radome is being applied to most of the base station antennas in order to reduce the apparent number of antennas for multisector cell sites. An example of this scheme is illustrated in Figure 4.35(a), which is a top view of a 3-sector antenna system consisting of three poles. In each of the poles, two radiating elements are installed.

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Figure 4.35 (a, b) Space and polarization diversity antenna configurations. ( 1997 IEICE.)

Pole 1 contains two vertical arrays, A1,1 and A3,2 , facing to azimuth angles of 0° for Sector 1, and 240° for Sector 3, respectively. The array A1,1 is to receive the signal Rx1,1 to be combined with Rx1,2 received by array A1,2 in Pole 2 for diversity of Sector 1. A1,1 is commonly used for transmitting the signal Tx1 for Sector 1. Tx2 is the signal transmitted by array A2,1 which is shared for receiving the signal Rx2,1 from 120° direction. Rx2,2 is received by array A2,2 in Pole 3 and combined with Rx2,1 for diversity of Sector 2. Transmission and diversity reception of Sector 3 will be made in the same manner. This configuration needs three antenna poles separated by 10 wavelengths each for 3-sector space diversity system. Polarization Diversity [39] Polarization diversity, on the other hand, does not require two spatially separated antennas. Multiple dipole elements with orthogonal polarization can be alternately mounted on a piece of dielectric substrate in a vertical radome. Elements for the orthogonal polarization may be of vertical/horizontal or of +45°/−45° cross dipoles depending on the particular design. Polarization diversity is a well-known diversity technique but has not traditionally been used in cellular phone systems. This is because it was not necessarily effective in the vertical polarization environment where mobiles with vertical trunk-lid antennas on board cars dominated. The number of cellular phone subscribers exceeded 93 million in September 2006 in Japan. Now, more than 99.9% of phones are of hand held type. Users hold their phones in a tilted position like 60 degrees from zenith when they engage in conversation. Moreover,

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antennas in mailing and browsing mode are declined to the point of being almost horizontal. Under such circumstances, it is clear that a strong horizontal component is dominant and uplink polarization diversity should be effective. Figure 4.35(b) shows a top view of the newly developed polarization diversity antenna with a compact structure, which should be compared with Figure 4.35(a) of space diversity. This is an antenna consisting of a single piece of vertical pole containing 3-sector transmitting and receiving arrays with diversity capability inside. Three vertical arrays A1, A2, and A3, are bound together in a shape of triangular pillar and are accommodated in a cylindrical radome. Array A1, for example, contains a set of vertical and horizontal dipoles stacked alternately on a dielectric substrate to receive vertically polarized signal Rx-V1 and horizontally polarized signal Rx-H1 for Sector 1. Either vertical or horizontal dipole elements of array A1, or both, can be shared for transmission of the signal Tx1. The other two arrays, A2 and A3, have identical configuration to A1.

4.2.4.2 Development of New Polarization Diversity Antenna [40] Based on the positive test results described in the previous section, new polarization diversity antennas were developed which were aimed at achieving superior reception performance in the hand-held mobile environment and, at the same time, reducing their size and weight to cope with the stringent environmental requirement. Now, this type of polarization diversity antenna is being used by all CDMA base stations of KDDI Corporation in Japan [41, 42]. This has resulted in easy cell site selection and a reduction of construction costs because of the antennas’ simple structure and superior performance. Figure 4.36 shows an example of an actual installation of a 16-dBi gain antenna in Tokyo. Figure 4.37 shows a simplified inner structure of this type of antenna. Arrays for three sectors are accommodated in the shape of triangular tube in a cylindrical radome of 23-cm diameter. In this example, three subarrays are stacked for achieving high gain. Down tilting is performed by changing the phase angles of top and bottom subarrays in opposite sense relative to the central subarray. The tilting is given independently to each of transmission/reception and vertical/horizontal polarization combination. Each subarray consists of three vertically polarized dipoles and horizontally polarized dipoles stacked alternately [43]. Table 4.2 lists the specifications of the newly developed antennas. Care has to be taken for reducing the coupling between dipoles of different polarization elements as well as that between transmitting and receiving ports. Also important is the intermodulation product that would be generated when multiple carriers are fed to the antenna. One of the potential features of this antenna is the capability of transmission diversity using different polarizations. This shall be readily applied to the 3G CDMA mobile communications system.

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Figure 4.36 Polarization diversity antenna. (Courtesy of KDDI Corp.)

4.2.4.3 Dual Band Polarization Diversity Antenna KDDI Corporation started CDMA2000 1x and 1xEV-DO services in the 2-GHz band in April 2003, in addition to CDMA2000 1x and 1xEV-DO services in the 800-MHz band. These services are being extended from the Tokyo area to all other regions of Japan. In order to reduce the construction cost, dual band polarization diversity antennas of 800-MHz and 2-GHz bands were developed and installed at the base stations, besides the antennas dedicated to the 2-GHz band. The old 800-MHz band polarization diversity antennas were replaced by those of dual band polarization diversity. In order to provide similar service areas to those of existing coverage in the 800-MHz band, antennas of various beam widths were developed (see Table 4.3). For the 2-GHz band, however, only antennas of 80° 3-dB beam width were provided, since only a 3-sector system would be used in this band [44]. Figure 4.38 shows an example of the simplified inner structure of this type of antenna. In the vertical array, the 2-GHz band elements are inserted between the 800-MHz band elements. The antenna elements existing for 800 MHz were resized for the use in 2 GHz. Polarization diversity is provided by either V-H pol. pair or +45°/−45° pol. pair

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Figure 4.37 Inner structure of polarization diversity antenna.

elements (Figure 4.39). Antennas of different beam width were realized by adjusting the width of ground plane in the subarray. The external appearance of the modified dual band antennas is almost the same as the old ones. Figure 4.40 shows the replacement of an existing 800-MHz band polarization diversity antenna with a dual band polarization diversity antenna. It does not look so different before and after the replacement. 4.3 ADAPTIVE ANTENNA SYSTEMS 4.3.1 Personal Handy Phone System [45] Adaptive array antennas increase antenna gain in the direction of a desired signal and suppress interference in the undesired directions. Therefore, use of such antennas would

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Table 4.2 Specifications of Polarization Diversity Antennas Electrical Specifications Type Antenna element Frequency range Gain VSWR −3-dB beam width (vertical plane) −3-dB beam width (horizontal plane) Cross polarization ratio Isolation between V-H ports Isolation between sector beams

AN-951-1 Array of two printed dipoles 818–958 MHz 16 dBi Less than 1.5 Approx. 18° Approx. 70° Greater than 30 dB Greater than 40 dB Greater than 40 dB

Mechanical Specifications Type Radome height Radome diameter Rated wind velocity Connector Weight (with electrical tilt box)

AN-951-1 Approx. 3,200 mm 239 mm 60 m/s (Survival) BFX-20D Approx. 124 kg

Table 4.3 A Set of Horizontal Plane 3-dB Beam Width of the Newly Developed Dual Band Polarization Diversity Antennas

Type Type Type Type

1 2 3 4

800-MHz Band

2-GHz Band

50° 60° 70° 90°

80° (for 3-sector)

(for (for (for (for

6-sector) 6-sector) 3-sector) 3-sector)

save transmission power and increase the system capacity. Willcom introduced this antenna into the Personal Handy Phone System (PHS) in 1998. Four antennas are used for transmission and reception. For the output signal of each antenna, suitable weight produced in the CPU module is given to achieve the best bit error rate (BER) value by means of CMA algorithm. Maximum antenna gain is obtained in the direction of the desired signal, and pattern nulls are created in the undesired directions. Since the TDD system uses the same frequency band for the uplink and downlink, propagation conditions of both links are expected to be the same. Therefore, the determined weight for the uplink to suppress the interference can also be effective for the downlink to increase the gain.

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Figure 4.38 Inner structure of dual band polarization diversity antenna.

The base station antenna configuration is shown in Figure 4.41. Each antenna has gain of 10 dBi. Antenna spacing is 5 wavelengths. The radio unit is installed at the foot of the antenna. Antenna radiation patterns with and without interference are shown in Figure 4.42 [46]. A deep pattern null is generated in the direction of interference, ensuring adaptive performance of the antenna system. A field test was conducted for the case of a conventional space diversity antenna having a transmission power of 500 mW and for an adaptive antenna with a total transmission power of 125 mW [47]. Since the same communication quality in the same cell size should be obtained for both cases, a 6-dB increase in antenna gain was confirmed. The average of the received power by the adaptive antenna was 3 dB higher than that of the conventional diversity antenna. 4.3.2 W-OAM Willcom Optimized Adaptive Modulation (W-OAM) is the newly developed system to enhance the PHS capability. This system uses modulation of ␲ /2 shift BPSK, ␲ /4 shift

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Figure 4.39 Dual band polarization diversity element. ( 2004 IEICE.)

QPSK, D8PSK, 16QAM, 32QAM, 64QAM, and 256QAM in addition to ␲ /4 QPSK used in the conventional PHS system. For the case of 64QAM, the obtained data rate is approximately 768 Kbps. Figure 4.43 shows the adaptive array antenna of a W-OAM system base station. This system is composed of an array of eight antennas, to make up one common channel and seven traffic channels. W-OAM is a two-multiaccess space division multiple access (SDMA) system using adaptive array antenna technology (Figure 4.43). 4.3.3 i-Burst System The i-Burst base station is equipped with a smart antenna for both transmission and reception by using 12 antennas, and it realizes a triple-fold multiple access of SDMA. The feeder loss of the base station is reduced by the architecture of separating the power amplifier (PA)/low noise amplifier (LNA) and base unit (controller). The container of the

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Figure 4.40 (a, b) Appearances of dual band polarization diversity antenna. (Courtesy of KDDI Corp.)

base station equipment is waterproof and can be conveniently installed at any possible site. A base station unit is shown in Figure 4.44. The algorithm adopted in an i-Burst adaptive array antenna system is minimum mean square error (MMSE). It is possible to determine the array weight at the transmission timing by using the weight that was generated at the reception timing. As a result, high SINR condition is provided for mobile stations, considering the propagation characteristics, and communication quality is greatly improved [48, 49]. The data rate was measured in Sydney, Australia, where commercial service had been started and interference was low. The results in Table 4.4 show that the effective data rate with 24 mobile stations was 29.569 Mbps in a 5-MHz bandwidth and the spectrum efficiency was 5.91 bit/sec/Hz/sector (= 29.569 Mbps/5 MHz). This value is close to that of the ideal condition with no loading. On the other hand, the simulation result of 19 base stations with loading is 3.40 bit/sec/Hz/sector for the forward link and 2.24 bit/sec/Hz/ sector for the reverse link [50]. The measurement condition of the drive test is shown in Table 4.5. We focused on the SDMA mode where the degradation of the downlink performance with fast-moving MS is significant. The cumulative distribution function (CDF) of SINR in the urban area is shown in Figure 4.45. The gain at 0.1% of CDF is seen to be approximately 5 dB [51].

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Figure 4.41 PHS adaptive base station antenna configuration. (Courtesy of Kyocera Corp.)

4.3.4 Experimental System of Adaptive Array for WCDMA The WCDMA system has the features of coherent adaptive antenna array diversity (CAAAD) and adaptive antenna array (AAA). In the CAAAD, the antenna weight is controlled to minimize the MSE using Rake combining, by using the pilot symbol after inverse spreading and the data symbol for information decision. For the transmitting antenna, the AAA uses the weight that is generated based on the receiving antenna weight in the CAAAD receiver. A series of experiments on the adaptive array antenna system for WCDMA were made in Ichikawa-City, Japan, by NTT DoCoMo. Six antenna elements of 120° beam width were aligned linearly at a base station 50m high. The interelement spacing is halfwavelength of the received signal (Figure 4.46). Figure 4.47 shows the receiving patterns at the base station corresponding to the maximum received power of CAAAD, where there were four mobiles on the uplink. For one mobile station traveling along the test course, the average of received Eb/No was 25 dB at one time in the base station antenna and the average of received SIR was −15 dB at another time. As shown in Figure 4.47, the main beam direction varied, from

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Figure 4.42 Antenna radiation pattern of interference suppression.

the peak of the solid line to that of the dotted line in this case, as the mobile station moved. Though the interfering mobile station was fixed at +40° from the base station, the null location varied according to the movement of the mobile station. It is thought that using the receiving antenna weight to obtain maximum SINR, the antenna pattern changed so that the gain in the desired signal was raised, but the location of null shifted from the direction of arrival of the interference signal [52, 53]. 4.3.5 Experimental System of Adaptive Array for CDMA2000 1xEV-DO In order to apply adaptive antennas to SDMA packet cellular system using CDMA2000 1xEV-DO, KDDI evaluated two kinds of antenna systems. One is a circular array antenna that could uniformly control the radiation patterns in all directions for a tower top antenna. The other is a planar array antenna installed on the building wall. Figure 4.48 shows (a) the circular and (b) the planar array antennas. The capability of the arrays was evaluated for controlling both beams and nulls in the directions of the desired mobile stations and

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Figure 4.43 W-OAM adaptive antenna system. (Courtesy of Kyocera Corp.)

the undesired mobile stations, respectively. Figure 4.49 shows the radiation patterns directing beams to desired mobile stations and nulls to interference. Figure 4.50 shows the simulation results of the throughput in the SDMA systems using the circular and the planar array antennas. Comparing this with the system that uses omnidirectional antenna without SDMA, it is seen that the cell throughput of SDMA with planar array antenna is improved by a factor of three to four for both cases of beam control only and beamand-null control. On the other hand, the improvement of throughput of SDMA with the circular array antenna is four to six times for beam control only, and from four to eight times for beam-and-null control [54, 55]. 4.4 DESIGN AND PRACTICE II (EUROPEAN SYSTEMS) Antennas are an essential part of every wireless communication system. As these systems are standardized, developed, and refined, enhanced antenna solutions will, in many cases, greatly contribute to the success of concepts and products [56]. In a number of systems, improved antenna solutions will be needed due to, for example, increased capacity and coverage requirements, new frequency allocations with possibly fragmented spectrum, cost and size reductions, and new applications.

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Figure 4.44 i-Burst adaptive antenna system. (Courtesy of Kyocera Corp.)

Table 4.4 Data Rate with 24 Mobile Stations in Sydney, Australia

Forward link Reverse link Sum

Average/MS (Max.)

Average/BS (Max.)

942 Kbps (1,061 Kbps) 277 Kbps (346 Kbps) 1,219 Kbps (1,407 Kbps)

22,616 Kbps (24.403 Kbps) 6,953 Kbps (8.304 Kbps) 29,569 Kbps (32.707 Kbps)

Table 4.5 The Measurement Condition of Drive Test BS antenna height MS antenna height Distance Velocity

30m 1m 0–1.8 km 0–50 km/h

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Figure 4.45 CDF of SINR in normal drive test.

4.4.1 Antenna Configurations In mobile communications, advanced antenna design is an area of high importance where different array antenna solutions are being studied. One example is the possible use of multibeam antennas, where substantial C/I improvements have been shown in GSM and TDMA cellular systems [57]. Another example is MIMO or spatial multiplexing antenna systems, which can be used to further increase data rates without requiring any extra bandwidth by transmitting multiple parallel streams to a user [58, 59]. Signals are processed adaptively at either the transmit side or the receive side, or both, to exploit the spatial or polarization dimensions of the radio channel. 4.4.1.1 Sector Antennas Conventional macro base stations use three antenna beam patterns to cover the service area, or cell, around a base station site—the area is said to be divided into sectors. This

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Figure 4.46 Adaptive array antenna based on WCDMA. (Courtesy of NTT DoCoMo Corp.)

will typically require three or six separate sector antennas, depending on if polarization or space diversity is used on uplink from terminal to base station. Sector antennas are typically implemented as vertical linear arrays of radiating elements and provide basic azimuth filtering of interfering signals. For low-capacity applications, omnidirectional antennas may be of interest, particularly to achieve coverage during early roll-out. However, omnidirectional antennas provide no azimuth filtering. Sample cell plans and sectorization resulting from three different base station antenna configurations are presented in Figure 4.51. These are omnidirectional antennas, 120° half-power beam-width antennas in the ‘‘Bell cell plan,’’ and 65° half-power beam-width antennas in the ‘‘Ericsson cell plan.’’ Contour plots show intercell signal-to-interference ratio from high to low, which depends on antenna gain and path loss under an ideal propagation model without fading, when all antennas transmit equal power. For each antenna beam, the corresponding sector is defined as the area where that beam provides the highest field strength. From an interference point of view, the capacity of a sectorized system is approximately proportional to the number of sectors per site. Sectorized systems can be improved by introducing adaptive signal processing—a basic form of which is reception diversity— which can be used to minimize the effects of fading on uplink. To achieve better performance, two or more sufficiently uncorrelated signals are received. Space diversity, in which two sector antennas are separated some 10 to 20 wavelengths apart, and polarization diversity, in which a dual polarized sector antenna is utilized, are two common physical realizations to provide uncorrelated signals. The signals are then combined typically at baseband to optimize the desired received signal level.

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Figure 4.47 Radiation patterns in the case of receiving r.

Current cellular systems support transmit diversity, which is a technique to improve also the downlink performance by transmitting signals via two or more sector antennas. For example, by adjusting the relative phase of identical signals, transmitted via different base station antennas, the total signal received by the user terminal is optimized. This can be seen as a generalized phase-steered interferometer in which the relative phase value is continuously modified to counter the effects of multipath induced fading. 4.4.1.2 Two-Dimensional Array Antennas Improved azimuth filtering can be achieved by using two-dimensional array antennas. Planar array antennas are in use, but there is also an increasing interest in nonplanar, conformal, antennas—for example, circular-cylindrical arrays. Array antennas can be divided into two major categories: fixed-beam antenna and steered-beam antenna configurations.

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Figure 4.48 Smart antenna.

Figure 4.49 Radiation patterns in the case of receiving.

Fixed-beam array antennas provide a number of fixed, nonoverlapping, orthogonal beams using a fixed beam-forming network in the antenna, and each port of the antenna provides signals associated with a beam. A fixed-beam antenna can be used as part of an adaptive antenna system, in which uplink signals from the beams are optimally combined

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Figure 4.50 Maximum cell throughput. ( 2006 IEICE.)

Figure 4.51 Cell plans and sectorization (thick, solid lines) resulting from three different base station antenna configurations: (a) omnidirectional antennas; (b) 120° half-power beam-width antennas pointing toward the symmetry point between three base stations (‘‘Bell cell plan’’); and (c) 65° half-power beam-width antennas, each pointing toward a neighboring base station along a sector border of that base station (‘‘Ericsson cell plan’’). The scales from high (around the center) to low (around the end of the circles) in the different plots are not identical.

and the envelope of the gain pattern of all fixed beams define a cell. Downlink beam selection can be based on uplink signal properties or feedback information from the terminal. A fixed-beam antenna can also be used in a conventional sectorized system, by assigning one cell to each beam. Fixed-beam antennas are referred to as noncoherent

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since each beam is generated at RF in the antenna unit and the relative phase values of feeder cables do not affect the shape of the beams. A sample contour plot of the ratio between intracell and intercell signal levels, when all base stations use four narrow beam antennas within each 120° sector, is shown in Figure 4.52. The contour plot shows intercell signal-to-interference ratio from high to low, which depends on antenna gain and path loss, assuming an ideal propagation model without fading, when all antennas transmit equal power. In steered-beam array antennas, a number of subunits (elements, columns, or subarrays) are individually fed. Beams are formed by weighting the fed signals to each subunit with respect to phase, amplitude, and time. This can be used to optimize the antenna system performance by maximizing the gain or reducing the interference. Each signal path has to be characterized with respect to phase and amplitude since the system has to apply proper weights in order to achieve the desired beam shape. Steered-beam antenna systems are therefore referred to as being coherent. A physical fixed-beam antenna can be used in a coherent system if desired; such a system combines signals coherently from the individual fixed beams to provide beam steering. 4.4.1.3 Antenna Polarization Radiated waves have polarization directions given by the properties of the antenna elements and objects in their immediate vicinity. Both single and dual polarized base station antennas are common, most of them generating linearly polarized fields. Dual polarized antennas can be designed to radiate a number of orthogonal polarizations. The most common polarization combinations are vertical and horizontal, and ±45° (cross-polarized). The latter is often favored. It provides the base station with two uplink signals of equal amplitude on average. Antennas with other polarization combinations, such as right- and left-hand circular, are also feasible in base station applications. The ability to adapt to the strongest polarization would be the optimal choice [60].

Figure 4.52 Cell plan and sectorization resulting from four-beam planar array antennas.

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4.4.1.4 Calibration Calibration refers to the process by which array antenna systems correct for variations in behavior of the different signal paths between the array antenna and baseband processing equipment during operation. A calibration process involves three steps: test signal injection, test signal measurements, and calculation and application of correction terms. Calibration methods differ mainly with respect to how and where test signals are injected and what type of test signals is used. The calibration loop (i.e., the part of the signal path through which the test signal is transmitted) includes those components which can have variations in performance such as amplitude, phase, and time, large enough to critically affect antenna system operation and performance. Calibration will include all analog parts except for the antenna and antenna-installed feed networks. Digital parts may be included, typically because test signal injection is facilitated if done digitally. A block diagram representation of an array antenna system with calibration is shown in Figure 4.53. An extra feeder cable is required for transmission of test signals to/from the array antenna. The same feeder is used for calibrating both transmit (TX) and receive (RX) signal paths. In the calibration coupling unit (CCU), which is located directly at the antenna ports, all signals paths could be connected to the extra feeder cable using a splitter/ combiner. No parts of the signal paths beyond (in the direction of the antenna aperture) the point where the CCU is located can be calibrated. Calibration is usually not necessary in systems without explicit beam-forming. Fixed multibeam array antenna systems (noncoherently implemented) must have similar amplitude differences between uplink and downlink signal paths for each beam, since downlink beam selection is based on processing of uplink data, for example, on received

Figure 4.53 Adaptive array antenna system including calibration parts.

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uplink power. When the amplitude variations in the components before the duplex filter (looking towards the antenna) are negligible, calibration is not necessary. Steered-beam or coherent array antenna systems have to be calibrated to have similar amplitude, phase, and time-delay performance for each signal path. The calibration is done separately for transmit and receive chains. Calibration of the receive chain is necessary to allow for correct calculation of uplink weight vectors when direction information is desired. Downlink beam-forming requires a set of coherent signal paths, which is why transmit chain calibration is required. Once all signal paths are calibrated, correct weight vector calculation can be performed and beams can be generated in arbitrary directions. The required calibration accuracy depends on the system implementation. The use of hardware beam-formers at RF, such as Butler matrices, may reduce the accuracy requirements for a steered-beam system.

4.4.1.5 Characterization Pattern characterization refers to predeployment procedures performed on an antenna measurement range. It is typically performed only once, since it describes effects related to antenna features which are essentially time-invariant. Based on the characterization, fixed compensations can be introduced in the system to handle effects due to mutual coupling and finite aperture size. Antennas have to be properly characterized, since calibration cannot compensate for unknown built-in deficiencies affecting the radiation pattern. Antenna systems without explicit beam-forming require little or no characterization beyond conventional radiation pattern measurements, except for feeder cable characterization. Since, by design, there are no degrees of freedom available during operation, which depend on well-defined phase differences between different signals path, there are no phase effects that need to be compensated for using calibration. This is true both for sectorized systems and systems using transmit diversity. In some systems, feeder cable time delays should be known to an accuracy of one-tenth of the chip period, or better. In conventional fixed-beam array antenna systems, a separate feeder cable is associated with each beam. Therefore, no means for compensating pattern shape deficiencies exist, which means that the antenna pattern performance is decided during the antenna design phase. Fixed multibeam array antennas may still need to be characterized with respect to gain and pattern shape for each beam, depending on manufacturing tolerances. Steered-beam array antenna systems must have known amplitude, phase, and timedelay performance for each signal path. More than one signal path is typically used to generate each specific beam. A steered-beam antenna may need to be characterized with respect to its scattering parameters including mutual coupling, as well as radiation pattern performance, on a port-by-port basis.

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4.4.2 Antenna Solutions 4.4.2.1 Cell Systems Base station antenna installations for mobile communication systems are very dependent on site type and other requirements. One basic division that can be made is in antennas for macro cells, micro cells, and distributed antennas. Traditionally, the following antennas have been used for these different situations. Macro Cells In early system deployment, an installation may consist of omnidirectional antennas, mounted in pairs separated some meters apart to obtain space diversity in uplink. When capacity demand increases, 3-sector sites may be introduced, having antennas installed with 120° angular separation. The azimuth half-power beam width of these antennas is either 60° or 90° depending on the cell planning principle that has been chosen. Antenna lengths are from a few up to almost 20 wavelengths. A commonly installed antenna is about 8 wavelengths long and has a gain of 18 dBi. In recent years, dual polarized antennas have become more and more common with the introduction of compact polarization diversity installations (see Figure 4.54). With new frequency bands allocated for mobile communication systems, a more recent demand for colocation and multiband antennas has shown up, in order to better use existing sites. In areas where high capacity is needed, a dense network of sites with smaller coverage area is used. The visual, aesthetic aspects of antennas in these locations are coming more into focus. There are also regions where deployments of any type of standard antenna masts are seen as a major aesthetic problem. In such areas, specially designed antenna masts have been developed which resemble other objects, for example trees, cactuses, and flag poles [61]. Micro Cells For antenna installations below rooftop level, or indoors, a separate class of antennas has been developed. For this type of cell, the visual aspect of antenna design is highly important. As the propagation circumstances also are very different from a macro cell, the typical antennas have pairs of small omni-directional monopoles, separated a few wavelengths apart for diversity reception. An alternative solution has been sector antennas with relative wide azimuth beams (half-power beam width wider than 120°), which have been used for wall installations. Space diversity is often used, but polarization diversity can be a preferred solution in

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Figure 4.54 Dual polarized 3-sector base station antenna installation.

these environments, if antenna isolation requirements can be fulfilled. Depending on the scenario, antennas are installed within the enclosure of the micro base station (Figure 4.55) or as separate external antennas.

Distributed Cells At some locations the most efficient antenna solution is a cell with distributed antennas. This category contains installations where a larger number of radiators are connected to the same base station. Examples of typical installations are inside buildings and tunnels with several local antennas connected through an RF distribution network, or a leaking cable solution [62].

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Figure 4.55 Space diversity wide-beam sector antenna installation for micro cell applications.

4.4.2.2 Array Antennas Driven by an expected need for increased capacity and coverage in mobile communication systems, two-dimensional antenna arrays have been brought up as a possibility. These antenna arrays have a horizontal extension that enables narrow antenna beams to be created in the azimuth plane. The interference situation is reduced (improved C/I) with this spatial filtering technique (Figure 4.56) and the capacity is thereby increased in the network. Array antennas also provide increased coverage as the two-dimensional antennas typically have larger area and thereby higher antenna gain than traditional wide-beam sector antennas. Antenna arrays that are connected to a base station utilizing the capability to form narrow multibeams or steered beams are often referred to as smart or adaptive antennas [63]. Adaptive Antennas An example of an adaptive antenna is a fixed multibeam array antenna with RF beamforming, as shown in Figure 4.57. The antenna is a dual polarized five-column array,

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Figure 4.56 Base station array antenna for interference suppression.

Figure 4.57 Fixed dual polarized multibeam adaptive antenna principle.

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where four columns are used for creation of narrow interleaved beams in azimuth while the fifth column is used for broadcast transmission. The radiating elements are dual polarized microstrip patches, separated 0.5 wavelength apart between columns and 0.9 wavelengths within columns. A triangular grid provides greater element spacing compared to a square lattice before the onset of grating lobes in a planar phased array [64]. These adaptive antennas use a fixed multibeam architecture, where the narrow beams are created in the antenna by using RF beam-forming networks of the Butler matrix type. The used polarization directions in this case are slanted ±45° linear orientation. In each polarization, this gives a set of orthogonal beams covering a 120° sector (Figure 4.58). In uplink it is possible to use the total information from all interleaved beams in order to maximize reception performance and to estimate direction-of-arrivals (DOA) for communicating terminals. Based on the DOA estimate, a single beam can be selected for downlink transmission. By selecting a single transmission path, it is possible to avoid coherency requirements in the feeder cables between the array antenna and the base station as beams are formed at RF in the antenna. In many communication systems, base stations have to occasionally transmit a control or common channel over the entire sector. This requirement can be accomplished with a separate sector antenna function integrated as part of the adaptive array antenna system. An effective solution uses an additional column of radiating elements next to the array antenna columns. The sector antenna beam pattern and the array antenna multibeam envelope need to track each other closely. Due to the polarization diversity technique, and the integration of sector coverage and narrow beam apertures within the same enclosure, only a single antenna unit is needed in every sector (Figure 4.59). This gives a less aesthetic impact and a simpler installation procedure, while still giving large improvements in system performance. Cylindrical Antennas To date, mainly planar antenna arrays have been used for advanced antenna applications in mobile communication systems. Nevertheless, it is an advantage in many systems to

Figure 4.58 Azimuth radiation patterns of a dual polarized adaptive array antenna with eight interleaved fixed beams and a sector beam.

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Figure 4.59 Fixed multibeam adaptive array antenna with integrated sector beam.

use an antenna that combines high gain and low interference levels with a very large angular coverage. This may be achieved with a cylindrical antenna array (Figure 4.60) that is able to direct a narrow beam over a large angular sector. Over this large angular sector, no pattern degradation occurs that otherwise would result in lower gain and broader beams leading to limitations in capacity and area coverage. In other cases, it may also be desirable to make the radiating elements of an array antenna cylindrical. The reasons may be wind-load related or aesthetic, for example when mounting base station antennas on a pole or cylindrical mast. Before the introduction of cylindrical antenna arrays, the electromagnetic properties of such arrays (e.g., mutual coupling) have to be fully understood to be able to facilitate a high-quality design. Furthermore, the beam-forming properties are also different from a planar antenna array. With proper feeding, the cylindrical array can give simultaneous narrow and omnidirectional beams [65]. 4.4.2.3 Antenna Integrated Amplifiers Antenna integrated amplifiers are here referred to as antennas with the power amplifiers very close to the radiating aperture of the antenna. Such antennas can be configured in different functional complexities, from single element antennas to adaptive arrays.

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Figure 4.60 Cylindrical array antenna.

Active Sector Antennas Traditionally, mobile communication base stations are attached to mast-mounted passive antennas. It has been necessary to use amplifiers with very high output power together with low-loss feeder cables to obtain sufficient effective isotropic radiated power (EIRP) levels from these antenna installations. Although high power amplifiers may have relatively high efficiency, the overall power efficiency of a traditional base station installation is reduced due to feeder losses between the power amplifiers and the radiating elements. As a consequence of the reduced efficiency, heat is generated in the base station cabinets, resulting in unnecessary cooling needs. Air conditioners have to be installed, further reducing the overall efficiency of the base station installation. As feeding losses are system critical, low-loss feeder cables are preferred. The introduction of antenna integrated amplifiers (Figure 4.61) with power amplifiers located close to the radiating elements in the antenna array has the potential to give much better overall power efficiency. This also leads to a much smaller total volume of base station equipment and makes the site well suited for areas where large installations are a concern. The antenna and the base station cabinet may be installed separately and connected with low-loss thin coaxial cables. Antenna integrated amplifiers also allow for graceful performance degradation in the event of an amplifier fault. Only a small reduction of the effective radiated power level results from the failure of one of the power amplifiers. Active Adaptive Antennas Adaptive antenna systems with passive antennas have proven to be very efficient in increasing capacity in mobile networks, but they still deal with the same drawbacks as

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Figure 4.61 Conventional antenna (top) and antenna integrated amplifiers (bottom) connected to a radio base station (RBS).

traditional base station arrangements regarding efficiency and power consumption. High output power amplifiers, air conditioners, and low-loss feeder cables are required to get a reasonable level of equivalent radiated power. The introduction of power amplifiers located close to the radiating elements would greatly improve the overall power efficiency also for adaptive antennas. There are different ways to configure such an active adaptive antenna array. The configurations described here have the beam-forming function located in the antenna unit, with the argument that correct beam positioning requires phase coherency to the radiating elements. It is an advantage to perform RF beam-forming within the antenna unit instead of having it done in the base station cabinet at baseband to avoid the need for feeder cable calibration. Two different configurations of active adaptive antennas are with power amplifiers either between the feeder cable and the RF beamformer, or between the RF beamformer and the radiating elements. If a uniform tapered beamformer is used, both these configurations have the important characteristic that every signal fed into any of the beam ports are divided equally to all power amplifiers. An even load over all power amplifiers minimizes peak power and intermodulation requirements on the amplifiers. As the purpose of using active adaptive antennas is to increase capacity, it is essential that the system can handle several simultaneous carriers. The described power amplifiers must therefore be so called multicarrier power amplifiers (MCPA). These MCPAs are critical components in order to achieve efficient active adaptive antenna systems. One of the potential advantages of active adaptive antennas is intermodulation products (IM) from the different amplifiers that are not correlated. When that is the case,

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the IM signal is not directed into a narrow high-gain beam. The suppression of effective isotropic radiated power for the IM signal with respect to EIRP for the carrier signal is then enhanced, and the requirements on the individual MCPA can be relaxed. Distributed Active Antennas With the introduction of antenna integrated amplifiers, feeder losses do not put as strict limitations on system performance. It is therefore possible to distribute antenna integrated amplifiers at large distances from the base station, even splitting energy in power dividers along the way. A cell from a base station can be built up using a number of distributed antenna integrated amplifiers that are installed close to the expected terminal positions, for example in a building. The required output power is in such case kept low, which makes it possible to design very small units with low power consumption. The signal transmission between the base station and the distributed antenna integrated amplifiers can be of different kinds (e.g., RF, digital, or optical). 4.4.3 Antenna Units The most common base station antenna in use today is the sector antenna with fixed azimuth beam-pointing direction, which consists of a single vertical linear array of radiating antenna elements fed using a corporate feed. The sector antenna may be dual polarized and may also have built-in fixed or variable down-tilt functionality. A wide variety of technologies exist for realizing antenna units, although only a few of these fulfill all the requirements of a commercial base station product. Advanced and adaptive array antenna systems may use more than one column of antenna elements. Still, antenna units for advanced antenna systems will share a number of properties with a sector antenna. This means that technologies used for conventional sector antennas often can be used also for advanced antenna solutions. The antenna unit is divided into a number of subunits (see Figure 4.62). These subunits are discussed individually; in reality, choosing a particular technology for one subunit may limit the number of suitable implementations of other subunits. The subunits are described, starting at the antenna aperture and moving towards the base station. 4.4.3.1 Radiating Elements Two types of array antenna radiating elements are in common use: microstrip patches and dipoles, as sketched in Figure 4.63. Many products employ microstrip patches, and for a number of reasons: they are easy to analyze with commercial software, they can be manufactured with high precision at low cost, and they can, if aperture-fed, provide large bandwidths while being low profile [66]. Dipoles may be more suitable for multiband

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Figure 4.62 Array antenna subunits shown for one polarization. From left to right: calibration coupling unit (CCU), azimuth beam-forming network (BFN; not necessary present in a phase-coherent array antenna system), vertical feed and beam-tilt network (FN), and array aperture with radiating elements. The CCU position can be closer to the aperture and may also be combined with the BFN. Sector antennas typically include the same subunits, expect for those associated with azimuth beam-forming (i.e., CCU and BFN).

Figure 4.63 Top view sketch of the two most common dual polarized antenna elements used in base station antennas: (a) aperture-fed square microstrip patch; and (b) crossed dipoles.

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array antennas, particularly from a building practice point of view. Multiple dipole feeds, for multiple frequency bands, are typically easier to integrate than multiple feed slots for aperture-coupled patches. The radiating antenna elements, and the backing ground plane they are mounted on, affect a number of electromagnetic performance parameters, such as element radiation pattern shape, matching bandwidth, polarization orthogonality, and ohmic losses. The first one of these, the radiation pattern of the radiating element, is particularly important since the capacity of cellular systems using sector antennas can be optimized using the beam width [67]. The same type of optimization can be performed for array antenna systems, although the element pattern beam width may be less important for array antennas since the array configuration itself provides most of the beam shaping. 4.4.3.2 Aperture Shapes Most existing adaptive antenna systems use planar array antennas as radiators [see Figure 4.64(b)]. All arrays are shown with rotated micro-strip patch elements, which are suitable for ±45° dual polarized radiation. The planar array is fairly easy to analyze and is commonly modeled in terms of a product between the element factor and the array factor, which depends on the relative positions of the individual antenna columns. A conventional planar array uses radiating elements placed in a regular grid to achieve the desired radiation performance. The array antenna can then be thought of as an array of identical sector antennas, connected together in the horizontal dimension via a beam-forming network, either explicitly at RF or implicitly at baseband. For some applications, nonplanar array apertures may be more suitable. One example of a nonplanar array is the circular-cylindrical array [Figure 4.64(c)], which can produce narrow beams in arbitrary directions around the cylinder axis. Combined with a proper beam-forming network, the circular-cylindrical antenna can produce simultaneous narrow

Figure 4.64 Sector and two-dimensional array configurations: (a) sector antenna; (b) planar array; (c) circular-cylindrical array; and (d) faceted array.

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and omnidirectional beams, with the former used for dedicated channels and the latter for common channels [65]. Another implementation of a cylindrical array antenna is the faceted aperture [Figure 4.64(d)]. This can be used either to generate individual sector beams from a common antenna structure, using one facet for each sector pattern and desired coverage region, or to produce omnidirectional coverage [68] with facets along the entire circumference of a cylinder. From a practical point of view, the performance of the circular-cylindrical and faceted array apertures is quite similar. The array configurations outlined in Figure 4.64 are shown with micro-strip patch radiators, but any suitable element can be used, for example, dipoles. In the case of nonplanar arrays, the radiation patterns do not necessarily cover all azimuth angles, but can also be designed to cover only fractions of 360°. 4.4.3.3 Feed Networks The RF feed network is a passive transmission-line distribution/combining network that connects two or more antenna radiating elements in a column to a common feed point. In aperture-fed micro-strip antennas, the feed network is commonly a micro-strip or stripline network running on the rear side of the ground plane of the radiating elements. Other feed network solutions are also in common usage, for example, coaxial cables. The vertical array complex excitation of all elements in an array antenna controls the shape of the elevation radiation pattern. A feed network for each polarization should provide the desired array distribution over the operating bandwidth. One solution which can provide good bandwidth is the corporate feed—that is, a network of parallel feed lines where the phase lengths, from the feed point to each radiating element, are identical if a constant-phase array distribution is desired. See Figure 4.65(a). The basic vertical array distribution is uniform with respect to both amplitude and phase. This generates a sector-beam with its elevation peak pointing in the plane orthogonal to the vertical array axis. Down-tilting the main beam is often desirable in order to achieve suppression of interference from neighboring sites. Beam-tilt can be achieved by mechanically tilting the entire antenna unit shown in Figure 4.65(b). By applying a progressive phase-shift or true time delays over the radiating elements in the vertical direction, an electrical beamtilt can be generated [Figure 4.65(c)]. A phase-shift can also improve the matching as well as the isolation between orthogonal polarizations. Mechanical and electrical beamtilts can also be combined to improve the front-to-back ratio, which is another source of interference. In antennas with adaptive down-tilt, each radiating element or group of elements in elevation is fed individually from a phase-steering unit. A more general vertical aperture distribution can also be used, with nonuniform amplitude and phase values. This can be used for shaping the elevation pattern, for example to achieve null-filling or an asymmetric main beam shape.

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Figure 4.65 Schematic layout of corporate feed networks connecting radiating elements in a single vertical antenna column: (a) identical path lengths give uniform phase; (b) mechanical beam-tilt; and (c) electrical beam-tilt using time delay.

4.4.3.4 Beam-Forming Networks A beam-forming network generates desired azimuth beams from a two-dimensional array antenna by controlling the horizontal phase and amplitude distribution over the array columns. In a communication system with many users connected simultaneously, a beamforming network has to support coverage of the entire served cell. This means that, in a noncoherent array antenna system, the number of ports of a fixed beam-forming network has to be equal to the number of beams that is generated. Two common RF beam-forming networks are the Butler and Blass matrices, which are passive networks [69]. The Butler matrix depends on line lengths and hybrid couplers to generate a uniform amplitude distribution and a progressive phase-shift from column to column, the magnitude of the phase-shift being different for each input port (Figure 4.66) [70]. An example of a beam-forming network for a circular-cylindrical array is shown in Figure 4.67. This network provides simultaneous sector and omni-coverage by combining two Butler matrices back to back [65]. In a coherent array antenna system, beam-forming is performed at baseband, although a fixed beam-forming network may also be used to achieve potentially better performance. The horizontal aperture distribution can be a function of the desired radiation pattern on

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Figure 4.66 A Butler matrix RF beam-forming network for feeding an eight-element array antenna. Boxes denote 90° hybrid couplers, and fixed phase-shifter values are in degrees.

Figure 4.67 A Butler matrix-fed circular-cylindrical array antenna with simultaneous sector and omnidirectional beams.

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a user-by-user basis. During the design and characterization of an array antenna, it may be necessary to establish to what degree the aperture distribution can be controlled; mutual coupling and other effects can limit the potential for radiation pattern shaping. 4.4.3.5 Power Amplifier Configurations Load Balancing Two basic implementations of power amplifier configurations can be distinguished. The first one has individual power amplifiers located in equivalent element space, whereas the second one has power amplifiers located in equivalent beam space. The two implementations have fundamentally different behavior in terms of load balancing and coherency requirements. Power amplifiers in element space, shown in Figure 4.68, will be a pooled resource for several independent signals. The primary effect of this is load balancing, which means that a specific signal is amplified by more than one power amplifier. This results in efficient power usage as well as graceful degradation of performance. For example, if one power amplifier is lost, output power is still available for all independent signals, at the expense of power reduction and loss of orthogonality between signals for all signals. The minimum required number, M, of power amplifiers in a pool is equal to the maximum number, K, of independent signals to be amplified, if independency shall be

Figure 4.68 Power amplifiers in (a) element space and (b) equivalent element space.

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retained after amplification. Pooling in element space requires coherency within the pool. Element space does not necessarily mean that the power amplifiers are directly feeding the antenna radiating elements but that the signals are transformed to a space equivalent to element space before amplification, as shown in Figure 4.68(b). An advantage of implementing the power amplification resource in beam space (Figure 4.69) is that there are no coherency requirements for the different amplifiers. A disadvantage is that the amplifier resource is not pooled, since each unit will amplify independent signals. This means that there is no load balancing and a faulty amplifier may result in a beam becoming unavailable. The power amplifier configurations considered here are characterized by the number of antenna elements, the number of beam ports, and the number of power amplifiers. When the number of antenna elements and beam ports are equal and the beam-former is a Butler matrix, as many orthogonal beams will be generated as there are antenna elements. When the number of elements is larger than the number of used beam ports, there are still orthogonal beams but not all of the beams are used. This may be a desired property, for example, if the area to cover has a different shape compared to the element radiation pattern. When the number of element ports is less than the number of beam ports, the beams will no longer be orthogonal. In addition, the system will suffer from severe power loss in downlink if the power amplification precedes the RF beam-former.

Figure 4.69 Power amplifiers in (a) beam space and (b) equivalent beam space.

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Dedicated and Common Channels In conventional sectorized systems, the dedicated and the common channels are always transmitted over the same area, which defines the cell. This type of system uses the same antenna beam port and power amplifier resources for both channel types. More sophisticated systems adapt the transmission of the dedicated channel to the downlink radio channel of each user. This adaptation, which may be performed by explicit beamforming or TX diversity, is based on using more than one beam port. Load balancing over dedicated channel beams in a fixed-beam system can be achieved by using a power amplifier resource in element space. The common channel can be handled by a separate power amplifier, as illustrated in Figure 4.70(a). In order to have load balancing, not only over beams but also over channels, a common power amplifier resource has to be used. If the dedicated and common channels use separate antenna elements, the power resources cannot be placed directly at the elements, since not all elements carry both dedicated and common signals. Instead, the common resource has to be located in an equivalent element space as show in Figure 4.70(b). A steeredbeam system can never be load balanced for both dedicated and common channels unless the common channel is transmitted with equal power over the array radiating elements. 4.4.4 Antenna Development Trends In order to be prepared for antenna systems architecture and design work needed in coming generations of mobile communication systems, it is important to understand the fundamental principle these systems are expected to have. There are several different options for future generation wireless systems to evolve. One common trend that is

Figure 4.70 Fixed-beam configurations with (a) separate and (b) common power amplifier (PA) resources for dedicated channel (DCH) and common channel (CCH).

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important for this evolution is the growing amount of data traffic in the mobile networks, and another one is extended coverage.

4.4.4.1 High-Gain Antennas A very cost-effective and easy-to-implement way to extend the coverage in communication networks is to increase the antenna gain, which is valid in both uplink and downlink. Increasing the antenna gain is accomplished by reducing the beam width. One potential method of decreasing the vertical beam width is by adding two or more antenna sections above each other in a modular fashion (Figure 4.71). The antenna sections are placed in such a way that the beam patterns add in equal phase and hereby increase the gain in the main beam direction without altering the azimuth beam width. The antenna arrangement further comprises an RF feed network for feeding the separate antenna sections. The feed network may be used to control the feeding of each antenna section allowing for beam shape control. However, other parameters besides the gain may be of considerable importance, such as, for example, beam shape, side-lobe level, beam-tilt, and null-fill. Some of the actions that can be implemented in the feed network are, for example, provision of amplitude taper, phase taper, or antenna section beam-tilt.

Figure 4.71 Modular high gain antenna principle.

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4.4.4.2 Multiband Antennas Multiband implementations can be envisioned on a number of complexity levels, from integrated single band antennas to diplexing before the feed network. Depending on the frequency bands, the former may be the only feasible solution. For example, GSM-900 and UMTS frequency bands may not facilitate straightforward sharing of signal paths in the antenna unit. A major issue is the element separation, which has to be such that grating lobes are sufficiently suppressed not to generate interference or cause gain reduction. This could be handled by choosing the element separation based on the requirements set by the highest frequency. Fractal antenna elements have been suggested as one way of achieving multiband performance on element level (Figure 4.72) [71]. Other techniques include reconfigurable plasma antennas and antennas with switches based on microelectromechanical systems (MEMS) technology. However, it may still be difficult to realize an element with sufficient bandwidth; one alternative is to have separate single-band elements in a common aperture for all frequencies (Figure 4.73). 4.4.4.3 Antenna Arrangement Migration During initial rollout, capacity may not be an issue. The three-sector omnicoverage antenna configuration (Figure 4.74) offers a low-cost solution for low-capacity requirements since

Figure 4.72 Fractal antenna element in monopole arrangement over ground plane.

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Figure 4.73 Triple-band sector antenna for simultaneous operation at 800, 1,500, and 2,000 MHz.

Figure 4.74 Three-sector omnicoverage antenna configuration with all sector antennas fed by a common power amplifier.

it reduces the required number of power amplifiers by a factor of three, compared to a conventional three-sector base station arrangement. Therefore, it could be of interest as a solution for early and rapid deployment. However, the interferometer effects caused by radiating the same signal from more than one sector antenna will put requirements on the minimum antenna separation distance for the omnicoverage solution. When more capacity is needed, the base station can be reconfigured to have three independent sectors. Array antennas also offer the possibility of implementing capacity improvements without changing the antenna unit hardware. The basic idea is to install antenna equipment that can provide one or more beams. This is illustrated in Figure 4.75, where a dual

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Figure 4.75 Dual polarized four-beam array antenna with Butler matrix (BM) RF beam-forming, configured for three different levels of system complexity: (a) one sector-beam, one TX-chain, 2 RXchains; (b) two pairs of beams, two TX, four RX; (c) four narrow beams, four TX, eight RX. Solid lines below Butler matrices indicate RX and TX; dashed lines indicate RX only. Effective radiation pattern behavior is sketched on the right.

polarized four-beam antenna is used for three different levels of system complexity. The lowest-complexity solution (i.e., sector coverage) is generated by connecting beams with alternating polarization, which is used to avoid undesired beam-forming by adjacent beams [Figure 4.75(a)]. If the side-lobe contributions are negligible, the net effect is that the beams are power-combined, and the radiation pattern is similar to that of a sector antenna. The next level of complexity is generated by combining pairs of beams [Figure 4.75(b)]. Again, to avoid undesired beam-forming, beams with orthogonal polarization can be used. Finally, the shown highest level of complexity is to use directly the beams that are generated by the Butler matrices [Figure 4.75(c)]. This could represent the final stage of a configuration scheme where the system is migrated in the direction of increasing capacity. A number of migration scenarios are possible, where the issue of simultaneous dedicated and common channel transmission using array antennas need to be considered. 4.4.4.4 Antenna Evolution Different possibilities for the evolution of base station antennas for future mobile communication systems are as follows:

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Advanced antenna concepts are proposed in system standardization and considered in network planning. MIMO systems offer higher capacity gains by using multiple antennas at both ends of the wireless communication links. Higher capacity requirements lead to larger demands on interference suppression, which can be improved by spatial filtering through adaptive antennas or smaller cells. More diverse antenna configurations—for example, distributed antenna systems (DAS)—are expected to be seen as different preferred solutions on different locations are identified. Migration possibilities of base stations and antenna configurations on site are important to facilitate the step from low capacity installations to high capacity installations. Real-time tuning of antenna parameters such as beam width, beam direction, and electrical down-tilt are introduced to optimize the cell plans when changes occur, even for changes from hour to hour. Multistandard and multifrequency antennas are sought for as more standards and more frequency bands are introduced. Antenna sharing is expected to take place between different operators, as site acquisition and site costs increase. Cylindrical antennas are a flexible and aesthetic technology for future installations. Adaptive antenna configurations are implemented in ad hoc/multihop network.

REFERENCES [1] Fujimoto, K., and J. R. James, Mobile Antenna Systems Handbook, 2nd ed., Norwood, MA: Artech House, 2001, Section 4.3.1, pp. 179–181. [2] http://www.tele.soumu.go.jp/rfeq/index.htm. [3] Iwamura, M., et al., ‘‘Optomal Beamwidth of Base Station Antennas for W-CDMA,’’ IEICE National Convention Record, 1999, p. 508 (in Japanese). [4] Fujimoto, K., and J. R. James, Mobile Antenna Systems Handbook, 2nd ed., Norwood, MA: Artech House, 2001, Section 4.3.3, pp. 192–193. [5] Nara, T., Y. Ebine, and N. Nakajima, ‘‘Beam Tilting Effect of Base Station Antenna,’’ IEICE National Convention Record, 1985, No. S5-15 (in Japanese). [6] Fujimoto, K., and J. R. James, Mobile Antenna Systems Handbook, 2nd ed., Norwood, MA: Artech House, 2001, Section 4.3.4, pp. 195–196. [7] Clark, R. H., ‘‘A Statistical Theory of Mobile-Radio Receptions,’’ B.S.T.J., Vol. 47, No. 6, 1968, pp. 957–1000. [8] Yamada, Y., K. Kagoshima, and K. Tsunekawa, ‘‘Diversity Antennas for Base and Mobile Stations in Land Mobile Communication Systems,’’ IEICE Trans., Vol. E74, No. 10, 1991, pp. 3202–3209. [9] Fujimoto, K., and J. R. James, Mobile Antenna Systems Handbook, 2nd ed., Norwood, MA: Artech House, 2001, Section 4.3.4, pp. 196–199. [10] Yamada, Y., Y. Ebine, and K. Tsunekawa, ‘‘Base and Mobile Station Antennas for Land Mobile Radio Systems,’’ IEICE Trans., Vol. E74, No. 6, 1991, pp. 1547–1555.

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[11] Fujimoto, K., and J. R. James, Mobile Antenna Systems Handbook, 2nd ed., Norwood, MA: Artech House, 2001, Section 4.3.4, p. 199. [12] Fujimoto, K., and J. R. James, Mobile Antenna Systems Handbook, 2nd ed., Norwood, MA: Artech House, 2001, Section 5.1.1.3, p. 233. [13] Fujimoto, K., and J. R. James, Mobile Antenna Systems Handbook, 2nd ed., Norwood, MA: Artech House, 2001, Section 5.1.1.3, pp. 228–229. [14] Arai, H., and M. Nakano, ‘‘Up-Link Polarization Diversity and Antenna Gain Measurement of Hand Held Terminal at 900 MHz,’’ MDMC’94, Niigata, Japan, November 21–22, 1994, pp. 93–96. [15] Nakano, M., T. Satoh, and H. Arai, ‘‘Up-Link Polarization Diversity Measurement of Hand-Held Terminal at 900 MHz,’’ Technical Report of IEICE, AP95-13, 1995, pp. 13–19 (in Japanese). [16] Nakano, M., T. Satoh, and H. Arai, ‘‘Up-Link Polarization Diversity and Antenna Gain Measurement of Hand-Held Terminal,’’ IEEE AP-S, Digest, New Port Beach, July 1995, pp. 1940–1943. [17] Nakano, M., T. Satoh, and H. Arai, ‘‘Up-Link Polarization Diversity and Antenna Gain Measurement of Hand-Held Terminal,’’ IEICE National Convention, B-11, 1995, p. 11 (in Japanese). [18] Anderson, J., and F. Hansen, ‘‘Antennas for VHF/UHF Personal Radio: A Theoretical and Experimental Study of Characteristics and Performance,’’ IEEE Trans. Veh. Technol., Vol. VT-26, No. 4, November 1977, pp. 349–357. [19] Fujimoto, K., and J. R. James, Mobile Antenna Systems Handbook, 2nd ed., Norwood, MA: Artech House, 2001, Section 5.1.1.3, pp. 234–238. [20] Moriyasu, S., et al., ‘‘Printed Dipole Pair with Modified Parasitic Elements and Reflector,’’ IEICE National Convention Record, 1999, B-1-150 (in Japanese). [21] Karikomi, M., and Y. Ebine, ‘‘Characteristics of Printed Dipole Antenna with Parasitic Elements,’’ Technical Report of IEICE, AP82-2, April 1989, pp. 9–16 (in Japanese). [22] Tefiku, F., and I. Oshima, ‘‘Feeding Technique for Arrays of Double-Sided Printed Strip Dipoles,’’ National Conference Convention of IEICE, B-47, 1996 [23] Michishita, N., et al., ‘‘FDTD Analiysis of Dual Frequency Printed Dipole Antenna,’’ IEICE Trans. Commun., Vol. J86-B, No. 1, September 2003, pp. 68–75 (in Japanese). [24] Sugimoto, Y., and Y. Ebine, ‘‘Design of a Triple Frequency Resonant Base Station Antenna for Cellular Mobile Radios,’’ Technical Report of IEICE, AP99-262, March 2000, pp. 81–85. [25] Nakamura, K., K. Komada, and Y. Ebine, ‘‘A Triple-Band Antenna for Mobile Base Stations,’’ National Society Convention of IEICE, B-1-66, 2000 (in Japanese). [26] Sugimoto, Y., and Y. Ebine, ‘‘A Triple Frequency Base Station Antenna with 60° and 120° Beam-width in Horizontal Plane for Cellular Mobile Radios,’’ Technical Report of IEICE, AP99-47, July 1999, pp. 35–42 (in Japanese). [27] Kakefuda, H., M. Karikomi, and Y. Ebine, ‘‘A 0.8/1.5/2GHz Base Station Antenna Having Dual 60 Degrees Beams for 2GHz Band,’’ National Society Convention of IEICE, B-1-65, 2000 (in Japanese). [28] Nara, T., Y. Ebine, and N. Nakajima, ‘‘Beam Tilting Effect of Base Station Antenna,’’ National Conference Convention of IEICE, No. S5-15, 1985. [29] Ebine, Y., and M. Karikomi, ‘‘A Phase Shifter Using Meander Lines Loaded High Permittivity Dielectric Plates,’’ Technical Report of IEICE, AP2000-47, July 2000, pp. 29–36 (in Japanese). [30] Michisuta, N., H. Arai, and Y. Ebine, ‘‘Mutual Coupling Characteristics of Binominal and Chebyshev Array Using 16 Element Patch Antennas with Choke of Repeater Systems,’’ IEEE Antenna and Propagation Society Int. Symp., June 2003. [31] Ebine, Y., ‘‘Development of Fiber-Radio Systems for Cellular Mobile Communications,’’ IEEE International Topical Meeting on Microwave Photonics, November 1999, pp. 249–253. [32] Arai, H., et al., ‘‘Bi-Directional Notch Antenna with Parasitic Elements for Tunnel Booster Systems,’’ IEEE Antenna and Propagation Society Int. Symp., June 1997, pp. 2218–2221. [33] Kanemoto, T., Y. Ebine, and M. Karikomi, ‘‘A Wide-Band Inverted-F Antenna,’’ National Society Convention of IEICE, B-1-88, 2000 (in Japanese).

210

[34] Ihara, T., and K. Tsunekawa, ‘‘Indoor Base Station Antennas for Mobile Communications with Rounder Semi Circular Element,’’ Technical Report of IEICE, AP97-70, July 1997, pp. 1–6 (in Japanese). [35] Jiang, H., and H. Arai, ‘‘FDTD Analysis of Disk Loaded Monopole Antenna with Matching Posts,’’ IEICE Trans. Commun., Vol. E85-B, No. 11, November 2002, pp. 2468–2475. [36] Rikuta, Y., H.Arai, and Y. Ebine, ‘‘Two-Layer Antenna with Dual Frequency,’’ IEICE Trans. Commun., Vol. J86-B, No.9, September 2003, pp. 1825–1832 (in Japanese). [37] Mitsumoto, K., T. Matsuoka, and M. Nakano, ‘‘Dual Polarization Bi-Direction Antenna for 900 MHz Indoor Use,’’ National Conference Convention of IEICE, B-1-68, 2000 (in Japanese). [38] Fujimoto, K., and J. R. James, Mobile Antenna Systems Handbook, 2nd ed., Norwood, MA: Artech House, 2001, Section 5.1.1.3, pp. 224–226. [39] Fujimoto, K., and J. R. James, Mobile Antenna Systems Handbook, 2nd ed., Norwood, MA, Artech House, 2001, Section 5.1.1.3, pp. 227–228. [40] Fujimoto, K., and J. R. James, Mobile Antenna Systems Handbook, 2nd ed., Norwood, MA: Artech House, 2001, Section 5.1.1.3, pp. 228–233. [41] Nakano, M., et al., ‘‘Up-Link Polarization Diversity Measurement for Cellular Communication Systems Using Hand-Held Terminal,’’ IEEE Antenna and Propagation Society International Symposium, Vol. 66.8, 1997, pp. 1360–1363. [42] Nakano, M., et al., ‘‘Up-Link Polarization Diversity for Cellular Communication System,’’ Technical Report of IEICE, 1998, A-P98-15, pp. 13–20 (in Japanese). [43] Nakano, M., et al., ‘‘Small-Sized Polarization Diversity Antenna for Cellular Base Stations,’’ National Society Convention of IEICE, 1997, B-1-42, p. 42 (in Japanese). [44] Nakano, M., et al., ‘‘A 0.8MHz/2GHz Dual Band Polarization Diversity Antenna for Base Stations,’’ National Society Convention of IEICE, B-1-138, 2004, p. 138 (in Japanese). [45] Fujimoto, K., and J. R. James, Mobile Antenna Systems Handbook, 2nd ed., Norwood, MA: Artech House, 2001, Section 5.1.2.2, pp. 239–242. [46] Uchibori, J., et al., ‘‘The Radiation Characteristics of Adaptive Array Antenna for PHS Base Station,’’ National Society Convention of IEICE, 1998, B-5-76 (in Japanese). [47] Fujitsuka Y., et al., ‘‘A Field Test for the PHS Base Station with Adaptive Array Antenna,’’ National Society Convention of IEICE, B-5-77, 1998 (in Japanese). [48] Yamazaki, C., et al., ‘‘Experimental Results on Mobility Improvement Effects of Adaptive Antenna Arrays Applied to Mobile Stations in SDMA System,’’ 2006 IEEE 64th VTC, 2006, DSP-1P-13. [49] Okino, K., et al., ‘‘Experiment and Simulation Results of Adaptive Antenna Array System at Base and Mobile Stations in Mobile Environment,’’ 2006 International Symposium on Antenna and Propagation, 2006, SA2-3. [50] ANSI ATIS 0700004-2005, High Capacity-Spatial Division Multiple Access (iBurst), 2005. [51] Okino K., et al., ‘‘Experimental Performance Results of Adaptive Antenna Array System at Base and Mobile Stations in Mobile Environment,’’ 17th IEEE PIMRC’2006, TJ1-2. [52] Tanaka S., et al., ‘‘Transmit Diversity Based on Adaptive Antenna Array for W-CDMA Forward Link,’’ Proc. CIC’99, 1999, pp. 282–286. [53] Sawahashi M., et al., ‘‘Adaptive Antenna Array Diversity for W-CDMA,’’ NTT DoCoMo Technical Journal, Vol. 8, No. 1, pp. 52–61 (in Japanese). [54] Amano, Y., et al., ‘‘A Novel Approach for Estimating Downlink Weights for FDD/SDMA based Wireless Packet Cellular Systems,’’ WPMC’03, WA12-1, 2003, pp. 425–429. [55] Nakano, M., et al., ‘‘Evaluations of SDMA Packet Cellular Systems with Circular or Planar Array Antennas for Base Station,’’ Technical Report of IEICE, A-P2005-188, pp. 93–98 (in Japanese). [56] Vaughan, R., and J. B. Andersen, Channels, Propagation and Antennas for Mobile Communications, IEE Electromagnetic Waves Series 50, London, U.K., 2003. [57] Andersson, S., et al., ‘‘Smart Antennas for GSM and TDMA Systems,’’ IEEE Personal Communications, Vol. 6, No. 3, 1999, pp. 74–86.

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[58] Winters, J. H., ‘‘On the Capacity of Radio Communication Systems with Diversity in a Rayleigh Fading Environment,’’ IEEE Journal on Selected Areas in Communications, Vol. SAC-5, No. 5, 1987, pp. 871–878. [59] Gesbert, D., et al., ‘‘From Theory to Practice: An Overview of MIMO Space-Time Coded Wireless Systems,’’ IEEE Journal on Selected Areas in Communications, Vol. SAC-21, No. 3, 2003, pp. 281–302. [60] Asplund, H., et al., ‘‘Propagation Characteristics of Polarized Radio Waves in Cellular Communications,’’ Proc. IEEE 66th Vehicular Technology Conference, VTC2007-Fall, Baltimore, MD, September/October 2007. [61] Corcoran, M., ‘‘Hiding in Plain Site,’’ Wireless Review, Vol. 18, No. 10, 2001, pp. 32–34. [62] Morgan, S. P., ‘‘Prediction of Indoor Wireless Coverage by Leaky Coaxial Cable Using Ray Tracing,’’ IEEE Transactions on Vehicular Technology, Vol. 48, No. 6, 1999, pp. 2005–2014. [63] Liberti, Jr., J. C., and T. S. Rappaport, Smart Antennas for Wireless Communications: IS-95 and Third CDMA Applications, Englewood Cliffs, NJ: Prentice Hall, 1990. [64] Sharp, E. D., ‘‘A Triangular Arrangement of Planar-Array Elements That Reduces the Number Needed,’’ IEEE Trans. on Antennas and Propagation, Vol. AP-9, No. 2, 1961, pp. 126–129. [65] Raffaelli, S., M. Johansson, and B. Johannisson, ‘‘Cylindrical Array Antenna Demonstrator for WCDMA Application,’’ Proc. International Conference on Electromagnetics in Advanced Applications, ICEAA, Torino, Italy, September 2003. [66] James, J. R., and P. S. Hall, (eds.), Handbook of Microstrip Antennas, Volumes 1 and 2, IEE Electromagnetic Waves Series 28, London, U.K., 1989. [67] Athley, F., ‘‘On Base Station Antenna Beamwidth for Sectorized WCDMA Systems,’’ Proc. IEEE 64th Vehicular Technology Conference, VTC2006-Fall, Montreal, Canada, September 2006. [68] Davies, D. E. N., ‘‘Circular Arrays,’’ Chapter 12 in The Handbook of Antenna Design, Volumes 1 and 2, A. W. Rudge, et al., (eds.), London, U.K.: Peter Peregrinus, Ltd., 1986, pp. 992–1023. [69] Tang, R., and R. W. Burns, ‘‘Phased Arrays,’’ Chapter 20 in Antenna Engineering Handbook, 2nd ed. R. C. Johnson, and H. Jasik, (eds.), New York: McGraw-Hill, 1984. [70] Shelton, J. P., and K. S. Kelleher, ‘‘Multiple Beams from Linear Arrays,’’ IEEE Trans. on Antennas and Propagation, Vol. AP-9, No. 2, 1961, pp. 154–161. [71] Puente-Baliarda, C., et al., ‘‘On the Behavior of the Sierpinski Multiband Fractal Aantenna,’’ IEEE Trans. on Antennas and Propagation, Vol. AP-46, No. 4, 1998, pp. 517–524.

Chapter 5 Antennas for Mobile Terminals Zhinong Ying, Masataka Ohtsuka, Yasuhiro Nishioka, and Kyohei Fujimoto

5.1 BASIC TECHNIQUES FOR MOBILE TERMINAL ANTENNAS 5.1.1 General With the remarkable progress in mobile communications systems, antenna systems have also advanced in the recent decade. The typical mobile systems are mobile phone systems, evolved from analog systems, called the first generation (1G) systems, to digital systems, called the second generation (2G) systems, and further to the third generation (3G) systems, which are capable of multimedia transmission. Now the 3G systems are advancing to the fourth generation (4G) systems through 3.5G systems, which are designed to stand between 3G and 4G systems in order to advance smoothly from 3G systems to 4G systems. In addition to the mobile phone systems, various wireless mobile systems (WMS) have been deployed and offer services in various areas. The services of WMS range from very short distances to intermediate distances, whereas mobile phone systems provide nationwide service. The operating frequencies used by WMS ranges from the kilohertz regions to as high as the gigahertz regions, depending on the system performance, complexity, transmitting media, and data. The WMS provide not only communication services, but also perform control, data transmission, identification, and sensing, either through their network or with their own structure. Typical short range systems are Bluetooth, near field communication (NFC) systems (including RFID), and UWB. Middle to long range systems include wireless local area network (WLAN), wireless metropolitan area network 213

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(WMAN), and mobile worldwide interoperability for microwave access (WiMAX) as the middle to long range systems. The WLAN, WMAN, and WiMAX systems feature in high data-rate transmission, even in motion with high speed, although the service areas differ. Various antenna systems have been developed for these mobile systems, and accordingly the antenna technology has made progress along with the deployment of these systems. Currently there have been two major trends in antenna design. One trend is in antennas for mobile phone systems, which require small, built-in, and multiband operation. Another trend is in antennas for WMS, which require various antenna performances, depending on the function and complexity of the system, its service areas, and the quality and quantity of data to be transmitted. Many WMS operate in shorter ranges and mostly indoors. Some of them have a simple structure and operate in very short distance, ranging from a few centimeters to a few meters. Other WMS, however, have relatively complicated structure and operate in longer distances, ranging from a few meters to a few kilometers. Types of antennas for these systems vary from very small ones to conventional ones. Most of them for the short range systems need small antennas, to which specific design is generally necessary to meet the system requirements. Nevertheless, the typical trend common to both mobile phones and WMS is personalization, which has been accelerated by the personal use of small mobile terminals that offer users easy access to personal information, media, and data. Antennas commonly required for WMS terminals are generally small size, compact, and lightweight, but yet remain functional. Recent mobile phones employ a built-in multiband antenna that can cover frequency bands of various systems such as GSM (800-, 900-, 1,800-, and 1,900-MHz bands), UMTS (1.8-, 1.9-, 2.1-, and 2.5-GHz bands), and GPS (1.5-GHz band). In addition, modern mobile phones install some functions other than telephony; for instance, entertainment functions such as games, television reception, and music listening, as well as functions that concern business, industry, and life, including electronic keying, electronic banking, sensing, and identifying, in addition to data transmission. One typical trend these days is to attach function to mobile terminals such as Bluetooth, NFC systems, including RFID, and UWB systems. Accordingly, the development of very small, yet compact and lightweight multiband antennas has been accelerated. There have been many mobile terminals incorporating functions other than telephony and hence employ a small antenna. In addition, increased demand attributed to required services for multimedia and large quantities of data has sped deployment of high-speed, high data-rate transmission systems, to which advanced antenna systems such as adaptive arrays and MIMO systems have been adopted. The propagation problem should include parameters related with not only path loss, but also signal transmission rate, bandwidth, delay spread, and Doppler shift in a Rayleigh fading environment. These parameters are particularly important in the digital modulation systems used in high data-rate transmission. Antenna design depends on the specifications required for the particular system in which antennas are employed. Interference rejection and mitigation of any unfavorable effects caused by the multipath fading are included in the requirements. Environmental

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conditions are also a serious issue to be considered in antenna design. Proximity effects due to materials near the antenna element such as circuit components, ground plane, and the operator’s hand and head may degrade antenna performance, while, on the other hand, they may contribute to enhancing radiation by acting as parasitic parts of the radiator. This is encountered particularly in a mobile terminal where a built-in antenna is used. One must be able to consider the proximity effect as either an advantage or disadvantage. In modern antenna design the proximity effect is treated with the integration concept, in which nearby materials are included in an antenna system as an integral part of the radiator. The propagation path can be treated equivalently as a transmission circuit, and the evaluation of it will assist in optimization of the communication link, by which the best signal transmission is realized. This is essential in designing communication links, especially in the Rayleigh fading environment and diversity antenna system. It is also true for a link using MIMO systems. Another important issue to be considered in designing antennas for mobile terminals is reduction of specific absorption rate (SAR) values, which should be as low as possible, especially against the human brain. Nowadays, it is a rather common understanding that an antenna is not an isolated element, but a system in which such factors as propagation concerns, system requirements, and environmental conditions are involved [1]. This concept is illustrated in Figure 5.1. 5.1.2 Brief Historical Review of Design Concept [2] Antenna technology has made progress along with the advancement of mobile phones and various WMS. Essential factors to be considered in antenna design are as follows: • • • • • • • • •

Small size; Light weight; Compactness; Low profile; Robustness; Flexibility; Multiband; Built-in; Integration with nearby materials. Two of the last items, multiband and built-in, are included according to the recent

trend. Notable changes observed in antenna design for mobile phones in the recent decade are selection of antenna type, mounting of antenna element in the mobile terminal, and the design procedure. Historically, a monopole was used in almost all portable equipment

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Figure 5.1 Parameters of the mobile terminal antenna design.

in the early days of mobile communications, and its use has been continued for a long time—although there was a misconception that the unit case could be treated as a ground plane so that a monopole would be considered as a half-wave dipole with its image. It was in 1968 when analysis revealed that the unit case should be treated as a part of the radiator and hence the monopole along with the unit case constitutes an asymmetrical dipole [3]. Later, an antenna system in which a monopole antenna is installed on a unit case was precisely analyzed and the antenna performance related to dimensions of both the antenna element and the unit case was clarified [4]. The analysis provided the design concept for antennas presently used in mobile terminals (MT) of Japanese PDC systems. In the mid-1980s, planar antennas were developed, and application of planar antennas has prevailed in portable equipment along with monopoles. In recent handheld terminals,

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planar antennas, typically a planar inverted-F antenna (PIFA), have been employed. The PIFA is an antenna that evolved from a half-wave slot on a rectangular conducting box [5]. MTs in PDC employ a built-in PIFA as a subantenna of diversity systems. Mobile phones of GSM systems have installed a built-in PIFA for more than 10 years. Their PIFAs are modified a great deal from the basic PIFA structure: the antenna no longer has a rectangular shape and rather has slots on the planar element which introduce different current paths on the planar element along the slots so that resonance at multiple frequencies occurs. Use of a built-in antenna has now become a worldwide trend in mobile phones. Figure 5.2 illustrates the historical trend of antenna design for mobile terminals. 5.1.3 Modern Antenna Technology The types of antennas used in mobile terminals of today’s 3G systems are not only a modified PIFA, but also a planar meander-line, a folded loop, modified dipole, and so forth. The design of these antennas is essentially based on the concept of small antenna technology, as the small dimensions are of prime importance, and yet these antennas

Figure 5.2 Historical trend of antenna design for mobile terminals.

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should have reasonably high gain and wide bandwidth for the practical applications. To make antenna dimensions small, one of the essential concepts is to construct an antenna element with a slow wave structure. The typical example is application of a traveling wave structure (e.g., helix, meander-line, zigzag, and so forth) in which the current path on the antenna structure is effectively extended so that the resonance occurs at lower frequency, even though the physical dimensions of the antenna are kept small. Another way of achieving slow wave structure is integration of device(s) or circuit(s) into the antenna structure, by which phase of the current on the antenna structure is altered so that the electric length is effectively increased and the resonance occurs at lower frequencies without change in the physical dimensions of the antenna. Another significant issue in using built-in antennas is the proximity effect, as it effectively determines the antenna performance, because of the current flow on the proximity materials. Materials existing near to the antenna element should be treated as part of the radiator and hence these are integrated into the antenna design. Typical materials are the ground plane, housing, and electronic components and accessories. Accessories such as a camera and a speaker, usually located near the antenna element, should also be included in the antenna design. Again it should be noticed that these materials either degrade antenna performance due to their loss or enhance it as a part of radiator. This integration technology is very important and should be used practically as one of the modern antenna design concepts. Among influential materials on the antenna performance, the ground plane is the most significant one. It can serve to enhance the radiation, or it may degrade antenna performance due to the body effect arising from operator’s hands, which may vary the current distributions on the ground plane and absorb the radiation power. One of the methods to mitigate the influence of the body effect is to reduce variation of the current distributions on the ground plane. This can be achieved by introducing a balanced structure to the antenna system [6]. This concept is another significant one in antenna design. Small antenna technology is also needed for designing the antenna systems for mobile wireless systems, which employ adaptive antenna systems and MIMO systems, where multiple antenna elements are used. An advanced concept of integration includes application of nonperfect electric conductor (PEC) as the ground plane. One of the typical examples is application of the electromagnetic bandgap (EBG) surface [7, 8]. Evaluating antenna performance has made progress according to necessity encountered in the conventional measurement. One of the serious issues is error caused by the proximity effect on the antenna performance. When evaluating a handset antenna performance, a voltage controlled oscillator (VCO) is often used inside the handset, instead of connecting a signal generator to the test antenna, in order to avoid some error due to the proximity effect, especially influence of the cable. A potential disadvantage of using a VCO is that the VCO itself could disturb the precise measurement, as the VCO body may act as a part of the radiator. In addition, no reference signal is available to the receiver by this means and thus the phase information in radiation patterns, for example, cannot

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be obtained. Furthermore, the measurement is limited to only one VCO frequency at one time. This would become serious disadvantage in designing diversity systems and multiband antennas. To replace the VCO, a method which uses an optical fiber cable has been adopted to the practical antenna measurement. A cable, connecting a test antenna and the signal generator, is replaced by an optical fiber so that no interference occurs and also that the reference signal, including information of phase as well as amplitude is available. A small electrical-to-optical (E/O) signal converter and a small optical-to-electrical (O/E) signal converter are used to transform a RF signal to an optical signal and vice versa. Then the RF signal (output of the O/E converter) excites the test antenna and the signal is transmitted to the receiver. By this system, multiple frequency measurement is possible and a significant time saving for the measurement can be expected. This method is discussed in Section 5.4. Another advanced measurement system is the reverberation chamber [9], in which a random field environment can be simulated and the evaluation of antenna performance in the Rayleigh fading condition can be performed. Notable advantages of this method are found in various ways: evaluation of small antenna gain, diversity gain, maximum capacity of MIMO systems, gain of active terminals, diversity gain of active diversity systems, receiver sensitivity, and so forth. The details of this method are discussed in Section 5.2. Electromagnetic simulators have made much progress and analysis of antenna systems used in an MT can be made with enough accuracy to verify the experimental results. The typical simulators are finite difference time domain (FDTD) method, method of moment (MoM), and finite element method (FEM). These are useful tool that assists in the design of antennas for MT, even when they have a complicated structure, having some material and also the human body near the antenna structure. Further advanced antenna systems expected in the near future may include the capability of controlling antenna performance adaptively to the environmental conditions so that degradation of antenna performance is minimized. Another notable example is a reconfigurable antenna system, which varies its performance to match the specified systems by reconfiguration of the antenna structure. For example, an antenna structure of a mobile phone system, operating in an area of the PDC system, is reconfigured to correspond to a CDMA system when it enters into and operates in a CDMA system’s area. The antenna structure can be reconfigured by means of switching, to which RF MEMS may be applied. Further development of advanced novel antenna systems are expected and will be introduced practically in small mobile terminals in the near future. 5.2 DESIGN AND PRACTICE OF ANTENNAS FOR HANDSETS I Since the 1980s, the mobile industry has experienced a dramatic growth. The first step was the development from the analog standard to the digital standard—for example,

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from analog standards such as Advanced Mobile Phone System (AMPS), Nordic Mobile Telephone (NMT), and ETACS to digital standards such as GSM, D-AMPS, and CDMA. The second step was the development from the single frequency band to multifrequency bands due to growing capacity requirements. For example, DCS (GSM 1800), PCS (GSM 1900), and GSM 850 were introduced since the mid-1990s. The third step was the development from voice to multimedia systems, 3G systems such as WCDMA, and their enhanced systems, which were introduced shortly after the beginning of the twenty-first century. Furthermore, the WCDMA system is proposed to be expanded to all cellular bands in the coming years. At the same time, more and more noncellular communication wireless standards have been introduced to the handset, such as FM radio, GPS, Bluetooth, WLAN, Wi-Fi, DVB-H RFID, and UWB. The trend of future mobile handsets will be a need for more integrated antennas for cellular and noncellular bands, diversity, or MIMO applications. In combination with many integration problems and the demand of an attractive industry design of the mobile terminals, practical antenna design work has become increasingly challenging. This section will discuss the multifrequency band antenna technologies for the mobile handset, size reduction techniques, antenna integration techniques in mobile handset, multichannel antenna system, diversity, human body effect, measurement techniques, and other practical issues. Section 5.2.1 will deal with some fundamental issues of small terminal antennas. Section 5.2.2 will describes the progress of different multifrequency band techniques for handset antennas. Section 5.2.3 will discuss the antenna integration issues and some practical engineering issues for the mobile terminal antennas. Section 5.2.4 will describe multichannel antenna systems, diversity, and MIMO. Finally in Section 5.2.5, the human body effect and some measurement techniques will be discussed. 5.2.1 Some Fundamental Issues 5.2.1.1 Downsizing Techniques for Terminal Antenna The small antenna types can be classified according to their geometry: dipoles, slots, and cavities. From these fundamental antenna types more complex geometries can be developed. The simplest omnidirectional type of antenna is the dipole. The external antenna on a mobile terminal can be considered as an unbalanced dipole. Usually we call it a monopole antenna, because the antenna element is much smaller than the actual handset chassis size. Slot antennas, also called magnetic dipoles, can be seen from a long, narrow opening on a metallic surface. Notch antennas and IFA antennas are type of slot antennas. The planar inverted-F antenna (PIFA) can be considered as a mixed dipole and slot antenna. The cavity antenna in its simplest cases can be a patch antenna or a DRA antenna. Many size reduction techniques for small antenna have been proposed and can be found in the literature [10]. The common techniques applied to reduce antenna size are folding configurations, surface etching, shorting walls or pins, or utilizing high dielectric

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material loading. However, there is always some performance degradation when reducing the size. For stubby antennas, wire folding and dielectric material loading are typical ways of reducing the antenna size [10]. For patch antennas, some basic design guidelines have been summarized: • • • •

A multilayered structure to obtain a multiresonant behavior; Cutting slots as a way of increasing the electrical path while maintaining the physical size; Short-circuiting wall or pin to utilize the half size reduction factor; Dielectric material loading, especially in packaged applications.

5.2.1.2 Physical Limits of a Mobile Terminal Antenna The performance of an electromagnetic passive device is sensitive to its electrical size compared to the wavelength; that is, given an operating wavelength and certain performance requirements, a small antenna cannot be made arbitrarily small. The bandwidth, losses, and dimensions of the antenna are closely interrelated. When the antenna size is smaller than a half wave dipole, the performance (bandwidth and efficiency) will be reduced when size is reduced. Due to the existence of losses in the antenna (the ohmic loss will increase when the antenna size is reduced) and impedance mismatching, only a part of the power sent by the transmitter is radiated into open air. Losses such as ohmic loss, passive loading loss, and mismatch loss in the antenna can be characterized by the radiation efficiency, ␩ , which is defined as the ratio of the total radiated power to the net power input to the antenna from the connected feed line [11]. Another parameter for small antennas is the bandwidth, which is related to the quality factor (Q). The quality factor Q is defined as the ratio of the time-average, nonpropagating energy to the radiated power of an antenna [12]. This parameter is a quantity of enormous interest when designing small antennas because of its lower bound, which provides knowledge of how small an antenna can be constructed for a given certain bandwidth. The evaluation of antenna Q can be traced back to Chu’s classical work, which derived the theoretical value for an ideal antenna enclosed in the smallest circumscribing sphere [12]. Chu’s approximate result was quoted by others in later related works [13]. In 1996 McLean [14] improved the work of Chu by deriving an exact result of the radiation Q-value using the fields of the TM01 mode directly. McLean’s result is nearly the same as Chu’s for very small antennas and predicts a slightly higher lower bound for antennas approaching a length of about 1/3 wavelength. To obtain a Q approaching the theoretical limit, an antenna must effectively use the entire volume within the bounding sphere. In this way, to use the handset chassis as the effective part of the antenna is the way to enhance the bandwidth. When the antenna is small compared with wavelength (i.e., ka Ⰶ 1), the maximum fractional bandwidth BW and its radiation efficiency ␩ have the following relationship:

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BW ⭈ ␩ ≅ (ka )3

(5.1)

where k is the wave number at the operating frequency (k = 2␲ /␭ ), and a is the diameter of the equivalent volume of the antenna. This implies that a larger bandwidth can be gained at the higher loss and lower radiation efficiency, once the size of antenna is constrained [13]. 5.2.1.3 Impact of the Ground Plane Size and Phone Form When considering antennas for mobile communications terminals in the practical case, the whole terminal, and even the human body, have their contribution to the radiation and losses [15]. The effective antenna size should be equivalent to the antenna element size plus a part of the ground plane (handset chassis). It is not a simple task to calculate the equivalent size of the antenna a in (5.1) in a real case. It is important to determine the fact that integrating an antenna in a terminal will affect its actual behavior regarding both bandwidth and radiation characterizations. The EM field generated by the antenna element will induce currents that flow on the different components of the terminal, in particular on a conductive chassis [16]. Figure 5.3 shows an example of current distributions at low and high bands of a PIFA antenna mounted on a 100 × 40-mm PWB of a bar phone. The front side of the PWB has less current compared with the PIFA antenna side, especially in the high band; this produces less body loss with a PIFA antenna in the talking position compared with a monopole antenna. The radiation patterns of the handset at 900 MHz and 1,800 MHz are shown in Figure 5.4 and Figure 5.5. It has a nearly omnidirectional pattern at 900 MHz, and has an irregular directive pattern at 1,800 MHz. The ground plane sizes have influence upon the matching characteristics, the impedance bandwidth, the radiation patterns and the interaction with the user. When the length of the handset is close to a half wavelength at the operating frequency, the antenna has a wide bandwidth and a uniform radiation pattern. Recent work has considered utilizing the ground plane of the terminal as a means to optimize the available performance, with the antenna element merely acting as a matching element to the wave modes of the metallic parts of the terminal [17–19]. Different phone forms have different ground plane features, which will affect the antenna performance. Bar-Type Handsets Most mobile phones are bar type since this type is conducive for implementing the components and functions that comprise a multimedia handset. The size of a handset is typically about 100 mm long and 40 mm wide. The length is not enough to get good bandwidth in both the low and high bands, so the antenna should be mounted at the end of the phone. For PIFA type antennas, the optimal length of ground plane is about 120 mm for the low bands and 80 and 140 mm for the high bands, as shown in Figure 5.12.

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Figure 5.3 The current distribution of a PIFA handset antenna on a finite ground plane: currents at the (a) back and (b) front sides of handset at 900 MHz; and currents at the (c) back and (d) front sides of handset at 1,800 MHz. The PWB contributes greatly to radiation from the whole handset. The current at the front side is less than at the back side. It has different distributions in low and high bands.

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Figure 5.4 The radiation pattern of a handset at 900-MHz band of the antenna shown in Figure 5.3. It has a uniform omnidirectional and nearly linearly polarized pattern.

Figure 5.5 The radiation pattern of the handset at 1,800 MHz of the antenna in Figure 5.3 has some irregular shape, the pattern is more directional, it has more minimums, and the pattern usually has high cross-polarization.

The artificial extension of ground, such as by adding a flange or slot flange or by adding a spiral loading, will improve the bandwidth of the antenna [20]. The grounded resonant metal structure (parasitic metal element), which is excited by the antenna element, can also improve the bandwidth. A monopole antenna has good performance when the ground plane is larger than quarter wavelengths. When the length is close to 120 mm, it has good bandwidth for both low and high bands. A matching network could be found to match the chassis to enhance the bandwidth if the length of PWB is between quarter and half wavelength. A monopole antenna has a quite strong induced current on both sides of the

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chassis, concerning the head loss in the talking position; the internal monopole antenna is usually mounted at the bottom of the handset. Clamshell Handsets The clamshell handset has different ground plane lengths at open and closed modes: about 80 mm when it is in the closed mode and 160 mm in the open mode. The length change will strongly affect the impedance matching of the antenna. In the open mode, the handset is composed of two small metal parts which are connected by a flex-circuit via the hinge; the two parts of the ground are inductively connected with some capacitive coupling which has a certain characteristic impedance. The length of the flex can affect the feature of the ground plane. Clamshell handset could have their antenna position at the top, on the hinge, or at the bottom of the handset. The ground plane has different impacts for different antenna positions. When the antenna is located near the hinge, the phone has good performance in free space if the hinge has less metal, but the drawback is high head loss. A metal hinge can usually improve the ground plane quality when the antenna is located at the end. Slider Handsets A slider handset also has two modes, open and closed, which will change the ground plane length. The upper parts and lower parts are coupled to each other and are DC connected by a folded flex-circuit. The impact of length change of ground plane can be controlled by introducing an LC loading circuit to minimize the negative impact of the imperfect ground plane in this case. 5.2.1.4 Extendable Antenna In all cases the near field is a concern when the human body is included. Less current on the chassis is desired in this case. To overcome the small antenna limitation and reduce the human body absorption and the near field (e.g., hearing aid compatibility will be discussed in 5.2.5.3) effects, an extendable antenna was commonly used, especially on lower cellular frequency bands. In the retracted mode, the bottom antenna, such as a helix or a meander, acts as a stubby antenna. In the extended mode, the whip antenna can significantly reduce the induced current on the phone chassis; and it has higher radiation efficiency in the talking position. The user absorption can be significantly reduced. An example of an extendable antenna is shown in Figure 5.6. A dual band mobile telephone will increase the complexity of the extendable antenna. The antenna has to meet the requirement for dual band performance in both retracted and extended modes. There are different ways to solve this:

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Figure 5.6 An example of an extended whip antenna for a mobile phone: the antenna has a stubby element in the retracted mode, and has higher efficiency in the extended mode especially in the low frequency when the phone size is less than 1/2 wavelength. (Courtesy of Laid Technology.)

1. To find a matching network for the whip and the stub—this is not always easy since the whip and the helical stub antenna have different lengths; 2. To use the capacitive coupling method and couple the whip to the base antenna [22]; 3. Mechanical switching for different matching networks for the whip and the stub. 5.2.2 Various Multiband Antenna Concepts Multiband antenna technology is an interesting topic for antenna engineers. This type of technology is widely used in multiband wireless applications. Since the mid-1990s, due to the demands of multiband and multistandard in mobile applications, multiband antenna technology has become one of the key technologies in a modern mobile handset. In this section, we will describle various types of multiband techniques which have been developed and widely used in the mobile industry since the mid-1990s. This reflects the evolution and trend of this antenna technology in the mobile industry. 5.2.2.1 The External Antenna In 1996, Z. Ying first proposed a dual band nonuniform helical antenna [23, 24], which later became a popular antenna for dual band mobile phones worldwide. The helix has quarter wavelengths at the low band functioning as a quarter wave monopole, and it has

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a nonuniform pitch angle or diameter to control the second resonance frequency band. The antenna has a high efficiency and is cheap to manufacture and it has been used in more than a billion mobile terminals worldwide. The antenna prototype is shown in Figure 5.7. At almost the same time, P. Haapala proposed a dual band mono-helix antenna [25, 26]. It is a helix that has a straight wire in the center axis. The antenna has the helix operating at the low band, and the center wire operating at the high band, both nearly quarter wavelength. This antenna is worth mentioning because it has also been used in a large number of cellular phones during the 1990s. The antenna structure is shown in Figure 5.8. G. Hayes proposed a meander or helix with a parasitic element to realize dual band performance [27]. In this case the total length of the antenna is a quarter wave monopole at the low band, and the parasitic element creates the second resonance at the high band. C.G. Blom proposed a spiral broadband impedance transformer to realize a good dual band stubby antenna, which has been in mass production [28]. In 1997, Z. Ying proposed a branch multiband antenna [29], which has the long branch about quarter wavelength at the low band and the short branch about quarter wavelength at the high band. By using the printing technique, the meander flex film is easy to make and has a good repeatability. The antenna can be rolled to be a stubby antenna or folded into a certain low profile shape in the chassis to be an internal antenna in the mobile terminal, as shown in Figure 5.9. This type of antenna becomes the basic

Figure 5.7 A nonuniform dual band stubby antenna, which was invented in 1996 by Z. Ying (Ericsson), Patent number US-6212102, WO-9815028. It has high efficiency, and it is cheap and easy to manufacture. It has reached a penetration of more than several hundreds of million products worldwide. (Source: [24]. Courtesy of Sony Ericsson.)

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Figure 5.8 A mono-helix antenna. It was a popular stubby dual band antenna which was invented in 1996 by P. Haapala [25].

Figure 5.9 A printed branch meander multiband monopole antenna, which can be rolled to be a stubby or folded to be internal antenna [29].

variant of the internal branch meander monopole antenna later on. J. Andersson has proposed several broadband low profile antennas since 1999 [30]. Later, those antennas have been proposed to be implemented popularly as low-profile monopole internal antennas in mobile terminals [31]. 5.2.2.2 The Internal Antenna The internal antenna can increase the mechanical robustness of a mobile terminal, and the internal antenna housing can also be used as an acoustic cavity to improve the audio

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performance. It follows the trend, since the late 1990s, of mobile phones becoming multimedia mobile handsets. These factors have led to a widespread market acceptance worldwide. Today, the external stub antennas are used only in the low-end models and in clamshell phones in the Asian and U.S. market. The internal antenna needs certain cubic volumes to have a good performance, so the bandwidth is limited because of the size restriction. The PIFA type antenna is more easily detuned when the user places a finger near the antenna element as compared with a monopole antenna. Due to the integration with other components in the terminal, the optimization of an internal antenna becomes more complex and time consuming compared with an external antenna. The main types of internal antennas for stick-type handsets are the PIFA antenna [32] and the folded monopole antenna [31]. The PIFA antenna is usually on a ground plane, and it may have a feeding pin and several ground pins. The radiation pattern is affected by the ground plane, which can be directive, especially in the high frequency range. Figure 5.10 shows an example of a mobile phone equipped with an internal PIFA antenna. The antenna is located at the top of the phone behind the PWB. The folded monopole antennas are usually used at the bottom of the handset to have less head loss, as was mentioned earlier. Foldable type phones have different designs such as clamshell, slide, and swivel styles. The internal antenna design in those is even more complex. The foldable feature creates changeable ground plane sizes, which makes antenna design more complicated. The different antenna locations will give different performances: 1. An antenna at the top of the phone: In this case, the antenna design is similar to a stick phone, and the PIFA antenna is the major antenna concept because of the head absorption. Because of the limited thickness of the phone, the antenna bandwidth is small.

Figure 5.10 A stick mobile phone having an internal multiband PIFA antenna; the simulated currents can be found in Figure 5.3. (Courtesy of Sony Ericsson.)

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2. An antenna at the middle of the phone: In this case the IFA and folded monopole are the major solutions. Often these include meander patterns and/or multiple branches to achieve multiband performance. It can offer a good bandwidth and good radiation performance in a free-standing position, but it has greater body loss in the talking position and more open/close matching shift or detuning. 3. An antenna at the bottom of the phone: Both monopole and PIFA can be used. Good radiation and bandwidth can be achieved if the hinge is well designed. Sometimes it is in conflict with the system connector. 5.2.2.3 The Dual Resonance PIFA In 1997 P. S. Hall proposed a PIFA composed of two separate patches of different sizes to achieve dual band performance. The antenna has two separate feedings: one for the low and one for the high band [33], and the initial idea of a common feed for a dual band PIFA antenna was mentioned. In most of the applications, it is necessary to have a common antenna feed point for the dual band wireless terminals. A lot of work was done to realize this for cellular application in later years. Between 1997 and 1999, several dual band PIFA antennas based on a slot cutting patch were developed in the mobile phone industry, and many patents were filed [34–37]. The antenna can meet the GSM/DCS dual band application with a 5-6 cc antenna volume. I. A. Korisch proposed a J shaped dual band PIFA in July 1997; I. Pankinoha proposed a switchable PIFA for multiband application in July 1997; Z. Ying proposed an internal printed spiral antenna and a dual band printed antenna in July 1998. The antenna has two branches, which have different lengths. It functions as a shunt-fed monopole antenna, and the two branches give dual band performance. A slot cutting version was proposed later in 1998 by Z. Ying [38]. In the beginning of 1999, S. Tarvas proposed a similar antenna which has a single feed and a single shorting pin [39]. The antenna was used in the first commercial dual band mobile phone with an internal antenna. The antenna current distribution at the 900-MHz band is shown in Figure 5.11. The antenna has the long arm working at the low band and the short patch at the high band. The feeding pin and ground pin form a loop. The loop size will determine the inductance that can compensate the patch capacitance to get a good matching. The bandwidth of a PIFA antenna has been studied widely in recent years [40–43]. The following factors are important for the bandwidth: 1. The antenna element size: The bandwidth of a PIFA antenna depends on the size of the antenna element (i.e., the patch size and the thickness of the PIFA). The meander shaped or spiral shaped loading can help to minimize the antenna size, but the bandwidth of the antenna will be reduced. A typical dual band PIFA antenna element for a cell phone is about 20 × 40 mm2. The thickness of PIFA has more weight to enhance the bandwidth. The typical thickness of PIFA for a cellular band

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Figure 5.11 The current distribution of a branch dual band PIFA antenna in a mobile phone at the 900-MHz band. It is a typical branched PIFA structure, with the long branch resonant at the 900-MHz band.

is 4 to 10 mm. The larger the antenna volume is, the larger the bandwidth that can be achieved. Thickness is more important compared with the patch area. 2. The feeding position: The feeding position of a PIFA antenna will affect the bandwidth. Usually the feeding point should be located at the edge of the PCB. The position is located at the end of the handset for cellular antennas. 3. The PWB size: As mentioned in the previous section, the PWB ground plane is a part of the antenna in a mobile terminal. It has a different influence on the 900-MHz band and the 1,800-MHz band. Figure 5.12 shows the bandwidth of a 16 × 38 × 8 mm3 dual band PIFA antenna on different sizes of the ground plane. It was found that the bandwidth has a maximum when the PWB length is about 120 mm for GSM 900 MHz, while for DCS 1,800 MHz the bandwidth is relatively stable; a good bandwidth is observed with the lengths 80 and 140 mm, respectively. Many products have been developed by using the branch PIFA antenna design. In Figure 5.13, some modified versions of this type of antenna with specific features are shown in [45–47]. Figure 5.13(a) shows a typical branch PIFA which has the long arm for the 900-MHz band and the short arm for the 1,800-MHz band; the extra small part can improve bandwidth at the high band. Figure 5.13(b) shows the long arm in the branch PIFA shaped to optimize the radiation pattern with more directivity at the high band to have a lower head loss [45]. Figure 5.13(c) shows the very popular center-fed C design, which allows the low band branch to resonate at high band frequencies to gain bandwidth [46].

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Figure 5.12 The bandwidth of a dual band PIFA antenna depends on the length of PCB (ground plane). The maximum bandwidth for 900-MHz band occurs when the PCB length is about 120 mm, and the maximum bandwidth for 1,800-MHz band are at PCB length 80 and 150 mm.

5.2.2.4 The Multiresonance PIFA Antenna In order to keep a small antenna volume and still fulfill the multiband applications, more development efforts have been made to enhance the bandwidth of the PIFA antennas. One way is to cut more slots on the patch to create multiresonant traces, as shown in Figure 5.14. A slot is introduced between the feeding and grounding posts to get more bandwidth in the high frequency range [48]. Another way is to introduce a parasitic element coupling to the main antenna, which was proposed by Z. Ying and A. Dalhsto¨rm in 2000 [49, 50]. The practical antenna can meet the application of GSM/DCS/PCS triband as shown in Figure 5.15 [51, 52]. Some summarizing work was done to classify the PIFA antenna variants in the mobile terminal applications, as shown in Figure 5.16 [53]. A PIFA antenna with one slot has dual resonance, mainly used in dual band applications. The average volume of such an antenna is about 5.4 cc in the application. A PIFA antenna with two slots (one slot is between feeding and grounding posts) can have an extra resonance at the high band. It could be used in tri-band applications. The average volume of such an antenna is 6.7 cc in the application [48]. A PIFA antenna with a parasitic element can have an extra resonance at the high band. It could be used in tri-band applications, and it has the average volume of 5.2 cc [52]. The third design is a compact solution, the drawbacks of which are three connections and restrictive mechanical tolerance. The second design has only two connecting pins, it has less severe mechanical tolerance, it is simpler to make,

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Figure 5.13 (a) A typical branch PIFA which has the long arm for the 900-MHz band and the short arm for the 1,800-MHz band—the left extra small part can improve bandwidth at high band; (b) the long arm in the branch PIFA was shaped to optimize the radiation pattern with more directivity at high band to have lower head loss [45]; and (c) the very popular center-fed C design, which allows the low band branch to resonate at high band frequencies [46].

and it has good radiation efficiency. These three antennas have been very popular solutions in handset applications. 5.2.2.5 The Capacitively Fed Patch Antenna The conventional PIFA antenna is electrically equivalent to a fixed LC circuit. The bandwidth is mainly determined by the antenna thickness and the surface area of the patch (i.e., the volume of the antenna). It was discovered that in a capacitively fed patch antenna [54], by introducing a distributed C in the antenna with an optimized design, the patch antenna bandwidth could be enhanced. To meet the global system requirement to have four GSM bands and five WCDMA bands, a so-called penta-band (824–894, 880–960, 1,710–1,880, 1,850–1,990, 1,920–2,170 MHz) C-fed antenna concept was proposed in 2001 by Z. Ying and A. Dahlsto¨rm [55], then later optimized by F. Nu´n˜ez Calvo [56]

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Figure 5.14 A tri-band PIFA antenna with a slot between the feeding and ground pins to create an extra resonance in order to enhance bandwidth in high bands. (Source: [48]. Courtesy of Laid Technology.)

Figure 5.15 A tri-band PIFA antenna with a parasitic element to enhance the bandwidth in high bands [49]. The design was proposed by Z. Ying and A. Dalhstorm in 2000 and has been applied in many high end mobile terminals. A 2.4-GHz Bluetooth antenna is integrated in the same antenna package. The isolation was better than 15 dB. (Source: [49, 51]. Courtesy of Sony Ericsson.)

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Figure 5.16 The summary of slot cutting PIFA antennas: (a) A PIFA antenna with one slot has dual resonance mainly used in dual band applications. The average volume of such antennas is 5.4 cc in the application. (b) A PIFA antenna with multislots (one slot is between feeding and grounding posts) can have extra resonance at high band, and it could be used in tri-band application. The average volume of such antennas is 6.7 cc in the application. (c) A PIFA antenna with a parasitic element can have extra resonant at high band, and it could be used in tri-band application. The average volume is 5.2 cc. (Source: [53].)

and J. Berge and M. Eriksen [57]. In their designs, the dual resonance feature was found in both the low band and the high band by combining the active element resonance and passive element resonance. The antenna does not have a matching loop as in a common PIFA antenna, the upper patch being fed by a lower layer patch where they are coupled to each other. The coupling is very critical to get a good bandwidth in this design. An example of the C-fed patch antenna is shown in Figure 5.17(a), and the measured and

Figure 5.17 A capacitive fed patch antenna. By introducing a distributed C in the antenna with an optimized design the patch antenna bandwidth can be enhanced. It was proposed in 2001 by Z. Ying and A. Dalhstorm from Ericsson [55], then optimized by F. Nu´n˜ez Calvo [56] and J. Berge and M. Eriksen [57]. In these designs, the dual resonant features are found in both low band and high band. The antenna can be used for penta-band application. (Source: Courtesy of Sony Ericsson.)

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simulated VSWR are shown in Figure 5.17(b). It has a good bandwidth and can be used for penta-band applications. In 2004, D. Iellici, S. P. Kingsley, and his associates proposed a high dielectric loading capacitive feeding antenna concept [58], the capacitive feeding realized by a small dielectric pellet under the patch, as shown in Figure 5.18. By optimizing several parameters, the antenna has a small size and wide bandwidth. The manufacturing complexity and cost of such antennas are higher than that of a conventional PIFA. 5.2.2.6 The Internal Monopole Antenna The branched monopole antenna mentioned in Section 5.2.1.3 can be built with a very low profile form. Then it is possible to make it integrated with the housing of a mobile terminal. The design of the antenna is similar to the external antenna and can easily be found in [29, 30], which were proposed by Z. Ying and J. Andersson. The antenna usually has a long branch to tune the low band and a short branch to tune the high band. It can also combine a nonuniform meander and a branch concept to enhance the bandwidth [30]. When the chassis of the handset is less than a quarter wavelength, the bandwidth will be reduced. Figure 5.19 shows the bandwidth of an 15 × 7 × 40-mm internal monopole antenna on different sizes of PWB, showing that the bandwidth of the monopole is much larger than a PIFA antenna. The monopole can be capacitively coupled or matched to the terminal housing to excite the chassis mode to gain some extra bandwidth when the chassis length is larger than a quarter wavelength, by combining the antenna modes and chassis modes of the multiresonant structure. When the height of the low profile monopole is too low, the antenna becomes capacitive, an extra ground pin is needed to form a matching loop, and the antenna becomes a branch IFA or nongrounding PIFA.

Figure 5.18 Another C-fed PIFA with high dielectric loading. D. Iellici et al. from Antenova proposed a capacitive feeding and high dielectric loading concept [58]. The capacitive feed is realized by a high dielectric loading. The antenna has a small size and good bandwidth.

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Figure 5.19 The bandwidth of a 15-mm height internal monopole antenna varies with the size of the ground plane; it has good bandwidth when the PCB length is near 120 mm due to the contribution of chassis resonant modes.

Figure 5.20 shows an example of a branched monopole with a parasitic element. The long meander branch is designed for the low bands and the short branch and parasitic element are for the high bands [30]. The bandwidth of the low profile antenna depends on the height of the antenna and length of the ground plane. If the antenna height and PCB have good size, the antenna has a good bandwidth when the antenna mode is combined with the chassis mode.

Figure 5.20 A low profile branch monopole antenna which is based on the design concept by Z. Ying and J. Andersson in [29, 30]. The long meander is resonant at low band, the short branch has the first resonance at high band, and the parasitic element has the second resonance at high band. (Courtesy of Sony Ericsson.)

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Figure 5.21 shows an IFA (nonground plane PIFA) multiband antenna that was proposed by K. Ishimiya [59]. This antenna is similar in principle to the design I shown in Figure 5.21. The loop between the feed and the ground can be tuned to a second resonance on the high band; the long arm can be tuned to the low band. The short arm is tuned to the high band. The optimized design has a very good bandwidth on both the low and high band [59]. E. Cassel in 2002 proposed a folded PIFA antenna, which has multiresonance. The antenna has separated feeding and ground pins with a wide bandwidth feature [60]. Recently, K. Ishimiya and Z. Ying proposed a folded dipole antenna which can also cover five bands. The design is based on the multimodes of the loop to get one resonance at the low band and two resonances at the high band [61]. 5.2.2.7 The Tuneable and Switchable Multiband Antenna In order to enhance the antenna bandwidth, one can shift the antenna resonance with loading change by using a varactor diode or pin diode [62]. The tuning could be by changing of the loading to the ground pin or the loading in the open end of the PIFA. The short circuit tuning is based on the series capacitive or inductive loading at the narrow short circuit pin of the PIFA. The open-end tuning can be implemented using parallel capacitive loading of a completely short circuited capacitor plate. Several tunable internal antenna designs were patented in recent years. J. Hayes and others proposed a length reconfigurable PIFA antenna [63]. Z. Ying and K. Hnkansson have also proposed to tune the series loading on the parasitic element [64]. Later, capacitive loading tunable PIFA and length changeable PIFA were proposed [65, 66], and the integration of the switch on the antenna element was proposed in [67]. For semiconductor switches some additional current consumption will be introduced [62]. The loading position of those switches is important to be as close as possible to the points where there is a low current flow.

Figure 5.21 An IFA antenna which includes a branch structure and a loop. This was invented by K. Ishimiya [59]. The antenna has the long arm resonant at low band, and the short branch and loop have two resonances at high band. (Courtesy of Sony Ericsson.)

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Another issue which is of concern is the RF distortion caused by the pin-diode components. The ideal solution to replace the diode is to use RF MEMS, which have low loss, high isolation, low power consumption, and high linearity. The MEMS applications are still in the lab stage [68, 69]. The remaining issues for this technology are higher driving voltage, reliability, and packaging. 5.2.3 Antenna Integration and Some Practical Issues In this section we will discuss the integration of antennas with other nearby components in a mobile terminal. The internal antenna in a mobile terminal is not in an ideal environment, but operates, rather, in a ‘‘dirty’’ surrounding environment. Starting around 2000, multimedia features were introduced to mobile phones with ever increasing complexity. As more and more features were introduced to the handset, the internal antenna in the mobile terminal had to be integrated with other components. The components act like lossy passive loading elements and sometimes as a noise source to the antenna. They will not only detune the matching and introduce losses to the antenna, but also, because some components such as the camera, the display, the keypad, and the flex-films are high-speed digital components, they will actively emit RF noise, which will degrade the receiver performance through the antenna. In this section, some of the integration issues will be discussed. An example of such a modern mobile handset with a lot of integration is shown in Figure 5.22. In the figure, a PIFA antenna is integrated with the acoustic components (speakers). It is close to a digital camera, a vibrator and a flex-circuit. Also some practical issues about mechanical and PA will be discussed in this section.

Figure 5.22 A PIFA antenna integrated with acoustic components (speakers) and external connector, it is close to a digital camera, vibrator, and a flex-circuit. (Courtesy of Sony Ericsson.)

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5.2.3.1 The Antenna Integrated with Acoustic Components The PIFA antenna needs a certain cubic volume over the ground plane to make it work. The volume between the antenna patch and the ground plane is naturally considered to be used as the acoustic cavity in order to enhance the audio quality. The idea to have the antenna and the speaker be integrated in the same housing was proposed first by B. H. Bjerre and I. Marqvardsen [70]. This design can improve the audio quality significantly, but at the same time it introduces some extra loading on the PIFA antenna, and usually has a negative impact on its performance. The loudspeaker usually consists of a coil with a permanent magnet. It has two balanced connections to connect to the audio base band circuit on the PCB. The antenna will be detuned and have some ohmic loss due to the existence of the loudspeaker. To minimize this effect on the antenna, the speaker should not be close to the dense current area of the antenna body; a simple and direct connection should be better in this case, and a coil connection should be avoided. Some EMC filtering network may be needed for further decoupling in the speaker network. The speaker position should be codesigned by antenna engineers and acoustics engineers to optimize the audio performance with a minimum impact to the antenna. A numerical model was built in [71] to simulate the effects of the speaker as shown in Figure 5.23. The new flat panel speaker technology can possibly remove the problem. By using this technique, a flat panel (which could be the antenna carrier) has an actuator to drive and produce the acoustic signal. Z. Ying and W. Shi proposed this design concept in 2002 [72].

Figure 5.23 A dual band PIFA antenna is integrated with a louder speaker which is simulated by a coil with magnetic loading. (Source: [71].)

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5.2.3.2 The Effect of the Camera Imaging is an important multimedia feature. The camera stick phone usually has a camera lens on the backside, near the internal antenna element. An example can be found in Figure 5.22. The digital camera module is a high data speed component and will not only affect the antenna performance passively (e.g., detune the antenna, introduce ohmic losses, reduce the effective volume of the PIFA antenna) but also create RF emission to degrade the receiving sensitivity. Figure 5.24 shows an example of the emission from a 3.2-megapixel camera in a phone at low band, which can degrade the radio sensitivity performance when it is placed near an internal antenna. The emission level is 10 to 20 dB above the noise floor, which can cause serious problem on the receiving bands. So a grounding or screening of the camera should be applied when it is close to the antenna. The flex-circuit connection is critical. It should be as short as possible and well shielded. For clamshell phones, the flex-circuit near the hinge should be extra carefully designed if the camera is near the hinge position.

Figure 5.24 The spurious emission from a 3.2-megapixel camera in a mobile phone. The test signals are tested in different positions. The noise floor is at 120 dBm. The emission is quite high in some RF channels in GSM 900 MHz Rx band.

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5.2.3.3 The Effect of the Battery The battery is another component that influences the internal antenna. Some studies were performed to find the effect of the battery on a PIFA antenna in a small bar type handset [73]. The investigation shows that the position of the battery affects the high band and the Bluetooth or WLAN band performance significantly regarding the bandwidth and gain. This is due to the battery occupying a big portion of the handset, which has some induced current because of EM coupling. The battery size is usually close to a half wavelength of high bands, or 2.4-GHz band, and may have a large impact on the antenna performance in those bands. The distance between the antenna and the battery should be well defined, usually more than 4 to 6 mm. If a vertical ground screen wall is introduced, the effect of the battery loss loading can be less, but the effective antenna volume will be reduced. 5.2.3.4 The Effect of the Display A multimedia phone usually has a large display which occupies a big surface area of the handset. Since the phone body is a part of the antenna, the display has some induced currents on it, which can have some impacts on the antenna and the radio performance. First, the LCD is made of a lossy dielectric material, which is parallel to the PCB ground plane and thus can change the electromagnetic near field of the mobile handset even though the cellular antenna is not in the same side as the display. Still, it can change SAR or hearing aid compatibility (HAC) performance since the display is close to the human head. Second, the LCD is a high data speed component which may cause disturbing RF emission to the radio. Figure 5.25 shows a measurement of the emission at the 900-MHz GSM receiving band from a display in a bar type phone, which can degrade the radio sensitivity performance. The emission level is 10 to 20 dB above the noise floor, which may cause serious problems on the receiving band. The connecting flex-circuit has to be carefully designed in order to minimize the emission. 5.2.3.5 Simulation and Test of the RF Sensitive Components Some of the theory was proposed to simulate the loading effect of the nearby objects to an antenna [71]. The problem is tackled by using equivalent circuit models and matrix representations that are determined from full wave simulations. The idea is based on using port responses determined by full wave simulations or measurements. Then the model is built by using this data. These can either be discrete circuit models that can be used directly in ordinary circuit simulators such as Spice, or the data can be used by modified or specially developed software. The main advantage with this approach is that only a few full wave simulations or measurements are needed to build a model, which can be used for determining responses for any excitation and loading at the defined ports. This

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Figure 5.25 The spurious emission from a LCD display in a mobile phone. The noise floor is at 120 dBm; the emission is quite high in some RF channels in 900-GSM Rx band.

technique can be applied to the problems where the geometry is fixed and where there is a need for investigating different excitation and/or loading conditions. One example of such a problem is the testing of different filter configurations on the lines feeding the loudspeaker amplifier circuitry in order to reduce interference caused by the coupling to the antenna [71]. This shows a possible way to simulate complex antenna integration and EMI problems.

5.2.3.6 The Antenna Integrated with Switches and RF Circuits Reconfigurable antennas and tunable antennas need switches to control the radiation patterns and resonance frequencies. The integration of the antenna and switches will be combined with the integration of the RF circuit. Due to the low insertion loss, high isolation, low power consumption, and high linearity, MEMS is the candidate to replace semiconductor switches in this application. Also, MEMS can remove the risk of harmonic emission from semiconductor switches [68, 69]. An antenna integrated with RF circuitry may reduce the insertion loss and enhance the RF performance. This kind of integration will be very useful for future multiband and multistandard radio. Adaptive matching circuits and software radio are required for those applications [74]. This presents a challenge for developing radio technology.

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5.2.3.7 Mechanics and Quality Issues Once the antenna is electrically designed for a terminal, a big portion of the work is put on the mechanical solution to realize the electrical design. There are many practical issues that have to be considered to ensure a high quality antenna, especially for an internal antenna in a mobile phone. Since it is an extremely high-volume product, the mechanical contacting and tolerance are very important to make a good product. To realize good quality in an antenna, the design has to be as simple as possible, with as few contact points as possible in order to maintain a short tolerance chain. Since the position of the antenna relative to the PWB affects the RF performance, all efforts should be made to hold it within tight tolerance. This can be done either by attaching the antenna directly to the PW, or by otherwise keeping a big distance between the feeding pin and the ground pins to improve tolerance. When it is possible to attach the antenna to the frame, it may be possible to position the antenna on the carrier. The internal antenna element should be robustly positioned relative to the plastic and metal elements as well, as all loading affects the antenna tuning and its overall performance, as we mentioned in the previous sections. The material of the antenna carrier should be low loss, as well as mechanically and thermally stable. As mentioned, the bandwidth and efficiency of a PIFA antenna are dependent upon its height and size, with the height playing a major role. Since the height of an antenna is very critical, a robust antenna carrier is needed. Sometimes, removing some ground under a PIFA antenna can generally improve the bandwidth, but this will decrease the front-to-back ratio. When a big portion of the ground plane is removed, we call it a nonground PIFA antenna, as in [59]. The antenna element must keep a good distance from metal objects and loss objects in order to ensure a good performance. Because of the dense integration in the handset, the antenna has to be integrated or be very close to other objects. A low-pass inductor or ferrite might be placed in series between the metal and the PWB to reduce the impacts. The antenna element, being close to the battery, camera, speaker, connectors, will affect tuning and loading effect. For an antenna element, the common ways to make the conductive part are as follows: • •

•

Printed metal on the flex film: This process is very flexible, with good repeatability, it is cheap to produce, but it usually needs extra connectors. Punched metal sheet: This process can offer a very good metal contact, the connectors can be a part of the metal which can make it cheaper, and the mechanical tolerances are larger than with a flex-film solution. Printing metal on the antenna carrier directly: This process can offer a threedimensional robust antenna element, with very good repeatability, but the drawback is a low flexibility (long tooling time), and it needs extra connections.

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Connection is another key issue in the quality of a mobile terminal antenna. If possible the antenna should be connected directly to the PWB (i.e., soldered) to minimize the ohmic losses. Although it usually has less flexibility, it is commonly used for the noncellular bands such as Bluetooth and GPS by using a dielectric loading solution. A surface mountable device (SMD) solution is very attractive for such an antenna. If the connectors are used, the lubricant, the vibration limit, and the contact force have to be well controlled. The pads on the PWB should consist of a metal with thicker gold plating; adding contact lubricants can increase the life of the contact by three to five times or more [75]. As a connector solution, Pogo-pin is a common solution used to connect a flexfilm antenna to the PWB, such as the antenna in Figure 5.15. The connection is generally robust and easy to fit to the antenna element. The drawbacks are the cost and the contacting losses. Another common solution is spring connectors. Leaf spring connectors can either be integrated with the antenna element or mounted on the PWB. Leaf springs that have a minimum bending will allow for a cheaper and robust connection (electrically and mechanically). A long bending arm usually introduces extra distributed capacitance, and the uncertain tolerance will degrade the quality of antenna matching. 5.2.3.8 The Radio Impedance Interface Power amplifiers (PA) are generally calibrated to provide an output power at a reference impedance of 50 ohms. A small antenna in the mobile terminal cannot realize 50 ohms throughout multiband frequency ranges. So the output power from the PA will vary as a function of phase at non-50 ohm antenna impedances. If the phase of the non-50-ohm antenna is properly matched to the PA even when the antenna has a high VSWR, it is still possible to achieve a similar, or in some cases, higher output power than at 50 ohms. This method is through the use of phase shifters or delay lines to change the phase of the antenna to match to the PA impedance instead of using other matching networks, which usually brings in some insertion loss. The advantage can help to improve the performance at band edges where there is a higher VSWR [75]. For the receiving frequency range, the matching can be improved by adding some LC circuitry when the antenna is not 50 ohms. The spurious emission is the major factor degrading the radio receiving performance, as discussed previously in this section. This is primarily an integration problem. The major emission components are LCD, camera (and camera flex), keypad, system connector, and any additional flexes. 5.2.4 The Multichannel Antenna Applications The main mobile business feature changed from voice to data communication modes in the beginning of the twenty-first century. Next generation mobile system and MIMO

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systems will offer much higher data rate via the wireless channels, which need a better channel quality or independent multispace channels. The key devices for transmitting and receiving signals are the antennas. By using the multichannel antenna technique, the antenna diversity feature and MIMO can be realized and the performance of the system can be significantly improved. How to characterize the multichannel antenna performance in an engineering way is very important for the mobile industry and the antenna manufactures [76–79]. In this section, we will discuss some fundamental technologies for this application. To be able to characterize the performance of a multichannel antenna in a mobile environment, some parameters have been defined in the early academic journals [76, 77, 80, 81]. The mean effective gain (MEG) is a statistical measure of antenna gain in the mobile environment. In MEG computation, the incoming wave propagation model is defined to describe a specific mobile environment. The correlation coefficient of the antennas is another important parameter to describe the correlation of the antennas. These parameters can be calculated from the three-dimensional far field complex radiation patterns of the antennas, which can be obtained from a numerical method or an advanced measurement system. Recently, some theoretical work has shown that in some simple cases of the environment, such as a uniform random field case, the correlation coefficient can be calculated from near field parameters such as the S-parameters [82, 83]. This is a very simple way to calculate multichannel antenna correlation evaluations. The random field measurement methods such as the reverberation chamber are important tools to repeat a random field environments [84–86]. Mean gain and diversity gain in a random environment can be tested in this kind of chamber [87–89]. The uniformly distributed random field chamber for engineering applications has been developed in recent years [90]. 5.2.4.1 Different Gain Definitions in a Multichannel Antenna System Mean Gain Mean gain (MG) is the average antenna gain in the whole space and it is equivalent to the antenna efficiency. The maximum value is unity, or 0 dBi. It can be calculated from an integration of the radiation pattern files as in (5.2) or measured from a scattering field chamber as an average value by using a standard antenna as a reference as in (5.3). 2␲ ␲

冕冕

MG =

(G ␪ (␪ , ␾ ) + G ␾ (␪ , ␾ )) (sin ␪ ) d␪ d␾

0 0

4␲ MG =

Pmean standard P mean

⭈ MG standard

=␩

(5.2) (5.3)

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where MG standard is the mean gain of a standard antenna, G ␪ (␪ , ␾ ), G ␾ (␪ , ␾ ) are the antenna power gain patterns. Mean Effective Gain MEG is a statistical measure of the antenna gain in a mobile environment. It is defined by the ratio between the mean received power of the antenna and the total mean incident power when moving the antenna over a random route. It can be expressed by 2␲ ␲

MEG =

冕冕冉

冊

XPR 1 G (␪ , ␾ ) P ␪ (␪ , ␾ ) + G (␪ , ␾ ) P ␾ (␪ , ␾ ) (sin ␪ ) d␪ d␾ 1 + XPR ␪ 1 + XPR ␾

0 0

(5.4) where XPR is the cross-polarization power ratio, which is defined by the mean incident power ratio Pv /Ph where Pv and Ph are the mean incident power ratio of the vertically and horizontally polarized waves, respectively. And P ␪ , P ␾ are the theta and phi components of the normalized angular power density functions of the incoming plane waves. If the antenna efficiency is 100%, and the antenna is located in a uniform random environment (XPD = 1, P ␪ = P ␾ = 1), MEG = 1/2 (i.e., −3 dB). It should be mentioned that the MEG here is for an isolated antenna case and the mutual coupling loss is not included. Diversity Gain The effectiveness of diversity is usually presented in terms of the diversity gain (DG). The diversity gain can be defined as the improvement in time-averaged signal-to-noise ratio (SNR) from combined signals from a diversity antenna system, relative to the SNR from one single antenna in the system, preferably the best one. This definition is conditioned by the probability that the SNR is above a reference level. A detailed description can be found in [91]. It is also called apparent diversity gain in some literature [89]. Effective Diversity Gain and Actual Diversity Gain In the DG definition, one of the branch signals is used as the reference. It is a relative value, and the antenna efficiency is not included. An effective diversity gain (EDG) [89] is defined to include the total antenna efficiency, EDG = DG ⭈ ␩ ant , where ␩ ant is the antenna efficiency of the best antenna, including reflection losses, ohmic losses, and mutual coupling losses; in this case, the diversity gain is relative to an ideal antenna [89, 92].

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The actual diversity gain is defined relative to an existing single antenna with certain efficiency; the mutual coupling loss and loading loss due to the second antenna are included [89]. Diversity Antenna Gain The most effective parameter to evaluate the diversity antenna performance is diversity antenna gain (DAG) since it includes the diversity gain and mean effective gain (channel information). It is defined as DAG = MEG ⭈ DG, and it can considered as the system gain when compared with the reference antenna in the same propagation environment. 5.2.4.2 Calculation and Measurement Methods Mean Gain and Mean Effective Gain In a simulation, a full wave solution by using MoM, FDTD, or FEM is needed. The antenna efficiency includes the ohmic losses, reflection losses, and mutual coupling losses in a multiple antenna system. In a measurement, 3D gain pattern is measured by a 3D test system when the other antenna elements are loaded; MG and MEG can be calculated from (5.2) and (5.4). A propagation model is included in the MEG computation. When a uniform random environment is assumed, MG can also be measured in a reverberation chamber, which represents the MEG when the incoming wave has a uniform distribution. Correlation Coefficient The correlation coefficient determines the quality of multichannels in the diversity and MIMO systems. For a two-antenna system case, it can be expressed from the far field patterns as:

␳e =

|冕冕 2␲ ␲

0 0

冠XPR ⭈ F ␪ 1 (␪ , ␾ ) F ␪*2 (␪ , ␾ ) P ␪ (␪ , ␾ ) + F ␾ 1 (␪ , ␾ ) F␾*2 (␪ , ␾ ) P ␾ (␪ , ␾ )冡 sin (␪ ) d␪ d␾

|冕冕 |冕冕 2␲ ␲

0 0

2␲ ␲

0 0

冠XPR ⭈ | F ␪ 1 (␪ , ␾ ) | 2 P ␪ (␪ , ␾ ) + | F ␾ 1 (␪ , ␾ ) | 2 P ␾ (␪ , ␾ )冡 sin (␪ ) d␪ d␾

| |

|

2

⭈

冠XPR ⭈ | F ␪ 2 (␪ , ␾ ) | 2 P ␪ (␪ , ␾ ) + | F ␾ 2 (␪ , ␾ ) | 2 P ␾ (␪ , ␾ )冡 sin (␪ ) d␪ d␾

(5.5)

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In a uniform random environment, it can be simplified as:

|冕 冕 2␲ ␲

␳e =

0 0

冠F ␪ 1 (␪ , ␾ ) F ␪*2 (␪ , ␾ ) + F ␾ 1 (␪ , ␾ ) F␾*2 (␪ , ␾ )冡 sin (␪ ) d␪ d␾

|

2␲ ␲

冕 冕 冠|

|

F ␪ 1 ( ␪ , ␾ ) | + | F ␾ 1 (␪ , ␾ ) | 2

2

冡 sin (␪ ) d␪ d␾

0 0 2␲ ␲

冕 冕 冠|

F ␪ 2 (␪ , ␾ ) | + | F ␾ 2 (␪ , ␾ ) | 2

2

|

冡 sin (␪ ) d␪ d␾

0 0

|

2

⭈

|

(5.6)

In this case, the correlation coefficient can be expressed with the near field S-parameters [80, 82] as:

␳e =

* S 12 + S 21 * S 22 | 2 | S 11

冠1 − | S 11 | 2 − | S 21 | 2 冡冠1 − | S 22 | 2 − | S 12 | 2 冡

(5.7)

In this way, the correlation coefficient can be calculated or measured quickly. From (5.7), an interesting result can be found: When the antennas have a strong coupling and both antennas are well matched, the correlation can still be very low, DG will be higher; but the coupling loss will be higher, then the effective diversity gain will be low. Diversity Performance Parameters Derived from Pattern Measurements. The diversity performance can also be calculated from the pattern measurement. The diversity gain is defined as the improvement in time-averaged SNR from combined signals from a diversity antenna system, relative to the SNR from one single antenna in the system, preferably the best one. This definition is conditioned by the probability that the SNR is above a reference level. The probability value is optional but is usually set to 50% or 99% reliability [92]. The general mathematical expression for diversity gain is as follows: DG =

冋

␥c ␥1 − ⌫c ⌫1

册

(dB )

(5.8)

P (␥ c < ␥ s /⌫ )

where ␥ c is the instantaneous SNR of the diversity combined signal, ⌫c is the mean SNR of the combined signal, ␥ 1 is the highest SNR of the diversity branch signals, ⌫1 is the mean value of ␥ 1 and ␥ s /⌫ is a threshold or reference level. The above definition of diversity gain is illustrated in Figure 5.26.

250

Figure 5.26 Probability for different number of branches of an M-port antenna system and diversity gain definition for M = 2 [92].

In order to derive the diversity gain of a general two-port antenna system, the probability function for the one-port antenna case is calculated and compared to the twoport antenna case. The difference in probability will define the achieved diversity gain of the system, as shown in Figure 5.26. Assuming uncorrelated signals with a Rayleigh distribution of the EM field at the antenna ports with equal noise, the probability function P for an M-port antenna system can be defined as follows [92]:

冉

冉 冊冊

P (␥ c < ␥ s /⌫) = 1 − exp −

␥s ⌫

M

(5.9)

As shown in Figure 5.26, this definition gives a theoretical diversity gain value of maximum 10 dB for 1% probability and approximately 2.5 dB for 50% probability. Antenna diversity gain as well as diversity system gain can be easily determined from calculations performed on simulated or measured data of the multiantenna system.

251

Multichannel Antenna Simulations The first method of computation is to calculate the S-parameters by using a MoM method, FEM, or FDTD method. The correlation and antenna mutual coupling can then be calculated according the above-mentioned theory. (In this case, a uniform random distribution is assumed.) The second method is to calculate the three-dimensional complex radiation patterns from a simulation, integration of (5.5) to get the correlation, and finally the diversity antenna gain by using the theory described in the last section. In this case different propagation models can be included. Multichannel Antenna Measurements The S-parameters are measured by using a network analyzer. The correlation and coupling can then be calculated by using (5.7), which represents the uniform distribution case. The S-parameter phase calibration has to be done carefully. The cable interference has to be minimized as much as possible. The diversity performance can also be calculated by using the measured threedimensional complex pattern as described in the last section. This technique requires more measurement time. This approach gives a complete repeatable and adequate method for measuring and comparing diversity performance of different devices and is therefore highly suited for the initial development of diversity antennas. To determine the diversity system gain of an antenna it is necessary to process the collected data by evaluation of the correlation including different propagation environments [80, 81]. The antenna performance is strongly affected by the nearby environment. The propagation possibilities in different terrain and surroundings can have both positive and negative effects on the performance. The presence of a user (body, head, and hand) is also a factor that generally makes a difference but is difficult to model in a satisfactory way. The conventional propagation models covering the most common environments for mobile terminals are defined in Table 5.1, to be included in a general antenna diversity performance calculation. The propagation effects are included in the expression defined as the power spectrum of the polarized incoming waves P ␪ (⍀) and P ␸ (⍀), expressed as statistical distributions. A reverberation chamber (scattering field chamber) is also proposed to test diversity performance [88]. It is a method to reproduce a three-dimensional isotropic random field environment. Mean gain and mean diversity gain can be evaluated in this kind of chamber. This is a convenient test indicator to compare multichannel antenna implementations [90]. 5.2.4.3 A Multichannel Antenna Performance in Different Incoming Wave Models Test Examples: Dual Slot Antennas and Dual Polarization Patch Antenna. Two manufactured two-port antennas in the 2-GHz band region are analyzed as an example. The first antenna consists of two slots in a ground plane employing space and polarization diversity

252

Table 5.1 Propagation Models and Scenario Parameters Propagation Model (Elevation/Azimuth) Laplacian/Uniform

Gaussian/Uniform Statistical Distribution −

P ␪ (␪ ) = A ␪ e

冤

−

P ␾ (␪ ) = A ␾ e

冉 冉

冤

␪−

␲ − mv 2

冊冊

2

2␴ v

冉 冉 ␪−

␲ − mh 2 2

2␴ h

2

冥

冊冊

2

冥

−

P ␪ (␪ ) = A ␪ e

冤

−

P ␾ (␪ ) = A ␾ e

P ␾ (␾ ) = 1

P ␾ (␾ ) = 1

P ␪ (␾ ) = 1

P ␪ (␾ ) = 1

冉 冉

√2 ␪ −

冤

␲ − mv 2

冊冊

2

␴v

冉 冉

√2 ␪ −

␲ − mh 2 2

␴h

2

冥

冊冊

2

冥

Elliptical

High directivity, 1/8 of the sphere: P ␪ (␪ ) = √A ␪ a ␪ 0 P ␾ (␪ ) =

√A ␾ (a ␾ 0 − b ␾ | ␾ | )

Low directivity, 1/2 of the sphere: 2

P ␪ (␪ ) =

√A ␪

P ␾ (␪ ) =

√A ␾

s ␪␪ 2

s ␪␪ + sin2 ␪ 2

Scenario Parameters Indoor

XPR = 5 dB m v = 10° m h = 10° ␴ v = 15° ␴ h = 15°

XPR = 5 dB m v = 10° m h = 10° ␴ v = 15° ␴ h = 15°

Outdoor

XPR = 1 dB m v = 20° m h = 20° ␴ v = 30° ␴ h = 30°

XPR = 1 dB m v = 20° m h = 20° ␴ v = 30° ␴ h = 30°

Isotropic

XPR = 1 dB m v = 0° m h = 0° ␴v = ∞ ␴h = ∞

XPR = 1 dB m v = 0° m h = 0° ␴v = ∞ ␴h = ∞

XPR = 5.5 dB s ␪␪ = 0.29 s ␾␪ = 1.06 s ␪␾ = 0.44 s ␾␾ = 1.18 a ␪ 0 = 0.16 a ␾ 0 = 0.7 b ␾ = 0.11

s ␾␪ 2

s ␾␪ + sin2 ␪

253

as shown in Figure 5.27(a). The second one is a dual polarization diversity patch antenna, shown in Figure 5.27(b) [81]. The scattering parameters have been analyzed by a method of moment code, and measured with a network analyzer and at a test range. The mutual coupling between the two ports is below −20 dB across a 200-MHz bandwidth. The correlation coefficient in a three-dimensional isotropic propagation environment based on calculated and measured scattering parameters by using (5.7) is below 0.01. The results agree very well with the far field method weighted with an isotropic environment by using (5.6). The diversity gain and the diversity system gain depend on the propagation environment. A number of propagation reference cases (see Table 5.1) have been analyzed in combination with the radiation pattern [81]. The 1% cumulative probability level based on simulations is presented in Table 5.2 for the switched diversity case. As can be seen, the gain does not vary much with the environment scenarios. The estimated antenna losses are 0.24 dB (95%) and 0.67 dB (86%), for the two-slot antenna and the two-feed patch antenna, respectively. From the results in Table 5.2, it was found that the two-element antennas with very low correlation and high diversity system gain have the diversity system gain of about 9 dB at the 1% probability level. Diversity gain is mainly determined by the signal correlation, signal imbalance, and diversity combining method, while diversity system gain includes the effects of antenna matching, losses, and mutual coupling. It is found that these parameters do not depend strongly on the propagation model used. Thus, the

Figure 5.27 Dual-port antenna prototypes: (a) two-slot antenna and (b) two-feed patch antenna. (Source: [81]. Courtesy of Sony Ericsson.)

254

Table 5.2 Switched Diversity Results at the 1% Level for the Two-Slot Antenna and the Two-Feed Patch Antenna

Propagation Models Gaussian/ Uniform Laplacian/ Uniform Elliptic Uniform

Outdoor Indoor Isotropic Outdoor Indoor Isotropic High directivity Low directivity

Two-Slot Antenna Effective Diversity Gain Diversity Gain (dB) (dB)

Two-Feed Patch Antenna Effective Diversity Gain Diversity Gain (dB) (dB)

9.1 8.6 8.8 9.0 8.5 8.8 7.4 7.2 9.0

9.7 10.0 10.0 9.8 10.1 10.0 8.0 9.6 9.8

8.8 8.3 8.6 8.8 8.2 8.6 7.2 7.0 8.7

9.1 9.3 9.3 9.1 9.4 9.3 7.4 9.0 9.1

isotropic random environment seems to be a good simplified scenario to evaluate diversity performance during the prototype phase. Test methods based on scattering parameters or scattering field chambers may be used to evaluate multichannel antenna performance. A detailed description may be found in [92, 93]. 5.2.4.4 Multiband Antenna Diversity For a multiband radio, multiband diversity and MIMO will be required. In Figure 5.3, a multiband, dual antenna system is shown [94]. The same methods we described in the previous sections are used to characterize the performance. The two antennas used in this study are based on previous antennas presented in [22, 52]. The left antenna (port 1) in Figure 5.28 is a PIFA-based antenna [52]. The long branch arm excites mainly the power for the lower (WCDMA850) band, while the upper branch controls the WCDMA1800 band. A shorted parasitic patch creates the resonance for the UMTS band. The right monopole-based antenna (port 2) in Figure 5.29 has a dense meandering patch on the right side, which controls the WCDMA850 band. The bigger branch on the upper side creates the resonance for the WCDMA1800 band while the smaller, shorted parasitic element creates a capacitive load and tunes the resonance for the UMTS band [30]. The dimensions for the PIFA (antenna 1) are 40 × 18 × 7 mm3 (xyz) (see Figure 5.29 for coordinates) and 40 × 16 × 7 mm3 (xyz) for the monopole (antenna 2). The spacing between the feed points of the two antennas is 84 mm or 0.24␭ (for WCDMA850). The length of the whole multiple antenna structure is 100 mm [94]. The S-parameters simulated in a MoM code and those measured in a network analyzer are given in Figure 5.29. The correlation coefficient can be calculated by using (5.7). The diversity gain and the diversity system gain are calculated from the measured

255

Figure 5.28 A multiband diversity antenna system in a small mobile terminal. The left antenna (port 1) is a PIFA-based antenna. The big patch excites the most power for the lower (WCDMA850 band) while the upper branch controls the WCDMA1800 band. A shorted parasitic patch creates the resonance for the UMTS band. The right monopole-based antenna (port 2) has a dense meandering patch on the right side, which controls the WCDMA850 band. The bigger branch on the upper side facilitates the resonance for the WCDMA1800 band, while the smaller, shorted parasitic element creates a capacitive load and tunes the resonance for the UMTS band. (Source: [94]. Courtesy of Sony Ericsson.)

(by using Satimo system) and calculated (by using MoM) patterns. The calculated and measured diversity gain results are shown in Table 5.3. It is shown in the table that significant diversity gains can be obtained, even at the lowest band where the antenna separation is merely 0.24␭ . In particular, effective diversity gains of up to 7 dB (at the 1% probability level) can be achieved for the WCDMA850 band; it should be mentioned the measured value is better than the simulated value due to the strong interference of feeding cables in the low band in this case. At the higher frequency bands of WCDMA1800 and UMTS the gains are as high as 8 to 9 dB [94]. Due to the limited available space for an antenna in a mobile terminal, the trend of applications is to have the multiband compact antenna to cover several cellular bands plus some noncellular bands. 5.2.4.5 MIMO System Using multiple antenna elements at the transmitter (TX) and receiver (RX) of a communications link in a multipath environment allows the system to have many independent channels in and out. This can result in a marked increase in channel capacity for the same utilization of bandwidth resources and transmitter power [95]. Such a scheme that makes use of multiple antennas at the input and output is called a multiple-input multiple-output, or MIMO, scheme, and is more advanced compared with a conventional diversity system. Consequently, it may therefore dictate the size, number, input-match, polarization, and

Figure 5.29 The amplitude and phase of S11, S21, S22 of the dual antenna system shown in Figure 5.28 by using simulation and prototype measurements: (a) amplitude of S11; (b) amplitude of S21; (c) amplitude of S22; (d) phase of S11; (e) phase of S21; and (f) phase of S22.

256

257

Table 5.3 Diversity Gain and Diversity System Gain at the 1% Probability Level (dB) from Simulations and Measurements Frequency [MHz]

869

881.5

894

1805

1842.5

1880

2110

2140

2170

10.16

10.13

10.11

10.09

10.20

10.20

10.20

10.17

Diversity Gain MoM (simulated) Satimo (measured) MoM (simulated) Satimo (measured)

6.53

7.02

7.48

10.15

10.17

10.08

10.10

9.95 10.20 10.20 Effective Diversity Gain

3.84

3.77

3.76

8.69

9.03

9.25

9.03

8.95

8.88

6.77

7.15

5.94

7.65

7.47

7.06

8.48

7.45

7.96

pattern requirements of the antennas for a given system. Fundamental to a MIMO scheme is the ability to achieve separate decorrelated paths between each of the TX and RX antennas [96, 97]. In order for the paths to be decorrelated, the environment has to be sufficiently rich in scattering so that independent fading will be exhibited in each of the paths between transmit and receive antennas [98, 99]. As we described in the diversity requirement, the MIMO antenna has to be designed with a low correlation [95]. One can use space, radiation pattern, or polarization diversities to make the channel to be decorrelated. Recent research shows that the maximum available capacity of a MIMO system is possibly measured in the simulated repeatable fading environment of a reverberation chamber, as described in [90, 93]. Furthermore, due to the lack of available space on the mobile terminal and the desire to use multiple elements, the antennas will have to be electrically small and may be multiband [95, 100]. The compact diversity antenna, smart beamform, and MIMO scheme might be used in a smart way to meet different enviroments. The research of compact MIMO antenna and MIMO system measurement are ongoing. 5.2.5 Human Body Interaction with Terminal Antennas and Some Measurement Methods The human interaction with an antenna is another important practical issue for mobile terminal antennas. Both scientists and the industry have spent much effort to study the near field, SAR, body loss, impact of antenna and handset types, and position of the antenna. The results can help to improve the communication quality and reduce the negative impact of human body effect [101–104]. In this section, we will introduce briefly the human body effects, and describe of some useful definitions and test methods. The detail of the measurement methods can be found in the literature.

258

5.2.5.1 Human Body Effects on Handset Antenna As mentioned in the previous sections, the radiations from a terminal are from both the antenna element and the terminal chassis. It is very important to measure the final radiation performance with the influences of the human body. The body absorption can be defined as head loss and hand loss for handset talking position. Since the hand effect is difficult to handle and do repeatable measurements, head loss is more often used in engineering tests. A phantom is usually used to do the evaluation test to be equivalent to human body tissue. A typical head phantom is the SAM phantom, which is defined for SAR measurement. Figure 5.30 shows the SAM head phantom that is commonly used in the engineering tests [105]. The body loss is dependent on the antenna type and chassis size and type. This is due to the current distribution on the chassis. The typical current distribution for a PIFA antenna on the bar handset can be found in Figure 5.3. Body Loss of Bar Handsets As mentioned in Section 5.2.1, for the bar type handset with a monopole antenna, a half wave monopole has less current on the chassis; therefore it has less body loss. A shorter monopole antenna has strong induced currents on the chassis; the head loss will be higher. With a PIFA antenna, there is less current on the front surface. Therefore it creates a

Figure 5.30 The SAM head phantom for radiation performance test. Constructed from reinforced fiber glass resin using inner and outer moulds produced from CAD files; biologic tissue-simulated liquid is filled in. (Source: [105].)

259

minor head loss, especially on frequencies above 1.8 GHz. The typical head loss (without hand loss) for PIFA antenna on a bar type of phone is about 5 to 7 dB on the 800- to 900-MHz band, and about 2 to 4 dB in the frequency range 1.8 to 2.1 GHz depending on the size of phone chassis. The hand effect can differ quite a lot depending on the user’s holding position. The low band has typically 2 to 5 dB extra loss, and the high band has 1 to 5 dB extra loss if the antenna is covered by the hand. For the case of a bar handset (typical 100 mm long) with a monopole antenna at the bottom, the typical head loss is 6 to 8 dB in the low bands, and 3 to 5 dB in the high bands. Body Loss of Clamshell Handsets For a foldable type of phone (clamshell phone), a PIFA antenna can be placed in the middle of the phone (near the hinge) or at the bottom. The clamshell phone in the open mode has quite a long chassis. An antenna located in the bottom of the clamshell phone usually has less head loss compare with hinge antenna positions. The hand effects are usually large for a hinge antenna in a clamshell phone. The body loss can also be simulated by using a numerical tool. The common methods are FDTD, FEM, or MoM. The time domain method is more effective for antenna body simulation. 5.2.5.2 SAR, Measurements, and Simulations Specific absorption rate is a value that measures how much power is absorbed in biological tissue when the body is exposed to electromagnetic radiation. The units are watts per kilogram of tissue elements. The maximum SAR is specified as applying to any 1g and 10g tissue elements. Governments around the world have agreed to define guidelines concerning SAR limits and in Europe’s case the International Commission on NonIonizing Radiation Protection (ICNIRP) organization is responsible for this. The SAR limit is set to be 2.0 W/kg over a 10g cube in Europe [105]. In the United States the limit is 1.6 W/kg over a 1g cube according to Federal Communication Commission (FCC). Spatial-peak SAR is defined as the maximum average SAR of a 10g or a 1g cubic volume of tissue. SAR is determined as follows: SAR =

␴ | E | 2 [W/kg] ␳

(5.10)

where E is the electric field (V/m), ␴ is the conductivity (S/m), and ␳ is the density (kg/m3).

260

SAR is measured on a complete head or flat phantom placed next to the mobile phone and the highest value detected decides how large the SAR value is. The measurements are made on a biologic-simulating liquid, which has a relative permittivity and a conductivity, which depends on the frequency. For GSM standard in Europe, SAR is measured for 900 MHz and 1,800 MHz. The values used for 900 MHz are ⑀ r = 42.5 and ␴ = 0.85 S/m and for 1,800 MHz ⑀ r = 41 and ␴ = 1.65 S/m [105]. The input average power of the antenna is different for different standards. For GSM it is 250 mW and for PCS and DCS it is 125 mW, which has to be taken into consideration when evaluating the results. The CDMA systems have similar power level. The DASY system shown in Figure 5.31 is a commonly used system for SAR measurements [107]. The tissue liquid is filled in the phantom; the probe tests the SAR distributions in a twin phantom. It is observed that exposure to magnetic fields (electric currents) rather than electric fields make a high value of SAR. The high electric RF current near the body usually results higher SAR in the body. So the antenna type, antenna position, phone style, material loading, and metal grounding will influence the SAR values. SAR can also be simulated by using numerical methods such as FDTD or FEM methods. In Figure 5.32 a handset

Figure 5.31 A DASY system for SAR measurement. (Source: [107].)

261

Figure 5.32 An example of SAR and body loss simulation by using FDTD method. (Source: [106].)

is in a touching position to a SAM phantom, and the SAR distribution is simulated by using FDTD method [106].

5.2.5.3 Hearing Aid Compatibility Mobile phones interfere with hearing aids both by their RF emission and their EM field. Modern hearing devices have an audio amplifier, which makes it easy for hearing aid users to use an ordinary phone. If the hearing aid is in a high level pulsed EM signal such as the talking position of mobile phone, the amplifier will generate some unpleasant buzzing noise. The U.S. FCC stated on July 10, 2003, that by September 2005, in the United States, all mobile phone manufacturers must offer at least two hearing aid compatibility (HAC)-approved GSM models [21]. By February 18, 2008, half of the mobile phone models sold in the U.S. market must be HAC approved. The manufacturer may specify how to use the phone to make it work according to the HAC standards, which means that if the phone can be used in different ways, such as a clamshell phone,

262

it is sufficient that it is HAC approved in one specific position [21]. If a mobile phone is used with accessories, such as headsets, it is not counted as a HAC phone. To measure HAC a 5 × 5-cm2 grid is centered over the acoustic output of the phone. The grid is divided into nine equal parts and it is scanned with a resolution of 2.5 mm, according to FCC’s requirements. The three contiguous parts of the grid with the highest values, which cannot include the center part of the grid, are removed from the results. The remaining results are used to calculate the highest E- and H-fields. HAC is defined by and measured as the highest E-field and the highest H-field detected in the near field area. The measurement probe is placed about 15 mm from the acoustic output of the mobile phone. The E-field and H-field are measured in decibels and since GSM has a modulation scheme that lies in the audio spectrum, 5 dB has to be added to the HAC value for GSM for comparison with the other frequency bands. The details of the measurement method can be found at the FCC homepage, since the standard is updated on an ongoing basis [21]. 5.2.5.4 Test Methods for Mobile Terminals The test method of a mobile terminal antenna is an important area. The conventional 3D pattern measurement has to be used to analyze the radiation pattern shape, field polarization, and average and peak gain. As the mobile terminals are often used randomly in scattered field environments, simple two-dimensional radiation pattern measurements are not good enough to evaluate antenna or system performance. The radiation efficiency or mean gain, which was defined in (5.2) and (5.3) in Section 5.2.4.1, is a more useful figure. Total radiated power (TRP) and total isotropic sensitivity (TIS) are RF system performance measurements in the active phone cases. Methods to measure the antenna performance and RF system performance in an efficient way are strongly required in the industry. Recently, many 3D pattern integration measurement systems, scattering field test systems, and Wheeler cap methods were proposed to improve the testing speed [84, 87, 108, 109]. The new methods for channel modeling and multiantenna system test were also proposed [80, 93]. 3D Pattern Integration Measurements As we described in Section 5.2.4.1, the radiation efficiency can be calculated from (5.2). When the three-dimensional radiation pattern is measured and compared with a reference antenna, the antenna radiation efficiency can be calculated from (5.3). If the measurement facility is calibrated, the total radiated power can be calculated by integration of the 3D radiation pattern. By following the mathematical definition of the radiation efficiency, the measurement result may be made very precisely. The accuracy is dependent on the resolution of the measuring angle increments. The details can be found in [84]. The three-dimensional

263

pattern measurement setup can be either a mechanical driving system or a near field spherical scanning system to scan a three-dimensional near field. With a fast Fourier transform (FFT) the far field pattern can be derived from the near field measurement. The Stargate system from SATIMO [110] is an example of such a system, as shown in Figure 5.33. The Random Field Measurement Method The random field measurement method is also called the scattered field test method. The measurement is usually performed in a well-defined scattered field environment. The reverberation chamber is a typical random field measurement facility [84]. A reverberation chamber is a metal cavity which is large enough to support several cavity modes at the operating frequency range. The modes can be stirred by mechanical stirrers or by mobile sampling techniques to create a Rayleigh distributed transfer function between two antennas inside the chamber. The average of the test power level is calculated by measuring the transfer function with a large number of different stirrer positions and a large number

Figure 5.33 The Stargate spherical near field scanning system is commonly used for handset radiation performance measurement. (Source: [110].)

264

of sample positions. A data sample or several samples are measured at different positions in the chamber for each stirrer position. The averaging of these levels is made from these large numbers of independent samples. Reverberation chambers with high Q have been usually used for EMC measurements. The Rayleigh statistics are also present in chambers with a low Q-value (down to 100). This makes them ideal to use for testing of mobile terminals designed for use in fading environments. The terminal antenna under test is compared with a standard antenna such as a dipole antenna in the same scattering environment. The relative gain can thus be calculated by using (5.3). Recently the application of reverberation chambers has been extended to obtain accurate measurements of small terminal antennas and active terminals in a Rayleigh fading environment. To get a good accuracy, hybrid mode stirring techniques are sometimes applied to get more independent samples during the same measurement time [87]. Different sampling techniques such as selective sampling and multisampling are used to improve the independent sample rate [111]. The new measurement applications of this kind of method include radiation efficiency of small antennas, diversity gains of multiantenna system, maximum available capacity of multiport antennas for MIMO systems, radiated power of active terminals, diversity gain of active terminals, and receiver sensitivity. All the measurements can be done with the terminal located in a position inside the chamber corresponding to free space and in talk positions near a head phantom or hand phantom [87, 90]. An example of a reverberation chamber which was developed in Sony Ericsson is shown in Figure 5.34. The mechanical stirring, platform stirring, and multiprobe sampling techniques [111] are used in the measurement in order to enhance the sampling speed of independent sample. The Wheeler Cap Method This method was proposed by H. A. Wheeler to measure the ohmic losses of small antennas by covering the antenna with a small metallic cap [112]. The measured resistance of the antenna covered by the cap is equivalent to the losses of the antenna, while the real part of the measured input impedance of the antenna in free space includes the radiation and losses. Thus, using these two measured results, the radiation efficiency of the antenna can be calculated by using the radiation resistance divided by the real part of the measured input impedance of the antenna without the cap. The accuracy of the method is dependent on the frequency. Avoiding a resonance between the antenna and the cap is important to have a good measurement [84]. A multisample test in a Wheeler cap can provide a good accuracy [109]. Figure 5.35 shows an example of a practical Wheeler cap [109]. System Performance: TRP and TIS Cellular Telecommunications & Internet Association (CTIA) is a U.S.-based international organization that serves the interests of the wireless industry by lobbying government

265

Figure 5.34 A reverberation chamber or mode stirring chamber for mobile terminal antenna measurement. (Courtesy of Sony Ericsson.)

agencies and assisting with regulations [113]. CTIA has established a certification program for mobile phones, which includes radiated performance testing. A working group, including operators, mobile phone manufacturers, and test equipment vendors, is evolving a detailed test plan. The most recent release of this is in [114]. According to the test plan, the total radiated power and total isotropic sensitivity are obtained by full sphere radiated measurements in an anechoic chamber. The test setup includes a base station emulator, which is used to establish a call to the mobile phone inside the anechoic environment. For transmit, the mobile’s effective isotropic radiated power (EIRP) can be recorded as a function of the direction of radiation using a narrow band power measurement device. For receiving, the base station emulator is used to record the receiver sensitivity as a function of angle of arrival. Integration of the EIRP and the sensitivity over the full sphere yields the TRP and TIS, respectively.

266

Figure 5.35 A practical Wheeler-cap test setup. (Source: [109].)

Total Radiated Power. ‘‘TRP is a measurement of the device’s transmitter performance. This procedure records EIRP every 15° for the total of 528 points covering the whole 3D space surrounding. As a result, a 3D pattern is obtained showing maximum and minimum points of transmitter performance. This gives much more information than peak EIRP (a single figure) alone’’ [115]. The value can be measured by the pattern integration method. It can also be measured in a scattering field chamber by using reference power level. As mentioned in Section 5.2, the reference level can be calibrated by a standard antenna with a standard power source. Total Isotropic Sensitivity. ‘‘TIS is a measure of the device’s receiver performance. This measurement records how much the base station’s transmitted signal can be reduced before mobile station reaches a BER (Bit Error Rate) of 2.44%. The power transmitted by the base station is recorded over 253 angle points. This measurement is crucial in understanding connection quality in a mobile network’’ [115]. Since the test values have to be measured in each point, one by one, TIS measurement is usually much more time consuming than a TRP measurement. A summary of some of the measurement parameters is shown in Table 5.4 [115]. 5.3 DESIGN AND PRACTICE OF ANTENNAS FOR HANDSETS The Personal Digital Cellular (PDC) system, which is the second generation mobile phone systems in Japan, started services in 1993. This system operates in two frequency bands:

267

Table 5.4 Measurement Parameters According to CTIA Standard Measurement Parameters Angular Sample Spacing Mobile Station Transmit

EIRP, TRP, NHRP

15° in ␪ and ␾

Mobile Station Receive

Sensitivity, TIS

30° in ␪ and ␾

Test Configurations Free space, phantom head (left + right ear) Free space, phantom head (left + right ear)

one in the lower bands of 810 to 885 MHz (except 828 to 870 MHz and 925 to 958 MHz, which are assigned for receiving and transmitting of mobile terminals, respectively), and another in the higher bands of 1,477 to 1,501 MHz and 1,429 to 1,453 MHz, which are assigned for receiving and transmitting of MTs, respectively. When the PDC services started, the antenna used in handsets was mainly a whip, which is a monopole, with a normal mode helical antenna (NMHA) placed on the top of the whip as shown in Figure 5.36. No electrical connection exists between the whip and the NMHA. When the whip is extracted outside the handset case, only the whip acts as an antenna by being fed at the bottom as shown in Figure 5.36(a). When the whip is retracted inside the handset case, only the NMHA acts as an antenna, to which the RF unit is connected as shown in Figure 5.36(b). In practice, the typical length of the monopole is either 3/8␭ or 5/8␭ , because with these lengths, relatively little current flows on the handset unit and appropriate impedance match to the load impedance is achieved as compared with other lengths. This is discussed in [116–118]. Current design of PDC MTs incorporate a diversity antenna system, to which a built-in PIFA has been used as the subantenna of the diversity system as shown in Figure 5.37. Other small planar antennas (e.g., a modified PIFA and other small antennas like a chip antenna) have also been used as the subantenna of the diversity system. Some of the modified PIFA structures are shown in Figure 5.38. A recent trend in MTs is to use a built-in antenna as the main antenna and consequently the second antennas are required to be smaller than the conventional ones. Thus, very small chip antennas have been adopted in many MTs as one of the typical second antennas [118]. In this section, recent technologies applied to antennas used in mobile terminals of PDC, 3G systems, and other mobile wireless systems (including WLAN) will be discussed. These are listed as follows: • • • • • •

Multiband and broad band; Novel diversity systems; Mitigation of human body effect; Reduction of SAR values; Omission of a balanced-unbalanced transformer (balun); Downsizing PIFA.

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Figure 5.36 Whip antenna: (a) extracted from the handset unit; and (b) retracted in the handset unit.

5.3.1 Multiband and Broad Band Antenna Technologies 5.3.1.1 Multiband Technique PDC mobile terminals use an antenna system of selection diversity for reception. The frequency band for the reception of MTs is from 810 to 885 MHz and its relative bandwidth is 8.9%. A single subantenna of a diversity system cannot cover this wide bandwidth, as the dimension of the antenna is downsized and the bandwidth is very narrow. However, the necessary bandwidth per link in the practical operation is less than 2%. Meanwhile, the terrestrial digital television (DTV) broadcasting service for mobile wireless terminals started in Japan in April 2006. That system uses the UHF band, which ranges from 470 to 770 MHz, and the relative bandwidth is 48.4%. The bandwidth of each channel is 6 MHz, which is further divided into 13 segments, with 12 of the 13 segments allocated for the fixed TV broadcasting and the remaining one allocated for

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Figure 5.37 Antenna elements used in a PDC handset.

Figure 5.38 (a–c) Small sizing of planar element.

broadcasting to mobile terminals. The latter is referred to as the one-seg broadcasting. The relative bandwidth necessary for receiving one channel of the one-seg broadcasting is only 0.1%. These two antennas (i.e., a subantenna for the diversity system and an antenna for the one-seg broadcasting) have common features as follows: • •

The bandwidth necessary for the practical operation is less than 2%, although the entire bandwidth of the system is 8.9% or wider. The dimensions of the antenna element are very small as compared to the wavelength.

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When an MT which contains a narrowband antenna is to be used in areas of wideband services, the MT will adopt a tunable antenna, which has a variable matching circuit (VMC) that varies impedance to match the load over the necessary frequency bands. Figure 5.39 shows an example of a VMC, which uses a variable capacitor diode (varactor) to control the frequency characteristics. This tunable antenna concept has been applied to built-in subantennas of PDC handsets. The circuit components such as the variable series capacitance, C m , and the shunt inductance, L m , play an important roll to achieve good impedance-matching to the antenna load. Other circuit components, C 1 , C 2 , Z 1 , and Z 2 , constitute the bias circuit, through which the voltage, Vcont , is supplied to the varactor, C m . The capacitor C 2 has low enough impedance at the RF, thereby providing the feed path to the antenna, while interrupting the dc bias current to prevent its flow into the feed circuit. The impedances, Z 1 and Z 2 , serve as the bias circuit, by which the appropriate bias voltage is supplied to the varactor. An example of a small antenna, to which a VMC is applied, is shown in Figure 5.40. This antenna is developed for receiving the one-seg broadcasting [119], and has dimensions of 27 × 5 × 5 mm3. Although the VMC cannot be seen in the figure, because it is located beneath a black sheet, a similar VMC to that shown in Figure 5.41 would be used. Figure 5.41 shows the return loss | S 11 | characteristics at the antenna input terminal by taking the voltage V, supplied to the VMC, as the parameter. Return loss lower than −10 dB is observed over the frequency range of 460 to 765 MHz for the voltage V from 0V to 5V. 5.3.1.2 Broad Band Techniques Folded Loop Antenna This antenna has a bent-folded dipole structure as shown in Figure 5.42(a). Originally it was aimed to construct an antenna of balanced structure in order to reduce the current

Figure 5.39 A circuit example of a tunable type of multiband antenna.

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Figure 5.40 The antenna element with variable matching circuit (VMC) for terrestrial DTV, fabricated by Murata Manufacturing Co., Ltd. The dimensions are 27 × 5 × 5 mm3.

Figure 5.41 The relationship between the voltage V impressed to the VMC and the return loss versus frequency at the antenna input terminal.

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Figure 5.42 (a, b) A folded loop antenna structure. (From: [121].  2003 IEICE. Reprinted with permission.)

distribution on the ground plane so that the degradation of antenna performance due to the human body effect would be reduced. However, as the folded dipole antenna has inherently wide bandwidth, attention was paid to an analysis of bandwidth characteristics when the antenna is placed on the ground plane as shown in Figure 5.42(b). The antenna impedance characteristics were analyzed with respect to the dimensions of the antenna, such as s, d, h, w 1 , and w 2 , which are denoted in Figure 5.43 [120, 121]. It has been clarified that the parameters s and the ratio of w 1 to w 2 , w 1 /w 2 , are very important factors to obtain wide band characteristics as shown in Figures 5.43 and 5.44, where (a) shows impedance on the Smith chart and (b) shows VSWR characteristics. It can be seen in the figure that the relative bandwidth of about 43% for VSWR ≤ 2 has been achieved. Use of the balanced structure has some disadvantage, because a balun is necessary, as the folded dipole is a balanced antenna system and a balun is used to connect to the RF front

Figure 5.43 Relationship between the impedance and the spacing s. (From: [121].  2003 IEICE. Reprinted with permission.)

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Figure 5.44 (a, b) Input impedance characteristics versus width ratio (w 1 /w 2 ). Other parameters are chosen as s = 20 mm, h = 8 mm, d = 1 mm, w 1 = w 3 = 1 mm, and w 2 = 4 mm. (From: [121].  2003 IEICE. Reprinted with permission.)

end circuit, which is generally an unbalanced system. Fortunately, a folded-dipole antenna having a ␭ /4 element shows a self-balanced nature, so that a balun is not really necessary. However, there are some differences in the impedance characteristics of these two types of antennas; the comparison of the balanced feed system and the unbalanced feed system in terms of VSWR is shown in Figure 5.45, where (a) show the balanced system and (b) shows the unbalanced system, from which the significant difference can be seen. Wideband Linear Inverted-F Antenna The linear inverted-F antenna has inherently narrow bandwidth. A method to increase the impedance bandwidth of this type of antenna [122] is described here. An example of the antenna is shown in Figure 5.46. The original antenna is a linear inverted-F antenna with a bent horizontal element and a wire element (element #2) is

Figure 5.45 (a, b) VSWR characteristics for optimized physical dimensions. (From: [121].  2003 IEICE. Reprinted with permission.)

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Figure 5.46 (a) Linear inverted-F antenna with an additional element, and (b) its resonance modes. (From: [122].  2003 IEICE. Reprinted with permission.)

added along with the horizontal element as shown in Figure 5.46(a). The bandwidth enhancement is achieved by the addition of the element #2, as a result of parallel resonance attained by this antenna structure as shown in Figure 5.46(b). The increase of the bandwidth attributes to the parallel resonance frequency approaching the serial resonance frequency of the fundamental mode of the inverted-F antenna. Figure 5.47 illustrates the frequency characteristics of input impedance and mismatching loss for cases where the antenna is constituted with and without the element #2. It can also be seen in the figure that the bandwidth enhancement by the addition of element #2 is verified by the calculation using MoM. Figure 5.48 shows the experimental results of the averaged actual gain in Y-Z plane and the mismatch loss, at the optimized physical dimensions shown in the figure. The dark lines show characteristics of the proposed antenna, and the dotted lines show that of the conventional antenna. The bandwidth in terms of actual gain of −1 dBd or higher obtained by the proposed method is 1.7 times wider than the original inverted-F antenna. 5.3.2 Diversity Antenna Technologies This section describes some novel technologies for implementing diversity function to mobile terminals.

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Figure 5.47 Comparison of input impedances. (From: [122].  2003 IEICE. Reprinted with permission.)

5.3.2.1 Diversity Antenna Using Variable Impedance Conventionally, PDC MTs have two antenna elements to function in diversity reception. By switching these two antenna elements, two different radiation patterns are produced and the diversity function is achieved. A unique method was introduced in 2002, in which

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Figure 5.48 Measured frequency characteristics of the averaged actual gain in Y-Z plane and the mismatching loss. (From: [122].  2003 IEICE. Reprinted with permission.)

only one antenna element was used to obtain two different radiation patterns [123]. Figures 5.49 and 5.50 show this concept, which was applied to a clamshell type MT having two separate ground planes; one is for the circuit board and another for the accessories. A radiating element (a built-in antenna) is placed on top of the lower ground plane. An RF line having a variable impedance circuit, which consists of two reactance components, Z 1 and Z 2 , is used to connect the lower ground plane with the upper ground plane. This RF line is placed inside the hinge of the clamshell housing. By switching Z 1 and Z 2 , the impedance of the RF line is varied and the current distributions on the upper and lower ground planes are changed. Consequently, the different radiation patterns depending on the impedance of the RF line are produced. Figure 5.51 shows the radiation pattern when the argument of Z 1 = 0°; that is, the RF line is open. This radiation pattern is similar to that of a dipole, because the current distribution on the ground plane is primarily in the z direction. Figure 5.52 shows the radiation patterns when the argument of Z 2 = 315°.

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Figure 5.49 Diversity antenna using two ground planes. (From: [123].  2002 IEICE. Reprinted with permission.)

Figure 5.50 Control of the impedance Z. (From: [123].  2002 IEICE. Reprinted with permission.)

The radiation patterns in this case are different from that shown in Figure 5.51, because the current distributions increase in the direction other than z. The correlation coefficient [defined by (5.5)] between the two radiation patterns shown in Figure 5.51 and Figure 5.52 is evaluated to be as low as 0.1. By this means, there is no need to consider antenna spacing, because only one antenna element is used to achieve the diversity function. This method has been applied to PDC MTs.

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Figure 5.51 (a, b) Radiation pattern of the diversity antenna in Arg (Z ) = 0°. (From: [123].  2002 IEICE. Reprinted with permission.)

The diversity system introduced in the explanation above can be improved by using an antenna structure shown in Figure 5.53 [124]. A previous model has been modified to have a variable impedance, consisting of a reactance switching circuit, which connects the GND 1 with the radiating element placed on top of the GND 1. With this structure two different radiation patterns are produced as the current distributions on the ground planes are changed, depending on the reactance that is varied by switching the two different reactances, Z 1 and Z 2 . Figure 5.54 shows both the calculated and measured radiation patterns when the argument of Z 1 = 0° corresponding to the circuit open, whereas Figure 5.55 shows those corresponding patterns when the argument of Z 2 = 60°. These are patterns when the clamshell chassis is opened. This method also obtains diversity effect

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Figure 5.52 (a, b) Radiation pattern of the diversity antenna in Arg (Z ) = 315°. (From: [123].  2002 IEICE. Reprinted with permission.)

because the radiation pattern changes drastically by switching the reactance. In the Z-X plane, the main polarization changes, as E ␾ patterns exceed E ␪ patterns. Figure 5.56 shows correlation coefficients with respect to the arguments of the reactance Z 2 , for cases when the clamshell chassis is either opened or closed, respectively. The correlation coefficient is observed to be less than 0.5 for the argument of Z 2 between 60° and 90°, with the clamshell chassis not only being opened but also when closed. Then, by this diversity system, good diversity performance is achieved with the clamshell chassis either opened or closed. Figure 5.57 shows the relationship between the crosspolarization power ratio (XPR) of arrival waves and the pattern-averaged gain (PAG) in the horizontal plane, for the handset at both talking position and browsing situation. The XPR represents the mean XPR of arrival waves, which is defined as the ratio of the

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Figure 5.53 One element diversity antenna: (a) chassis-close; and (b, c) chassis-open. (From: [124].  2005 IEICE. Reprinted with permission.)

mean power of the vertical polarization component to that of the horizontal polarization component. The results shown in Figure 5.57 are obtained by the calculation from measured radiation patterns when a handset user operates a model as shown in Figure 5.81. In the talking situation, the PAG was higher than −8.5 dBi for the arbitrary polarizations of the arrival-waves when the argument of Z is set to 0°. Also, the PAG was higher than 7.0 dBi in the browsing situation when the argument of Z is set to 60°. From these results, this diversity system can be said to be very effective in improving the receiving performance of MT without regarding polarizations of the arrival waves. Comparing this proposed system with the previous system [123], this system is seen to have less mechanical complexity, because no RF line with variable impedance needs to be placed in the hinge of the clamshell unit. This system is further advantageous, since it may be applied to not only the clamshell type MT, but also a mono-block type MT, because no connection is necessary between the GND 1 and GND 2, which does not exist

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Figure 5.54 Radiation pattern in Arg (Z ) = 0°. (From: [124].  2005 IEICE. Reprinted with permission.)

Figure 5.55 Radiation pattern in Arg (Z ) = 60°. (From: [124].  2005 IEICE. Reprinted with permission.)

in the mono-block type MT. This type of diversity system has already been adapted to PDC MTs. 5.3.2.2 Polarization Diversity Antenna This system has been proposed as a potentially useful one to be applied to MTs in either the WCDMA or the MIMO system, to which the polarization diversity algorithm is applied [125]. Figure 5.58 shows a model of this system, where two antenna elements (A and B) are placed on top of the ground plane (conducting plate), and fed through a 180° hybrid— a rat race circuit. Figure 5.59 explains the concept of this system. When port 1 of the

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Figure 5.56 Radiation pattern correlation coefficient versus Arg (Z ). (From: [125].  2006 IEICE. Reprinted with permission.)

Figure 5.57 Relationship between the PAG in a horizontal plane and the XPR of arrival wave in that talking and browsing situation. (From: [125].  2006 IEICE. Reprinted with permission.)

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Figure 5.58 Polarization diversity antenna. (From: [125].  2006 IEICE. Reprinted with permission.)

Figure 5.59 (a, b) Current action in each mode. (From: [125].  2006 IEICE. Reprinted with permission.)

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hybrid is fed, the two antenna elements are excited in phase, and the radiation pattern produced has V (vertical) polarization shown in Figure 5.60, because the current distribution on the ground plane is primarily in the z direction, as shown in Figure 5.59(a). On the other hand, when the two antenna elements are excited out of phase through port 2 of the hybrid, the radiation pattern in the y-z plane has H (horizontal) polarization shown in Figure 5.61, because the x component of current is dominant on the ground plane as shown in Figure 5.59(b). The correlation coefficient of this system was measured to be as low or lower than 0.27. By synthesizing the output signals of port 1 and of port 2 properly, this antenna system can have low polarization loss in whatever position the MT is held. This antenna system has the possibility of being applied to MIMO using dual polarization, because it produces two radiation patterns with different polarizations.

Figure 5.60 Radiation pattern in in-phase mode. (From: [125].  2006 IEICE. Reprinted with permission.)

Figure 5.61 Radiation pattern in out-of-phase mode. (From: [125].  2006 IEICE. Reprinted with permission.)

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5.3.2.3 Transmission Diversity Antenna When an operator’s hands touch a handset and affect the antenna performance, its degradation can sometimes become serious. If a PDC MT has only a single antenna as usual, it cannot deal with deterioration of transmitting performance, although it can improve the receiving performance by means of the single antenna diversity scheme, which is described in the previous sections. Proposed here is a unique technique of transmitting diversity, where two antennas of the MT are used for transmitting as well as receiving [126]. By this method, deterioration of the transmitter performance caused by the operator’s body effect is improved, and at the same time better receiving performance is expected by the receiving diversity scheme. However, the system is not effective against fading, because the frequency band for transmitting differs from that of receiving in PDC systems. A criterion for the improvement of the transmitting performance by this antenna switching system is discussed in [126]. The detail is described in Section 5.3.3. 5.3.2.4 Small Chip Antenna A PDC MT incorporates two antennas: a main antenna and a second antenna, which is a subantenna for diversity system. Recently, MTs tend to install a built-in main antenna, as the MT is tending to be downsized. Typical antennas are an inverted-F antenna, a PIFA, a modified PIFA, and a modified monopole antenna. As the antenna is required to be smaller, the second antenna is also required to be smaller, yet smaller than the main antenna. Accordingly, a small chip antenna has been adopted as the second antenna in many PDC MTs. The chip antenna is a small bulk antenna, which is composed of an antenna element encapsulated by either ceramic material or composite material of ceramic and resin having high relative permittivity. Generally, the antenna element is printed on the material substrate and encapsulated by the dielectric materials having higher relative permittivity greater than 5. The typical printed patterns of the antenna element are meander line, inverted-F, NMHA, and other planar patterns. The whole size of the small chip antenna is usually about a few millimeters. Figure 5.62 illustrates an example of a chip antenna mounted on a ground plane and the antenna pattern. This antenna is developed for small mobile terminals of Bluetooth and WLAN in 2.4-GHz bands, and the dimensions are 3 × 9.8 × 4 mm3 [127]. This type of antenna is usually laid on the ground plane and hence is called ‘‘on ground’’ type. The radiation efficiency is −1.2 to −0.3 dB and the bandwidth for VSWR ≤ 3 is 65 to 124 MHz, depending on the placement of antenna on the upper part of a 40 × 100 mm2 printed wiring board (PWB). Figure 5.63 shows another type of chip antenna placed on the ground plane, but a small part of the ground plane beneath the antenna element is removed. Hence, this type of antenna is called ‘‘no ground’’ type. In the figure, the antenna pattern is also depicted.

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Figure 5.62 Chip dielectric antenna (on-ground type). (Courtesy of Murata Mfg. Co., Japan.)

This antenna can be used similarly for Bluetooth and Wireless LAN in 2.4-GHz bands [127]. The dimensions are 3 × 9.8 × 4 mm3, which are smaller than the previous ‘‘on ground’’ type antenna. This ‘‘no ground’’ type antenna can be installed in any place on the PWB with any orientations, and the radiation efficiency is 3.2 to 0.7 dB and the bandwidth is 84 to 132 MHz for VSWR ≤ 3. Figure 5.64 expresses antenna positions with the indexes of 1 to 6 on the ground plane, and gives the bandwidth and the efficiency corresponding to the antenna position and direction. These chip antennas actually excite the ground plane and induce some currents on the ground plane so that the ground plane acts as a radiator. Thus, the antenna element and the ground plane are combined to constitute an antenna system. Accordingly, the antenna performance of an MT is determined by such factors as the size of the ground plane, position and direction of the antenna on the PWB, and the other materials existing on the PWB. These two types of chip antennas mentioned above can accommodate themselves to these environmental factors by means of tuning with a line on the PWB and a matching circuit. Figure 5.65 shows variation of frequency tuning of the ‘‘no

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Figure 5.63 Chip dielectric antenna (no ground type). (Courtesy of Murata Mfg. Co., Japan.)

ground’’ type antenna shown in Figure 5.63. The length L of a bent line, which connects the ground to the antenna element, is a factor to change the frequency and tune to the resonance frequency. Meanwhile, the fine tuning to the resonance frequency is done by adjusting a reactance, which is connected to the bent line in parallel. The impedance matching can be made by the matching circuit connected to the antenna terminals. The resonance frequency is changed from 2,140 to 2,630 MHz by way of rough tuning first and then fine tuning. The frequency variations attained by either rough tuning or fine tuning are shown in Figure 5.65. Figure 5.66 shows radiation patterns of a ‘‘no ground’’ type chip antenna placed at the position 3 (ANT3) shown in Figure 5.64. This sort of tuning facility gives inestimable flexibilities in designing chip antennas to be used in the MT. 5.3.3 Antenna Technologies Mitigating Human Body Effect It is well known that the actual gain of an antenna mounted on a mobile handset is strongly affected by a human body, especially an operator’s hand and head. The degradations

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Figure 5.64 BW and efficiency at each antenna position (no ground type). (Courtesy of Murata Mfg. Co., Japan.)

caused by the proximity effects are roughly classified into two factors: one is the increase of the impedance mismatching loss and another is the increase of the power absorption by the human body. The antenna impedance varies when the human body approaches the antenna and differs much from that in free space. In the following, three methods to improve the above problems will be described. 5.3.3.1 Adaptively Tunable Antenna An antenna system here adopts an adaptive matching scheme, by which impedance mismatch loss due to the human body effect can be reduced [128]. Although the study of this system is carried out in VHF (150 MHz) band, this system can generally be applied to conventional cellular handsets. Figure 5.67 illustrates the circuit to perform the impedance matching algorithm, in which a normal mode helical dipole antenna and a circuit consisting of four varactor diodes and their bias circuits are used. The antenna is fed through the variable impedance circuit, in which two varactor diodes C s’s are used in series and two other C p’s in parallel. A balun having the impedance transforming ratio of 1:4 is used at the input of the antenna circuit. The equivalent circuit, including the varactor diodes and a detector circuit, is shown in Figure 5.68(a). The n-type GaAs varactor diode is used, whose capacitance can be

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Figure 5.65 An example of resonance frequency tuning of the chip dielectric antenna (no ground type). (Courtesy of Murata Mfg. Co., Japan.)

varied in the range from 1 to 4.5 pF for the supply voltage of 0V to 10V. The impedance matching concept can be understood by using the impedance chart shown in Figure 5.68(b), where D denotes the distance between the antenna and the human body. The values of C p and C s , which are varied by the control-voltage V 1 and V 2 , respectively, are controlled by the steepest gradient algorithm as follows. The objective function y is y = | Vd − V 0 |

q

(5.11)

where V 0 is the detected voltage corresponding to the perfectly matched condition, q is a constant (which determines the shape of the objective function), and Vd is the detected voltage proportional to the square root of the reflected power, √Pr , which is a function of V 1 and V 2 . The problem is to determine the values of V 1 and V 2 so as to minimize the objective function, y. The formulated update equation for V 1 and V 2 is given by

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Figure 5.66 Radiation patterns of the chip dielectric antenna (no ground type) at position 3. (Courtesy of Murata Mfg. Co., Japan.)

V i (n + 1) = V i (n) −

⌬y ␦, ⌬V i (n)

i = 1, 2

(5.12)

where ␦ is the update step, which is determined by the convergence speed of the objective function and the residue after convergence. Here, the constant q in (5.11) is chosen to be 2, as an appropriate value. The flow chart to perform the control algorithm is given in Figure 5.69. Figure 5.70 provides experimental results, showing variation of frequency representing the change of the impedance matching condition before and after the control. Figure 5.70(a) shows the matching condition at 150 MHz in free space as the initial state. Figure 5.70(b) shows frequency shift to the lower frequency when the human body exists near the antenna. Figure 5.70(c) shows the frequency returned to the matching condition at 150 MHz by implementing the steepest gradient algorithm to the control circuit, even though the antenna is affected by the human body. Figure 5.71 shows the convergence response of the control voltages when the distance D between the antenna and the human body is 2.5 cm. Figure 5.71(a) shows the

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Figure 5.67 Configuration of the active normal-mode helical antenna. NHA1: P = 1.9 mm, N = 49, 2R = 7.5 mm, L = 93; NHA2: P = 1.9 mm, N = 52, 2R = 7.5 mm, L = 101. (From: [128].  2004 IEICE. Reprinted with permission.)

Figure 5.68 (a) Equivalent circuit including the varactor diodes and detector circuit, and (b) the impedance matching concept. (From: [128].  2004 IEICE. Reprinted with permission.)

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Figure 5.69 Flow chart for control algorithm.

Figure 5.70 (a–c) Change of the impedance match condition due to the proximity of the human body. (From: [128].  2004 IEICE. Reprinted with permission.)

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Figure 5.71 Convergence response of the control voltages when the distance between the antenna and the human body, D, is 2.5 cm: (a) initial values are of voltages that achieve the matching condition in free space; and (b) initial values are of voltages that achieve the matching condition close to the human body. (From: [128].  2004 IEICE. Reprinted with permission.)

convergence response when the initial values of V 1 and V 2 are set to the voltages that achieve the matching condition in free space. The number of iteration in this case is 68. On the other hand, Figure 5.71(b) shows the convergence response when the initial values are set to the voltages that achieve the matching condition when the distance D is 2 cm. The convergence speed is reduced by about half as compared with the case in (a). It should be noted that the computation for every update needed 2 ms. The actual gain was measured as a function of D, and the gain improvement of 5 to 12 dB has been obtained by the tunable antenna system as compared with the antenna without the control system. 5.3.3.2 Transmission Diversity Technique For the MTs of PDC systems, implementation of the reception diversity system has so far been specified; however, transmission diversity system has never been required and hence only a main antenna has been used for transmitting. Therefore, if the main antenna performance deteriorates, the received power at base stations degrades as well, and accordingly the uplink communication quality becomes insufficient. To improve this problem, a transmission system in PDC MTs was proposed (Figure 5.72) [126]. To implement the transmission diversity system to a MT, detection of the

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Figure 5.72 Concept of transmission diversity for mitigation of the human body effect. (From: [126].  2004 IEICE. Reprinted with permission.)

performance degradation in the transmitting antenna is needed; however, it cannot be done only by a MT. Figure 5.73 shows ⌬G, the improvement factor of the transmitting power, with respect to r r , the unequal median for reception. ⌬G corresponds to the improvement of the received power at a base station. r r is defined by r r = G r2 /G r1

(5.13)

where G r1 and G r2 are the mean effective gain of the main and subantennas, respectively, at the receiving frequency, and are assumed to be equal each other in free space. The parameter ␳ in the figure is the correlation coefficient between the two antenna branches. Although the diversity effect in the multipath fading environment cannot be obtained by this transmission diversity system if the frequencies of the transmitting and receiving are close each other, remarkable improvement could be achieved for a case where the proximity effect of a human body or objects affecting the antenna characteristics exist. 5.3.3.3 Mode Reconfigurable Antenna Formed on Hinged Dielectric Cover One of the effective methods to maintain the high radiation efficiency, even when the human body effect exists, is to keep electromagnetic sources physically away from the human body. For this purpose, one way is to install a hinged dielectric cover to a MT unit, as shown in Figure 5.74, with an antenna placed inside the cover. This method is useful, since users usually open the cover while talking and thus the antenna is kept away

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Figure 5.73 Relationship between the unequal median for reception, r r , and the improvement of the transmitting power to that of the unloaded main antenna, ⌬G. (From: [126].  2004 IEICE. Reprinted with permission.)

Figure 5.74 (a, b) A sketch of a handset with a hinged dielectric cover.

from the operator’s head and hands. A dipole antenna formed on the hinged dielectric cover has been proposed in [129]. Figure 5.75 shows a configuration of an antenna placed on the inner surface of the hinged cover, in which (a) shows the open cover status and so the antenna is separated

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Figure 5.75 (a, b) Configuration of the antenna formed on the inner surface of the hinged cover and its excitation condition. (From: [129].  2002 IEICE. Reprinted with permission.)

from the ground plane, whereas (b) shows the cover in closed status so the antenna is above the ground plane. The figure also illustrates how the antenna is excited [129]. The antenna consists of two bent conductors, which have the same shape and are symmetrically installed onto the cover. When the cover is opened, the two antenna elements are excited with equal amplitude but out of phase (balanced feed), so that the parallel element acts as a two-wire transmission line, which feeds the two bent elements that act as a dipole antenna (dipole mode). When the cover is closed, on the other hand, the impedance bandwidth becomes narrow if the antenna operates in the dipole mode. However, if the two elements are excited with equal amplitude and in phase, the elements will act as a monopole antenna along with the ground plane (monopole mode). The antenna mode can be switched by changing a 180° delay line, either open or short, as shown in Figure 5.76.

Figure 5.76 A switching circuit to change the mode. (From: [129].  2002 IEICE. Reprinted with permission.)

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When the antenna operates in the dipole mode, the antenna impedance is determined basically by the element length L d , and the length L f of two-wire transmission line, and it is independent of the length L g of the ground plane. However, when the antenna operates with the monopole mode, L g affects the antenna impedance significantly, because strong current flows are induced on the surface of the ground plane. Figure 5.77 shows the relationships between the input resistance R and the bandwidth BW (VSWR ≤ 3) with respect to the length L g . The bandwidth for the monopole mode operation can be increased by the appropriate selection of the length L g . By switching the excitation mode and the proper choice of L g , the bandwidths of 17.6% and 11% are obtained in the cover open and closed conditions, respectively. Figure 5.78 compares the radiation efficiency of the antenna (dipole mode) with that of a conventional monopole antenna when the antennas are located in the vicinity of a spherical phantom simulating a real human head and inside the hand model. The parameter d denotes the distance between the handset and the surface of the phantom. In the figure, the simulation model, in which an antenna is located near the phantom, is illustrated as follows: (a) the side view, (b) the front view, and (c) the antenna located

Figure 5.77 Relationships between the input resistance and bandwidth (VSWR < 3) and the length of the ground plane, L g . (From: [129].  2002 IEICE. Reprinted with permission.)

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Figure 5.78 (a–c) Comparison of the radiation efficiency of the antenna (dipole mode) and that of a traditional ␭ /4 whip antenna when the antennas are located in the vicinity of a numerical spherical phantom. (From: [129].  2002 IEICE. Reprinted with permission.)

inside the hand model. It can be seen that the radiation efficiency is much higher than that of the conventional monopole antenna and the high radiation efficiency is maintained even though the antenna is close to the phantom. It is also experimentally confirmed that the antenna has the pattern averaged gain of −0.9 dBi in the horizontal plane for the talking situation. 5.3.4 Antenna Technologies for Reducing SAR Two methods for decreasing the SAR value while maintaining good radiation efficiency in the MT antenna are introduced here. One is to control the direction of radiation so as to weaken radiation toward the human body, and the other is to disperse radiation against the human body. 5.3.4.1 Dividing Fed Directive Control Antenna This method proposes the decrease of SAR by controlling the direction of radiation of MT antennas [130]. It is applicable to a clamshell type handset as shown in Figure 5.79,

299

Figure 5.79 (a, b) Antenna structure. (From: [130].  2003 IEICE. Reprinted with permission.)

where (a) illustrates an antenna structure called chassis frame dipole antenna (CFDA) which has two radiation elements (#3 and #4) that are connected by a line (#5) with matching circuit, and (b) shows three radiating elements (#1, #2, and #3) constituted on the upper clamshell chassis. The element #3 is fed by a port of a power divider located on the lower clamshell chassis. The elements #1 and #2 are fed by another port of the power divider through a switch (SW) and a 90° phase shifter. The radiation pattern is varied by switching the element, selecting either #1 or #2. This structure is referred to as dividing fed directive control antenna (DFDCA) [130]. Figure 5.80 shows the radiation patterns in free space when either the elements #2–#3 or the elements #1–#3 are fed. In the figure, both experimental and calculated (using FDTD) results are shown. When #2–#3

300

Figure 5.80 Radiation patterns of DFDCA in free space. (From: [130].  2003 IEICE. Reprinted with permission.)

are fed, the radiation decreases toward directions of −Z, +Y, and +X. This is advantageous in the talking situation when an operator holds the handset with the right hand, because the power absorption of the human body would be reduced, as the radiation toward the shoulder (−Z, +Y directions) and the head (+X direction) is smaller. On the other hand, when #1–#3 are fed, the radiation decreases toward directions −Z, −Y, and +X. Accordingly, this case is advantageous when a MT is held by the operator’s left hand, because the power absorption by the human head (+X direction) and the shoulder (−Z, −Y directions) would be smaller. Figure 5.81 shows the talking situation with the coordinates, and the radiation patterns in talking situation are shown in Figure 5.82, where both measured and

Figure 5.81 Talking situation. (From: [130].  2003 IEICE. Reprinted with permission.)

301

Figure 5.82 Radiation patterns of DFDCA at speaking position. (From: [130].  2003 IEICE. Reprinted with permission.)

calculated (using FDTD) results are given. In this measurement a phantom was used. The radiation toward the shoulder (between −X ′ and −Z ′ directions) is about 10 dB lower than that in the opposite direction. The effect of controlling the direction of radiation shown in Figure 5.80 is confirmed by the results shown in Figure 5.82. Figure 5.83 shows the ratio of absorbed power in various parts of the human body measured by using the phantom simulating a human body. In this figure, three types of handsets are taken as the test model: they are a CFDA type, a two-element type (which is a type of DFDCA not having the element #3), and a DFDCA type. From these results, the absorption powers of the DFDCA type, especially for a case where the handset is operated with the right hand, are observed smallest in every part of the human body as compared with the other types. In addition, since the radiation efficiency of DFDCA is best among the other two types, the DFDCA can be said to be the most useful type to decrease the SAR values. 5.3.4.2 Folded Dipole Antenna with a Planar Reflector This method proposes reduction of the maximum local SAR by dispersing radiation toward the human head [131]. The radiation power toward the head is lowered by using a small reflector, which is placed between the antenna and the head, and which disperses the radiated power. Figure 5.84 shows the studied model [131]. In Figure 5.84, the human head model is simulated with a dielectric cube, which has the dielectric constant 42.5, the electrical conductivity 1.51 S/m, the mass density 1,030 kg/m3, and the side length 200 mm. A folded dipole antenna is placed 15 mm apart from the surface of the dielectric cube. Also, a planar reflector of 10 mm wide and 51 mm long is placed between the

302

Figure 5.83 Absorption ratio in each human part. (From: [130].  2003 IEICE. Reprinted with permission.)

Figure 5.84 (a) Folded dipole and (b) human head model. (From: [131].  2001 IEICE. Reprinted with permission.)

antenna and the dielectric cube, with the distance of 9 mm from the feed point of the folded dipole antenna. In this method a folded dipole antenna having high self-impedance is used in order to make matching effective, even when the input impedance of the antenna decreases extremely because of the proximity effect of the planar reflector. Figure

303

5.85(a, b), respectively, shows the input impedance (R: resistance and X: reactance) and VSWR of the folded dipole antenna with respect to the reflector length L either in free space or near the dielectric cube. As shown in Figure 5.85, the variation of the input impedance against the reflector length L is small and the input impedance is well matched to the load 50 ohms so long as the L is longer than 0.525␭ . Figure 5.86 shows the SAR distribution along the x-axis on the surface of the dielectric cube shown in Figure 5.84. The SAR was measured by the thermographic method and the results agree well with the calculated results. It is found from Figure 5.86 that there are two peaks where the maximum local SAR is higher than the case where no planar reflector is used. With the reflector, the radiation energy is dispersed mainly in three directions (+15°, 0°, and −15°), and in two of these directions (+15° and −15°) the radiation power exceeds that of the case where no reflector is used and the power concentrates in the direction of 0°. The dispersion is caused by interference between the direct wave from the antenna and the scattered waves by the planar reflector. Therefore,

Figure 5.85 (a, b) Input impedance of folded dipole with a planar reflector. (From: [131].  2001 IEICE. Reprinted with permission.)

304

Figure 5.86 SAR distribution on surface of human head model (L = 0.55␭ ). (From: [131].  2001 IEICE. Reprinted with permission.)

the peak values of three points vary with respect to the reflector length L, as the phase of the direct wave and the phase of the scattered waves change relatively. The maximum local SAR is minimized when the reflector length L becomes 0.55␭ and three peaks become equal. Figure 5.86 shows this case. Figure 5.87 shows the maximum local SAR with respect to the reflector length L. The smallest maximum local SAR is observed at L = 0.55␭ . The peak value at x = 0 is higher than that in L < 0.55␭ , and the peak values at x = −15 and +15 are higher than that in L > 0.55␭ . This method would be useful to reduce the maximum local SAR for MT, as it needs only a small reflector, and realizes good impedance matching even when an MT is operated near the head. 5.3.5 Technique of Omitting Balun 5.3.5.1 Reduction of Unbalance-Mode in Unbalanced-Feed Dipole Antenna In general, when feeding a balanced type of antenna such as a dipole antenna through an unbalanced transmission line, a balun (i.e., balanced-unbalanced transformer) must be

305

Figure 5.87 Element length of a folded dipole with a planar reflector versus maximum local SAR. (From: [131].  2001 IEICE. Reprinted with permission.)

inserted between the antenna and the transmission line, otherwise the unbalanced current may flow on the outer surface of the transmission line. Small chip baluns are often used in MTs, because it is convenient to install them in a small space, as the small dimensions of the MT restrict the installation of bulky components. However, the chip baluns usually have large loss, consequently decreasing the actual gain of the antennas. If a balancemode antenna can be used without a balun, deterioration of the antenna performance could be made smaller. One method of feeding a balanced antenna without a balun is described below [132]. Figure 5.88(a) shows the configuration of the antenna considered here, which is an unbalanced line-fed dipole antenna installed at an edge of a finite ground plane. This antenna is equivalently divided into two models as shown in Figure 5.88(b, c), where (b) shows the unbalanced model (monopole mode) fed in phase and (c) shows the balanced model (dipole mode) fed out of phase. In the figure, the equivalent circuit of each model is also shown. These models are useful to investigate reduction of undesired unbalanced current flow on the ground plane. The frequency of interest is 2.045 GHz. Figure 5.89 shows the calculated radiation patterns in Z-X plane for each model shown in Figure 5.88 where the length of the ground plane L g = 100 mm and the separation of upper part of the element ⌬L e = 0 mm. The radiation pattern of the original model (a) can be interpreted as the sum of model (b) and model (c). The impedance of the original model (a) can be given as that of the parallel circuit consisting of monopole-mode impedance Z u and the dipole-mode impedance Z b , as shown in Figure 5.88. It is therefore considered that the power ratio of the monopole-mode radiation to that of the dipolemode radiation is determined by the ratio of the conductance of the monopole-mode to that of the dipole-mode. Figure 5.90 plots the conductance-ratio R u as a function of the ground plane length L g . The conductance-ratio R u is defined by

306

Figure 5.88 (a) Configuration of an unbalanced line-fed dipole antenna installed at an edge of a finite ground plane; (b) an unbalanced feed model; and (c) a balanced feed model, and their equivalent circuits. (From: [132].  2004 IEICE. Reprinted with permission.)

Figure 5.89 (a–c) Radiation patterns in Z-X plane for each model shown in Figure 5.88. L g = 100 and ⌬L e = 0 in mm. (From: [132].  2004 IEICE. Reprinted with permission.)

307

Figure 5.90 R u versus L g , where R u = G u /(G u + G b ). (From: [132].  2004 IEICE. Reprinted with permission.)

R u = G u /(G u + G b )

(5.14)

where G u = Re [1/Z u ], G b = Re [1/Z b ]. R u is obtained by using two models: one is from the model (a) given by a line with square dots, and another is from combination of two models (b) and (c) given by a line with circular dots. It is confirmed that R u varies periodically in half-wavelength intervals, as the monopole-mode impedance varies periodically with the electrical length of the ground plane. Hence, the monopole-mode (unbalanced-mode) radiation can be reduced by choosing L g = 50 mm, 125 mm, or 200 mm. In these conditions, a balun would be no longer needed. This technique has been successfully applied to 3G MTs. 5.3.6 Technology of Downsizing PIFA Downsizing of a PIFA placed on the ground plane was tried by loading a magnetic material on the antenna structure. Figure 5.91(a) shows the geometry of a PIFA installed on a conducting box [133], which is the most basic and simplest installation of a PIFA. Figure 5.91(b) shows the calculated current distributions on the surface of the antenna model (conductors) at 900 MHz. It is found that the current concentrates around the feed pin and the short pin of the PIFA, and on the bottom surface of the radiating element. It is considered that an efficient way of lowering the resonance frequency is to put magnetic materials on the place where relatively large current flows and consequently the dimensions of the antenna structure can be reduced. Figure 5.92 shows the four types of magnetic materials to be applied to the PIFA: (a) a rectangular plate, (b) a small rectangular plate, (c) a small cube, and (d) a rectangular

308

Figure 5.91 (a) Geometry of a PIFA on a conducting box, and (b) surface current distribution at 900 MHz. (From: [133].  2004 IEICE. Reprinted with permission.)

Figure 5.92 Some magnetic material structures: (a) plate; (b) a quarter plate; (c) fragment; and (d) rectangular loop. (From: [133].  2004 IEICE. Reprinted with permission.)

loop. The constitutive parameters of the magnetic materials are ␮ r = 2.5, tan ␦ m = 0.01, ⑀ r = 21.0, and tan ␦ e = 0.01. The typical arrangements of the magnetic materials investigated in [133] are illustrated in Figure 5.93, where (a) shows a magnetic fragment attached to the shorting pin, (b) shows a magnetic plate attached to the bottom of the radiating element, and (c) shows a magnetic rectangular loop attached to the bottom surface of the radiating

309

Figure 5.93 (a–c) Arrangements of the magnetic material. (From: [133].  2004 IEICE. Reprinted with permission.)

element. The calculated results of the impedance bandwidth and the rate of downsizing are summarized in Table 5.5. The efficient way of miniaturizing a PIFA has been confirmed by numerical simulations using HFSS. 5.4 EVALUATION OF ANTENNA PERFORMANCE 5.4.1 Measurement Method Using Optical Fiber In MTs like mobile phones equipped with small antennas, more or less radiation current flows on the conducting materials such as ground planes of the circuit board, metal chassis, and so forth. When evaluating the antenna performance, these currents should be included along with the currents on the antenna element. Care must be taken when the antenna performances are measured, because currents induced on the outside of the RF coaxial cable connecting the test antenna and the instrument may disturb accurate measurement. Conventionally, the radiation pattern is measured by using a voltage controlled oscillator (VCO) in the MT, instead of using a coaxial cable to avoid the disturbance due to the extra current on the cable. However, this method has a disadvantage that it cannot measure the phase pattern, but only the amplitude pattern when measuring radiation patterns of the antenna under test, because it cannot transmit the reference signal to the receiver of the measurement system. In addition, the radiation patterns cannot be measured in multiple frequencies at the same time, because the frequency is fixed to that of the VCO. If multifrequency measurements should be performed, the frequency of the VCO should be

310

Table 5.5 Computed Results of the Miniaturization Due to the Magnetic Material

Arrangement

Downsize Coefficient (%)

Relative Bandwidth (%)

Volume of Magnetic Material (mm 3 )

Unloaded (reference) (a) (b) (a) and (b) (c) (a) and (c)

100 81.8 57.3 46.4 80.0 60.5

12.2 12.2 8.9 7.8 12.2 8.9

0 24.0 2,450.0 2,004.0 780.0 689.0

The dimensions of the radiating element, except for the height, are changed so as to obtain the same resonant frequency of 900 MHz.

changed each time for the different frequency measurement. A disadvantage of doing this is the extra time required. In order to avoid these complexities, a novel measurement method was developed [134]. Figure 5.94 illustrates this system, where RF signals from a synthesizer are transformed to light signals by a laser diode (LD), and the light signals are transmitted to a MT antenna through an optical fiber cable. The light signals are transformed back to RF signals by a photo diode (PD) placed near the MT antenna, and then the RF signals excite the MT antenna. The components used in the measurement are the same as those in the conventional measurement systems, except for the optical fiber system. This method has the following features: 1. It can obtain more accurate results than the conventional method, because no undesired RF currents flow on the optical fiber cable. 2. It can measure the phase pattern of the radiation, because the reference signals from the synthesizer are available at the receiver. 3. It can obtain the radiation patterns in multiple frequencies at the same time by sweeping frequencies of the synthesizer over the desired frequency range. This method contributes to a great deal of time saving. The C/N of the measurement system shown in Figure 5.94 is 34 dB in 800-MHz bands. The system uses the hardware shown in Table 5.6. The directivity of the terminal antenna, the standard antenna, is 0 dBi, and that of the receiving antenna is 6 dBi. The span of the measurement is 5m [134]. This system can be applied to frequency ranges up to 10 GHz, as LD and PD, which can be used up to 10 GHz, are available presently. The accuracy of this measurement system was verified by using a test antenna system shown in Figure 5.95, which depicts a monopole antenna placed on a rectangular conducting box. Figure 5.96 shows examples of measured amplitude and phase of radiation patterns. Figure 5.96(a) shows the results obtained by the conventional method using the

311

Figure 5.94 Measurement system for radiation pattern using optical fiber. (From: [134].  2003 IEICE. Reprinted with permission.) Table 5.6 Hardware Used in the Measurement System H/W

Model Number

Synthesizer Converter Receiver LD PD

83630B (Agilent) 8511A (Agilent) 8530A (Agilent) FU45SDF (Mitsubushi Electric) FU39SPD (Mitsubushi Electric)

RF coaxial cable, while Figure 5.96(b) shows the results obtained by the method using the optical fiber cable. It is observed in Figure 5.96(a) that the measured results differ significantly from the calculated results in both amplitude and phase in radiation patterns, because of the errors caused by the influence of the RF coaxial cable in the measurement. On the other hand, the measured results shown in Figure 5.96(b) agree well with the calculated results in both amplitude and phase in the radiation patterns. This method enables us to obtain higher accuracy in the measurement as compared with the conventional method.

312

Figure 5.95 An antenna on a portable telephone. (From: [134].  2003 IEICE. Reprinted with permission.)

313

Figure 5.96 Measurement results by (a) conventional method using RF cables and (b) the new method using optical fiber. (From: [134].  2003 IEICE. Reprinted with permission.)

REFERENCES [1] Fujimoto, K., and J. R. James, Mobile Antenna Systems Handbook, 2nd ed., Norwood, MA: Artech House, 2000, p. 418, Figure 7.35. [2] Morishita, H., Y. Kim, and K. Fujimoto, ‘‘Design Concept of Antennas for Small Mobile Terminals and the Future Perspectives,’’ IEEE Antenn. Propag. Mag., Vol. 44, No. 5, 2002, pp. 30–42. [3] Fujimoto, K., ‘‘A Loaded Antenna System Applied to VHF Portable Communication Equipment,’’ IEEE Trans. of Vehc. Tech., Vol. VT-17, No. 1, 1968, pp. 6–12. [4] Hirasawa, K., and K. Fujimoto, ‘‘Characteristics of Wire Antenna on a Rectangular Conducting Body,’’ Trans. of IEICE, Vol. J65-B, No. 9, 1982, pp. 1133–1139 (in Japanese). [5] Fujimoto, K., ‘‘Antennas for Mobile Communications,’’ in Encyclopedia of RF and Microwave Engineering, Volume I, K. Chang (Ed.), New York: Wiley, 2005, pp. 352–355. [6] Hayashida, S., et.al., ‘‘Characteristics of Built-in Monopole Antenna for Handsets,’’ IEICE Trans. of FUNDAMENTAL/COMMUN/ELECTRON/INF/& SYST, Vol. E85-A/B/C/D, No. 1, 2002, pp. 1–9.

314

[7] Sievenpiper, D., et al., ‘‘High Impedance Electromagnetic Surface with a Forbidden Band,’’ IEEE Trans. of Microw. Theory Tech., Vol. 47, No. 11, 1999, pp. 2059–2074. [8] Yang, F., and Y. Rahmat Samii, ‘‘Reflection Phase Characteristics of the EBG Ground Plane for Low Profile Wire Antenna Applications,’’ IEEE Trans. Antennas Propag., Vol. 53, No. 1, 2003, pp. 2691–2703. [9] Kouveliots, M. K., ‘‘Antenna Reverberation Chamber,’’ in Encyclopedia of RF and Microwave Engineering, Volume I, K. Chang, (ed.), New York: John Wiley, 2005, pp. 239–251. [10] Fujimoto, K., et al., Small Antennas, Letchworth, U.K.: Research Studies Press, 1987. [11] Siwiak, K., Radiowave Propagation and Antennas for Personal Communications, 2nd ed., Norwood, MA: Artech House, 1998. [12] Chu, L. J., ‘‘Physical Limitations of Omni-Directional Antennas,’’ Journal of Applied Physics, Vol. 19, 1948, pp. 1163–1175. [13] Hansen, R. C., ‘‘Fundamental Limitations in Antennas,’’ Proceedings of the IEEE, Vol. 69, 1981, pp. 170–182. [14] McLean, J. S., ‘‘A Re-Examination of the Fundamental Limits on the Radiation Q of Electrically Small Antennas,’’ IEEE Trans. Antennas Propagat., Vol. 44, 1996, pp. 672–676. [15] Skrivervik, A. K., et al., ‘‘PCS Antenna Design: The Challenge of Miniaturization,’’ IEEE Antennas and Propagation Magazine, Vol. 43, No. 4, 2001, pp. 12–27. [16] Sato, K., et al., ‘‘Characteristics of a Planar Inverted-F Antenna on a Rectangular Conducting Body,’’ Electronics and Communications in Japan, Vol. 72, No. 10, 1989, pp. 43–51. [17] Vainikainen, P., et al., ‘‘Resonator-Based Analysis of the Combination of Mobile Handset Antenna and Chassis,’’ IEEE Trans. Antennas and Propagat., Vol. 50, No.10, 2002, pp. 1433–1444. [18] Kivekas, O., et al., ‘‘Effect of the Chassis Length on the Bandwidth, SAR, and Efficiency of Internal Mobile Phone Antennas,’’ Microwave and Optical Technology Letters, Vol. 36, No. 6, March 2003, pp. 457–461. [19] Villanen, J., et al., ‘‘Compact Antenna Structures for Mobile Handsets,’’ IEEE Vehicular Technology Conference, 2003, pp. 40–44. ¨ jefors, ‘‘A Bandwidth Enhancement Technique for Mobile Handset Antennas [20] Lindberg, P., and E. O Using Wavetraps,’’ IEEE Trans. on Antennas and Propagation, Vol. 54, No. 8, pp. 2226–2233. [21] http://www.fcc.gov/cgb/consumerfacts/hac.html. [22] Ying, Z., ‘‘Design of High Efficiency Multi-Band Antenna for Modern Mobilephones,’’ Proceedings of AP 2000 Conference, Davos, Switzerland, April 2000. [23] Ying, Z., ‘‘Multi-Band Non-Uniform Helical Antennas,’’ October 1996, International Patent (granted first in USA), US-006122102, WO-9815028. [24] Egorov, I., and Z. Ying, ‘‘A Non-Uniform Helical Antenna for Dual Band Cellular Phones,’’ IEEE AP-S, Salt Lake City, UT, September 2000. [25] Haapala, P., ‘‘Helical Antennas for Multi-Mode Mobile Phones,’’ June 1995, Patent, EP-747990. [26] Era¨tuuli, P., P. Haapala, and P. Vainikainen, ‘‘Dual Frequency Wire Antennas,’’ Electronics Letters, Vol. 32, No. 12, 1996, pp. 1051–1052. [27] Hayes, G. J., ‘‘Dual Band Meander Antenna,’’ International Patent, US-0459553, WO-US08058, 1996. [28] Blom, C. G., ‘‘Spiral Impedance Transformer,’’ International Patent, SE-0001872, WO-9637007, May 1995. [29] Ying, Z., ‘‘Multi-Band Branch Meander Antenna,’’ International Patent, WO-09922419, WO-09922420, US-6329962 (granted first in USA), October 1997. [30] Andersson, J., ‘‘Multi-Band Antenna for Use in a Portable Telecom’s Apparatus,’’ Patent, November 1999, US-6504511; ‘‘Broad Band Low Profile Monopole Antenna,’’ March 2002, patent application, US-366514, EP-0005816. [31] Teng, P. L., and K. L. Wong, ‘‘Planar Monopole Folded into a Compact Structure for Very-Low-Profile Multi-Band Mobile Phone Antenna,’’ Microwave and Optical Tech. Lett., Vol. 33, No. 1, 2002, pp. 22–25.

315

[32] Taga, T., and K. Tsunekawa, ‘‘Performance Analysis of a Built-In Planar Inverted-F Antenna for 800 MHz Band Portable Radio Units,’’ IEEE Journal on Selected Areas on Communications, Vol. 5, No. 5, 1987, pp. 921–929. [33] Liu, Z. D., P. S. Hall, and D. Wake, ‘‘Dual-Frequency Planar Inverted F Antenna,’’ IEEE Trans. Antennas Propagat., Vol. 45, October 1997, pp. 1451–1458. [34] Korisch, I. A., ‘‘Planar Dual Frequency Band Antenna,’’ U.S. Patent, US-5926139, July 20, 1999. [35] Geerert, F., ‘‘Dual Band Antenna for a Hand Portable Telephone and a Corresponding Hand Portable Telephone,’’ European Patent, EP-1018777, December 10, 1999. [36] Pankinaho, I., ‘‘Double Resonance Antenna Structure for Several Frequency Ranges,’’ U.S. Patent, US-6140966, October 31, 2000. [37] Ying, Z., ‘‘Printed Twin Spiral Dual Band Antenna,’’ International Patent, July 1998 (granted first in USA), US-006166694, WO-200003452. [38] Ying, Z., ‘‘Printed Multi-Band Patch Antenna,’’ International Patent, September 1998 (granted first in USA), WO-200036700, US-0212259. [39] Tarvas, S., and A. Isohatala, ‘‘An Internal Dual-Band Mobile Phone Antenna,’’ 2000 IEEE Antennas Propagat. Soc. Int. Symp. Dig., pp. 266–269. [40] Salonen, P., M. Keskilammi, and M. Kivikoski, ‘‘New Slot Configurations for Dual-Band Planar Inverted-F Antenna,’’ Microwave Opt. Technol. Lett., Vol. 28, March 5, 2001, pp. 293–298. [41] Viratelle, D., and R. J. Langley, ‘‘Dual-Band Printed Antenna for Mobile Telephone Applications,’’ IEE Proc. Microwave Antennas Propagat., Vol. 147, October 2000, pp. 381–384. [42] Martinez-Vazquez, M., M. Geissler, and D. Heberling, ‘‘Volume Considerations in the Design of DualBand Handset Antennas,’’ 2001 IEEE Antennas Propagat. Soc. Int. Symp. Dig., pp. 112–115. [43] Hsiao, F. R., et al., ‘‘A Dual-Band Planar Inverted-F Patch Antenna with a Branch-Line Slit,’’ Microwave Opt. Technol. Lett., Vol. 32, February 20, 2002, pp. 310–312. [44] ICNIRP, ‘‘Guidelines for Limiting Exposure to Time-Varying Electric, Magnetic and Electromagnetic Fields (Up to 300 GHz),’’ Health Physics, Vol. 74, 1998, pp. 449–522. [45] Hayes, G., and S. Vance, ‘‘Branched Multi-Band Antenna,’’ Patent, US-6,903,686, 1999. [46] Talvitie, O., ‘‘Planar Inverted Antenna Structure for Mobile Stations, Includes Groove Formed in Radiator Such That Line Segment to Be Created Between Feed Point and Ground Point Cuts Groove,’’ U.S. Patent, US-6,597,317, October 27, 2000. [47] Vance, S., ‘‘Planar Inverted F Antenna for Wireless Terminal, Has Conductive Element Comprising Looped Track Element That Resonates at High and Low Bands,’’ U.S. Patent, US-6,909,402, June 11, 2003. [48] Ishimiya, K., J. Takada, and Z. Ying, ‘‘Progress of Multi-Band Antenna Technology for Mobile Phone,’’ Proceedings of 2006 IEICE Society Conference, Symposium, BS-1-2, September 2006. [49] Ying, Z., and A. Dahlstro¨m, ‘‘Compact Multi-band Printed Branch Antenna,’’ Patent, US-6456250 B1, WO-200191233, May 19, 2000. [50] Egorov, I., ‘‘Fold PIFA Antenna with a Parasitic Element,’’ Patent, US-6,614,400, August 7, 2000. [51] Ying, Z., et al., ‘‘Multi Band and Multi Antenna System,’’ Patent (granted first in Sweden), SE-0101067-7, SE-0102183-1, December 20, 2000. [52] Ying, Z., and J. Andersson, ‘‘Multi-Band, Multi Antenna System for a Modern Mobile Phone,’’ Proc. Nordic Antenna Symposium (Antenn’03 ), Kalmar, Sweden, May 2003. [53] Ying, Z., ‘‘Some Important Antenna Innovations in the Mobile Terminal Industry in the Last Decade,’’ Proceedings of EuCAP 2006, Nice, France, November 2006. [54] Rowell C. R., ‘‘A Capacitive Loaded PIFA for Compact Mobile Telephone Handsets,’’ IEEE Trans. on Antennas and Propagation, Vol. 45, No. 5, May 1997, pp. 837–842. [55] Ying, Z., and A. Dahlsto¨rm, ‘‘Multi-band Dual Layer C-Fed Antenna,’’ International Patent, (granted in USA), US-6650294, May 2001. [56] Nu´n˜ez Calvo, F., Z. Ying, and A. K. Skrivervik, ‘‘Design and Optimization of a Pentaband Terminal Antenna,’’ Proceedings of ICECom 2005, Dubrovnik, Croatia, August 2005.

316

[57] Bergh, J., and M. Eriksen, ‘‘Investigations of Compact Mobile Phone Antennas for Multi-Band Wireless Applications,’’ Master’s thesis, Lund Institute of Technology, August 2005. [58] Iellici, D., et al., ‘‘Hybrid Antenna Using Parasitic Excitation of Conducting Antenna by Dielectric Antennas,’’ Patent application, WO-2004/114462 A1, December 2004. [59] Ishimiya, K., ‘‘Multi-Band Antenna Device,’’ Patent application, JP-2005-309345, October 2005. [60] Cassel, E., ‘‘Antenna for portable Communications Device Has Hourglass-Shaped Inner Patch Portion Parallel to Ground Plane Back-Up Element And Metallic Wall Feeding Reflected Wave to Open Slot,’’ Patent, US-6,680,704, May 3, 2001. [61] Ishimiya, K., Z. Ying, and J. Takada ‘‘GSM + UMTS Penta-band Folded Dipole Antenna for Mobile Phone,’’ Proceedings of 2006 IEICE General Conference, B-1-132, March 2006. [62] Louhos, J.-P., ‘‘Practical Matching Techniques for Integrated Antennas,’’ Master’s thesis, Helsinki University of Technology, Finland, March 1999. [63] Hayes, J., et al., ‘‘Multi-Frequency Inverted-F Antenna for Mobile Communications Device,’’ Patent, US 6,662,028, May 22, 2000. [64] Ying, Z., and K. Hankansson, ‘‘Multi Band Antenna with Tuneable Loading Parasitic Element,’’ Patent (granted first in Denmark), GB-0007239, WO-EP02972, September 2000. [65] Louhos, J., and I. Pankinaho, ‘‘Electrically Tunable Multiband Planar Antenna for Example for Mobile Telecommunication Device,’’ Patent, US-6,693,594, April 2, 2001. [66] Kivekas, O., et al., ‘‘Radio Antenna for Personal Mobile Communication Terminal, Adjusts Frequency Shift from Radiating Element by Changing Length of Transmission Line,’’ Patent, US-6,650,295, January 28, 2002. [67] Thornell-Pers, A., ‘‘Antenna Device and Portable Radio Communication Device Comprising Such an Antenna,’’ Patent Application, International publication number: WO2005/07470 A1, October 2005. [68] Gupta, K. C., et al., ‘‘MEMS-Based Reconfigurable Slot Antennas,’’ Proc. of the Workshop on Affordability and Cost Reduction for Radar Systems, AL, April 5–6, 2000. [69] Cetiner, B. A., et al., ‘‘A Compact Wideband MEMS Switched Diversity Antenna for Indoor Mobile Channels,’’ IEEE MTT-S Int. Microwave Symp., 2003. [70] Bjerre, B. H., and I. Marqvardsen, ‘‘Acoustic Resonant Chamber Has Completely or Partly Located Within the Antenna Chamber,’’ Patent, WO-200038475, DK-0001700, December 1998. [71] Karlsson, K., and Z. Ying, ‘‘Combination of Full Wave Simulations and Equivalent Circuit Models in Predicting Coupling Between Antenna Element and Loudspeaker in Mobile Phone,’’ Proceedings of IWAT 2007, Cambridge, U.K., March 2007. [72] Ying, Z., and W. Shi, ‘‘Integrated Antenna Element as a Flat Panel Speaker,’’ Patent application, EP-04001567.9, 2002-12. [73] Urban, R., and C. Peixeiro, ‘‘Battery Effects on a Small Handset Planar Inverted-F Antenna,’’ IST Mobile & Wireless Communications Summit, Lyon, France, June 2004. [74] Sjo¨blom, P., and H. Sjo¨land, ‘‘An Adaptive Impedance Tuning CMOS Circuit for ISM 2.4 GHz Band,’’ IEEE Trans. on Circuit and System I (CAS-I ), Vol. 52, No. 6, June 2005, pp. 1115–1124. [75] Vance, S., Internal Antenna Design Guideline, Sony Ericsson Mobile Communication AB internal report, October 2006. [76] Vaughan, R. G., and J. B. Andersen, ‘‘Antenna Diversity in Mobile Communications,’’ IEEE Trans. Vehic. Technol., Vol. VT-36, No. 4, pp. 149–172, November 1987. [77] Taga, T., ‘‘Analysis for Mean Effective Gain of Mobile Antennas in Land Mobile Radio Environment,’’ IEEE Trans. Veh. Technol., Vol. 39, 1990, pp. 117–131. [78] Dietrich, Jr., C. B., et al., ‘‘Spatial, Polarization, and Pattern Diversity for Wireless Handheld Terminals,’’ IEEE Trans. Antennas and Propagat., Vol. 49, No. 9, September 2001, pp. 1271–1281. [79] Green, B. M., and M. A. Jensen, ‘‘Diversity Performance of Dual-Antenna Handsets Near Operator Tissue,’’ IEEE Trans. Antennas and Propagat., Vol. 48, No. 7, July 2000, pp. 1017–1024. [80] Ying, Z., et al., ‘‘Characrizing of Multi-Channel Antenna Performance for Mobile Terminal by Using Near Field and Far Field Parameters,’’ COST273, Go¨teborg, Sweden, 2004-6.

317

[81] Ying, Z., et al., ‘‘Diversity Antenna Terminal Evaluation,’’ Proceedings of IEEE AP-S 2005, Washington, D.C., 2005-07. [82] Blanch S., J. Romeu, and I.Corbella, ‘‘Exact Representation of Antenna System Diversity Performance from Input Parameter Description,’’ IEE Electronics Letters, Vol. 39, 2003, pp. 705–707. [83] Derneryd, A., and G. Kristensson, ‘‘Antenna Signal Correlation and Its Relation to the Impedance Matrix,’’ IEE Electronics Letters, Vol. 40, 2004, pp. 401–402. [84] Arai, H., Measurement of Mobile Antenna Systems, Norwood, MA: Artech House, 2001, Chapter 2. [85] Arai, H., and T. Urakawa, ‘‘Radiation Power Measurement Using Compact Shield Box,’’ Proc. ISAP, 1996, 4B-13. [86] Serafimov, N., ‘‘Radiation Efficiencies of Phone Antennas and Radiated Power of Mobile Phones Measured in Two Anechoic Chambers and Two Reverberation Chambers,’’ Master thesis, Department of Electromagnetics, Chalmers University of Technology, December 2001. [87] Rosengren, K., ‘‘Measuring Radiation Efficiency of Terminal Antennas in Reverberation Chambers,’’ Thesis for Licentiate degree, Chalmers University of Technology, April 2001. [88] Hallbjo¨rner, P., and K. Madse´n, ‘‘Terminal Antenna Diversity Characterization Using Mode Stirrer Chamber,’’ Electronics Letters, Vol. 37, 2001, pp. 273–274. [89] Kildal, P. -S., and K. Rosengren, ‘‘Electromagnetic Analysis of Effective and Apparent Diversity Gain of Two Parallel Dipoles,’’ IEEE Antennas and Wireless Propagtion Letters, Vol. 2, 2003, pp. 9–13. [90] Rosengren, K., and P.-S. Kildal, ‘‘Radiation Efficiency, Correlation, Diversity Gain, and Capacity of a Six Monopole Antenna Array for a MIMO System: Theory, Simulation and Measurement in Reverberation Chamber,’’ Proceedings IEE, Microwave Antennas Propagation, Vol. 152, No. 1, February 2005, pp. 7–16. [91] Vaughan, R., and J. B. Andersen, Channels, Propagation, and Antennas for Mobile Communications, London: IEE, 2003. [92] Plicanic, V., ‘‘Diversity Antenna Studies and Evaluations,’’ Master’s thesis, Lund Institute of Technology, 2004. [93] Kildal, P.-S., and K. Rosengren, ‘‘Correlation and Capacity of MIMO Systems and Mutual Coupling, Radiation Efficiency and Diversity Gain of Their Antennas: Simulations and Measurements in Reverberation Chamber,’’ IEEE Communications Magazine, August 2004. [94] Avendal, J., B. Lau, and Z. Ying, ‘‘Multiband Diversity Antennas,’’ Proceedings of iWAT 2007, Cambridge, London, U.K., March 2007. [95] Rogers, P. R., ‘‘An Electrically-Small Tuneable Annular Slot for Future Mobile Terminals,’’ Ph.D. dissertation, University of Bristol, Department of Electrical and Electronic Engineering, 2004. [96] Fletcher, P. N., M. Dean, and A. R. Nix, ‘‘Mutual Coupling in Multi-Element Array Antennas and its Influence on {MIMO} Channel Capacity,’’ Electronics Letter, Vol. 39, No. 4, February 2003. [97] Browne, D. W., and Y. Rahmat-Samii, ‘‘Antenna Topology Impacts on Measured MIMO Capacity,’’ Proc. IEEE 62nd Vehicular Technology Conference (VTC’2005-Fall ), Vol. 1, Dallas, TX, September 2005, pp. 339–344. [98] Kivinen, J., et al., ‘‘Experimental Investigations of MIMO Propagation Channels,’’ Proc. IEEE Int. Symp. Antennas and Propagation, 2002, pp. 16–21. [99] Sulonen, K., et al., ‘‘Comparison of MIMO Antenna Configurations in Picocell and Microcell Environments,’’ IEEE J. Select. Areas Commun., Vol. 21, 2003, pp. 703–712. [100] Waldschmidt, C., S. Schulteis, and W. Wiesbeck, ‘‘Complete RF System Model for the Analysis of Compact MIMO Arrays,’’ IEEE Trans. on Vehicular Technology, Vol. 53, No. 3, May 2004, pp. 579–586. [101] Jansen, M., and Y. Rahmat-Samii, ‘‘EM Interaction of Handset Antennas and a Human in Personal Communications,’’ Proceedings of the IEEE, Vol. 83, No.1, January 1995, pp. 7–17. [102] Manteuffel, D., et al., ‘‘Fundamental Aspects for the Design of Low-SAR Mobile Phones,’’ IEE AMS-2002—Technical Seminar on Antenna Measurement and SAR, Loughborough, U.K., 2002, pp. 25/1–25/5.

318

[103] Pedersen, G. F., ‘‘Antennas for Mobile Phones,’’ in Wireless Flexible Personalised Communications, COST 259: European Co-Operation in Mobile Radio Research, L. M. Correia, (ed.), New York: John Wiley & Sons, 2001. [104] Ogawa, K., T. Matsuyoshi, and K. Monma, ‘‘An Analysis of the Performance of a Handset Diversity Antenna Influenced by Head, Hand and Shoulder Effects at 900 MHz: Part I–Effective Gain Characteristics; and Part II–Correlation Characteristics,’’ IEEE Trans. on Vehicular Technology, Vol. 50, No. 3, May 2001, pp. 830–853. [105] http://www.sam.phantom.com. [106] Sangin, S., and H. ≈kesson, ‘‘Simulation of Handset Antenna and SAR,’’ Master’s thesis, Lund Institute of Technology, 2005. [107] http://www.speag.com. [108] Iversen, P. O., et al., ‘‘Real-Time Spherical Near Field Antenna Test Facility for Personal Communications Applications,’’ Proceedings of IEEE AP2000, Davos, 2000-04. [109] Geissler, M., et al., ‘‘An Improved Method for Measuring the Radiation Efficiency of Mobile Devices,’’ Antennas and Propagation Society International Symposium, 2003 IEEE, Vol. 4, June 22–27, 2003. [110] http://www.satimo.com. [111] Andersson, J., and Z. Ying, ‘‘Different Sampling Techniques in a Reverberation Chamber,’’ Proceedings of EuCAP 2006, Nice, France, 2006. [112] Wheeler, H. A., ‘‘The Radiation Sphere Around a Small Antenna,’’ Proc. IRE, Vol. 47, 1975, pp. 457–461. [113] http://www.ctia.org/. [114] Test Plan for Mobile Station Over the Air Performance—Method of Measurement for Radiated RF Power and Receiver Performance, Version 2.0, March 2003, CTIA Certification Department Program, http:// www.ctia.org. [115] http://www.cetecom.com. [116] Hirasawa, K., and K. Fujimoto, ‘‘Characteristics of Wire Antennas on a Rectangular Conducting Body,’’ IEICE Trans., B, Vol. J65-B, No. 9, 1982, pp. 1133–1139 (in Japanese). [117] Fujimoto, K., et al., Small Antennas, Letchworth, U.K.: Research Studies Press, 1987, Chapter 4. [118] Morishita, H., Y. Kim, and K. Fujimoto, ‘‘Design Concept of Antennas for Small Mobile Terminals and the Future Perspective,’’ IEEE Antennas and Propagation Magazine, Vol. 44, No. 5, 2002, pp. 30–43. [119] http://www.murata.co.jp/articles/ta05d1.html. [120] Kim, Y., et al., ‘‘A Folded Loop Antenna System for Handsets Developed and Based on the Advanced Design Concept,’’ IEICE Trans Commun., Vol. E84-B, No. 9, 2001, pp. 2468–2475. [121] Hayasida, S., H. Morishita, and K. Fujimoto, ‘‘A Wideband Folded Loop Antenna for Handsets,’’ IEICE Trans., B, Vol. J86-B, No. 9, 2003, pp. 1799–1805 (in Japanese). [122] Sekine, S., et al., ‘‘Design Method for a Broadband Inverted-F Antenna by Parallel Resonance Mode,’’ IEICE Trans., B, Vol. J86-B, No. 9, 2003, pp. 1806–1815 (in Japanese). [123] Fukasawa, T., K. Kodama, and S. Makino, ‘‘Diversity Antenna for Portable Telephone with Two Boxes,’’ IEICE Technical Report, Vol. TGA, P2002-78, July 2002 (in Japanese). [124] Nishioka, Y., et al., ‘‘Mobile Phone Antenna with Variable Radiation Pattern and Its Characteristics,’’ IEICE Technical Report, Vol. TGA, P2005-114, November 2005. [125] Soutome, H., et al., ‘‘A Polarization Diversity Antenna for Wireless Terminal,’’ Proc. the ISAP 2006, Singapore, November 1–4, 2006, SD2-2. [126] Iwai, H., K. Ogawa, and Y. Koyanagi, ‘‘A Proposal for a Transmission Diversity for Mitigation of the Human Body Effect of Handsets and Its Performance Evaluation,’’ IEICE Trans., B, Vol. J87-B, No. 9, 2004, pp. 1277–1286 (in Japanese). [127] Murata Mfg. Co., Japan. [128] Ogawa, K., et al., ‘‘Automatic Impedance Matching of an Active Antenna Near the Human Operator by the Steepest Gradient Algorithm,’’ IEICE Trans., B, Vol. J87-B, No. 9, 2004, pp. 1287–1298 (in Japanese). [129] Fukasawa, T., et al., ‘‘Characteristics of Dipole Antenna Built In Flip of Portable Telephone,’’ IEICE Trans., B, Vol. J85-B, No. 6, 2002, pp. 941–952 (in Japanese).

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[130] Haruki, H., Y. Koyanagi, and K. Egawa, ‘‘Radiation Characteristics of an Antenna System with a Dividing Feed Circuit for a Handy Phone in the Talk Position,’’ IEICE Technical Report, TGA, P2003-164, 2003 (in Japanese). [131] Okano, Y., et al., ‘‘A Study on the Reduction of the Local SAR by Use of the Folded Dipole Antenna with a Planar Reflector,’’ IEICE Technical Report, TGAP2001-159, December 2001 (in Japanese). [132] Fukasawa, T., et al., ‘‘Characteristics of the Dipole Antenna on a Portable Telephone Without a Balun,’’ IEICE Technical Report, AP2004-54, July 2004, pp. 79–83 (in Japanese). [133] Tanaka, T., et al., ‘‘A Study on Miniaturization of Antenna for Handset Utilizing Magnetic Materials,’’ IEICE Trans., B, Vol. J87-B, No. 9, 2004, pp. 1327–1335 (in Japanese). [134] Fukasawa, T., K. Shimomura, and M. Ohtsuka, ‘‘Accurate Measurement Method for Characteristics of an Antenna on a Portable Telephone,’’ IEICE Trans., B, Vol. J86-B, No. 9, 2003, pp. 1895–1905 (in Japanese).

Chapter 6 Radio Frequency Exposure and Compliance Standards for Mobile Communication Devices C-K. Chou and Ron Petersen

The study of the biological effects associated with exposure to electromagnetic energy has a rich history going back almost a century. Although much of the earlier work was carried out as a matter of scientific curiosity, since the mid-1950s the majority of the research has been focused on filling gaps in the knowledge-base regarding safety in order to develop rational radio frequency (RF) safety standards and guidelines to protect against established adverse health effects in humans. Members of the public and RF workers continue to raise questions about the safety of new RF technologies, including radar, radio and television broadcasting facilities, microwave ovens, point-to-point microwave radio, and satellite communications systems. The most recent concern is the safety of mobile and portable telephones and their base stations. Consequently, much of the bioeffects research carried out during the past 15 years is specific to conditions relative to exposure to portable telephones. The results of this research are used to ensure that contemporary safety guidelines and standards adequately protect the public and the worker, or if changes are necessary. Two types of standards directly related to the safety of mobile communication devices are described in this chapter: (1) safety standards that recommend limits to protect against harmful effects associated with RF exposure, and (2) conformance (or compliance) standards that describe protocols to ensure that RF-emitting devices, such as portable telephones, comply with the safety standards. 321

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6.1 INTRODUCTION Public awareness of the dramatic increase in the number of systems that emit RF energy frequently leads to questions about safety. For example, during the past few decades, questions have arisen about the safety of radar, radio and television broadcasting facilities, microwave ovens, point-to-point microwave radio, and satellite communications systems, and most recently, mobile and portable telephones and their base stations. The range of RF power at which mobile and portable wireless communication devices operate may be as low as a few mW for a Bluetooth device; a fraction of a watt for a mobile phone; up to 7W for two-way mobile radios; several tens of watts for mobile radio systems installed in motor vehicles; and up to 100W, or more, for certain mobile telephone and two-way radio base stations. Even though they operate at lower power than base station and vehiclemounted mobile radio antennas, handheld devices have the potential for producing higher exposures, especially to important organs such as the brain and eyes, because of their proximity to the caller’s body during normal use. Although exposure from base station antennas is far less than that from handheld devices, the public appears to be more concerned about the safety of base stations. Sound, science-based safety standards help to allay the fears of those who approach the RF safety issue with an open mind. In this chapter, the relevant parameters used to assess exposure, and the types of standards that address the safety of mobile communication devices are described— specifically safety standards that recommend limits to protect against harmful effects associated with RF exposure, and conformance (or compliance) standards that describe protocols to ensure that RF-emitting devices comply with the safety standards. For purposes of this chapter, the frequency range of interest is 30 MHz to 6 GHz, which includes the frequencies most commonly used for mobile communications. 6.2 PHYSICAL PARAMETERS Radio frequencies are loosely defined as frequencies between 3 kHz and 300 GHz—that is, frequencies below the infrared region of the electromagnetic spectrum. Because the photon energy associated with an RF electromagnetic wave is far below that required to remove an electron from an atom (ionization), RF exposure is characterized as nonionizing radiation, as is infrared radiation, visible light, and the longer ultraviolet wavelengths. The physical interaction of RF energy with biological material is complex, often resulting in highly nonuniform distributions of the induced electric (E) and magnetic (H) fields and the induced current density within the object regardless of the uniformity of the external exposure fields. The internal fields are related to a dosimetric quantity, called specific absorption rate (SAR), which was first proposed by the National Council on Radiation Protection and Measurements in 1981 [1], and defined as the time derivative of the incremental energy absorbed by (dissipated in) an incremental mass contained in a volume of a given density and is expressed in W/kg. The internal electric field strength,

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induced current density, and SAR are related to the physical and electrical properties of the absorbing object by the following equations: SAR =

E=

␴ 2 E ␳

冉 冊 ␳ SAR ␴

W/kg

(6.1)

1/2

J = (␴␳ SAR )1/2

V/m

(6.2)

A/m

(6.3)

where E is the root-mean-square value of the induced electric field strength (V/m) in tissue, J is the current density (A/m2) in tissue, ␳ is the tissue density (kg/m3), and ␴ is the dielectric conductivity of the tissue (S/m). In a tutorial on RF dosimetry, Chou et al. [2] discuss the relationship between SAR and the characteristics of the incident field and the geometrical and electrical properties of the absorbing object. SAR patterns, whole-body averaged SAR, and methods for the measurement of peak SAR, are also discussed. (Details for the measurement of peak SAR for mobile phones and other portable devices are described in Section 6.5.) In order to determine the thresholds for harmful effects and develop exposure limits to protect against such effects, it is necessary to know the magnitude and distribution of the SAR within the exposed object. The SAR depends not only on the properties of the incident field, including the magnitudes of E and H (or equivalent power density); it also depends on the dielectric properties, geometry, size, and orientation of the exposed object, the polarization and frequency of the incident fields, the source configuration, exposure environment, and time-intensity factors. Figure 6.1 shows the parameters associated with human exposure to RF energy. 6.3 TYPES OF RF SAFETY STANDARDS There are three types of RF standards related to human safety. The first type is the ‘‘safety’’ standard, which sets limits to protect against harmful effects associated with RF exposure. Currently two recognized international organizations develop RF safety standards and guidelines. One, now called the Institute of Electrical and Electronics Engineers (IEEE) International Committee on Electromagnetic Safety (ICES) Technical Committee 95, has a history of RF safety standard activities that traces back to the late 1950s. The first RF safety standard was published by this committee in 1966 [4]; four revisions have been published since then—the latest in 2006 [5]. This committee develops

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Figure 6.1 External and internal physical parameters of human exposure to RF energy. (Modified from Guy [3].)

standards through an open consensus process that is transparent at every level; that is, the committee is open to anyone with an expressed material interest, the meetings are open, and meeting records are posted on the Internet. A total of 130 members representing 24 countries were involved with developing the latest revision of this standard (IEEE C95.1-2005) [5], including members of government, academia, industry and the general public. (See Petersen [6] for a detailed historical record.) In 2006, this standard was approved by the American National Standards Institute and is recognized as an American National Standard (ANSI/IEEE C95.1-2006). The second international organization that develops RF safety guidelines is the International Commission on Non-Ionizing Radiation Protection (ICNIRP), which consists of 14 elected members from various government organizations and academia (but no members representing commercial interests). The ICNIRP guidelines, developed mostly in closed forums, are endorsed and promoted globally by the World Health Organization for adoption by national governments. Most countries in the world adopt the basic restrictions or derived limits of either the ICNIRP guidelines or the IEEE standard. Similarities and differences in the recommendations from IEEE and ICNIRP are presented in Section 6.4.3. The second type of standard is the product standard which recommends methodologies for ensuring products comply with the safety standards. The committees that develop international product standards for mobile communications devices are IEEE ICES Technical Committee 34 (TC-34) and International Electrotechnical Commission (IEC) TC-106. TC-34 is a relatively new committee established in 1995 (compared with ICES TC-95,

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which was established as an American Standards Association Committee in 1960); IEC TC-106 was established in 2000. Although TC-34 and TC-95 are both ICES committees, the TC-34 product standard for mobile telephones IEEE 1528-2003 [7] is used for determining compliance with TC-95 and ICNIRP recommendations to allow manufacturers to readily ensure that their products comply with these or similar requirements. The goal is to provide unambiguous procedures that yield repeatable results (e.g., similar to the procedure for certifying compliance of microwave ovens). In addition to standards for measuring the peak SAR associated with handheld mobile telephones, TC-34 is in the process of developing product standards for vehicle-mounted antennas, as well as for other devices using both measurement and numerical techniques [8]. Recent collaboration between ICES TC-34 and IEC TC-106 led to the development of the product standard for hand-held devices IEC 62209-1 [9], which is harmonized with IEEE 1528-2003. The third type of RF safety standard protects against indirect effects associated with RF energy. Examples of this type of standard include compatibility standards (e.g., standards for limiting electromagnetic interference with electronic equipment on aircraft or in medical environments). Compatibility standards, developed by the American National Institute of Standards, International Standard Organization, Consumer Electronics Association and others, are not discussed further in this chapter. 6.4 EXPOSURE STANDARDS As early as the mid-1950s, recommendations to limit exposure to RF energy were adopted by various agencies and organizations throughout the world. The first RF exposure standard published in the United States (USAS C95.1-1966) [4] limited RF-induced heating of the body. The recommended exposure limit was 100 W/m2 averaged over any 0.1-hr interval; the applicable frequency range was 10 MHz to 100 GHz. In the mid-1970s, dosimetry studies revealed that the interaction of RF energy with biological bodies is extremely complex, and a frequency-independent limit over a broad frequency range is unrealistic. The third revision of the 1966 standard (American National Standards Institute ANSI C95.1-1982) [10] incorporated dosimetry, which resulted in frequency-dependent limits based on whole-body-averaged and peak spatial-average SAR (to address localized exposure). In 1986, the National Council on Radiation Protection and Measurements (NCRP) adopted the 1982 ANSI standard as the upper tier for occupational exposure, but added an additional safety factor of 5 for a lower tier for exposure of the public [11]. The upper tier includes a 10-fold safety factor; the lower tier has an additional factor of 5 (i.e., a total safety factor of 50 below the threshold for effects considered adverse). The IEEE Committee adopted this approach, and the revision of the 1982 C95.1 standard (IEEE C95.1-1991) [12] also contains two tiers, as does the 1998 ICNIRP guidelines [13]. Although the ICNIRP guidelines and the 1991 IEEE standard are based on limiting the whole-body-averaged SAR to the same values of 0.4 and 0.08 W/kg for the upper and lower tiers, respectively, the peak spatial-average SAR limits differ, both in magnitude

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and in averaging volume. This discrepancy caused confusion for the general public, extra burdens for manufacturers, and discordance among the regulators. During the revision process that led to IEEE C95.1-2005 [5], consideration was given to harmonizing with the ICNIRP guidelines where scientifically justifiable. An important issue that was addressed is the peak SAR limits which are now essentially identical in the new IEEE standard and ICNIRP guidelines. The 1998 ICNIRP guidelines and IEEE C95.1-2005 are detailed in the following sections. 6.4.1 ICNIRP The most recent ICNIRP guidelines, approved in November 1997, were published in 1998 [13]. As in the case of the ANSI and IEEE committees, the ICNIRP guidelines are based on studies reporting established adverse health effects. In agreement with the rationale of C95.1-1991, ICNIRP also found that the relevant established effects are surface effects at the lower frequencies (e.g., electrostimulation, shocks and burns) and effects associated with tissue heating at the higher frequencies. Although a number of in vitro studies were reviewed, the focus was on in vivo studies. Epidemiological studies of reproductive outcome and cancer were reviewed but because of the lack of adequate exposure assessment and inconsistency of results, these studies were found to be of little use for establishing science-based exposure criteria. Studies reporting athermal effects, including ‘‘window effects’’ [e.g., effects associated with ELF amplitude modulated (AM) RF fields] were also considered, but ICNIRP concluded: ‘‘Overall, the literature on athermal effects of AM electromagnetic fields is so complex, the validity of reported effects so poorly established, and the relevance of the effects to human health is so uncertain, that it is impossible to use this body of information as a basis for setting limits on human exposure to these fields’’ [13]. The more recent review of the literature by IEEE led to the following conclusions regarding low-level effects: ‘‘Despite more than 50 years of RF research, low-level biological effects have not been established. No theoretical mechanism has been established that supports the existence of any effect characterized by trivial heating other than microwave hearing. Moreover, the relevance of reported low-level effects to health remains speculative and such effects are not useful for standard setting’’ [5, p. 82]. Standard-setting organizations (e.g., ANSI, IEEE) and organizations that develop recommendations and guidelines (e.g., NCRP and ICNIRP) have all determined that SAR is the appropriate dosimetric parameter over the broad whole-body resonance region and also found that the most reliable and sensitive indicator of potential harm was behavioral disruption, with a threshold SAR of 4 W/kg. A safety factor of 10 was incorporated for exposures in the workplace, and an additional factor of 5 for exposure of the general public yielding maximum whole-body-average SAR values of 0.4 and 0.08 W/kg, respectively (called basic restrictions). In addition, basic restrictions in terms of peak spatialaverage SAR of 10 and 2 W/kg averaged over any 10-g contiguous tissue are recommended for localized exposure. The ICNIRP peak spatial-average SAR values are based on the

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thresholds of cataract formation in rabbit eyes (about 10g) with safety factors of 10 and 50. The ICNIRP limits for high-peak, low-average-power pulsed fields are based on the evoked auditory response (microwave hearing [14, 15]) whereas the corresponding C95.1-1991 and C95.1-2005 limits are based on the stun-effect in small animals (with a suitable margin of safety) [16]. That is, while ICNIRP considers ‘‘microwave hearing’’ a harmful effect, it is not considered an adverse effect in the C95.1-2005 standard [5, pp. 81–82]. Table 6.1 shows the basic restrictions (SAR) of the ICNIRP guidelines for frequencies between 100 kHz to 10 GHz, both for occupational and for general-public exposure. Table 6.2 lists the derived limits (reference levels) for the incident fields. While compliance with the reference levels ensures that the basic restrictions are met, because of the conservatism built into the reference levels, exceeding the reference levels does not mean that the Table 6.1 1998 ICNIRP Basic Restrictions

Exposure Group

Frequency

Whole Body Avg. SAR W/kg

Occupational General Population

100 kHz to 10 GHz 100 kHz to 10 GHz

0.4 0.08

Local SAR (Head and Trunk) W/kg

Local SAR (Limbs) W/kg

10 (10g) 2 (10g)

20 (10g) 4 (10g)

Source: [13]. Table 6.2 1998 ICNIRP Reference Levels Frequency

E Field (V/m)

H Field (A/m)

Power Density (W/m 2 )

Occupational 3 to 65 kHz 0.065 to 1 MHz 1 to 10 MHz 10 to 400 MHz 400 to 2,000 MHz 2 to 300 GHz

610 610 610/f 61 3f 1/2 1.37

24.4 1.6/f 1.6/f 0.16 0.008f 1/2 0.36

10 f /40 50

General Population 3 to 150 kHz 0.15 to 1 MHz 1 to 10 MHz 10 to 400 MHz 400 to 2,000 MHz 2 to 300 GHz

87 87 87/f 1/2 28 1.375f 1/2 61

5 0.73/f 0.73/f 0.073 0.0037f 1/2 0.16

2 f /200 10

Source: [13].

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basic restrictions are exceeded. For additional details of ICNIRP recommendations, refer to the ICNIRP guidelines [13].

6.4.2 IEEE C95.1-2005 IEEE C95.1-2005 was approved on October 5, 2005, and published on April 19, 2006. The purpose of this standard is to provide recommendations to protect against established adverse effects to human health associated with exposure to RF electric, magnetic, and electromagnetic fields over the frequency range of 3 kHz to 300 GHz [5]. This revision (of C95.1-1991) is based on an evaluation of the scientific literature through 2003 (although the literature cutoff date was December 2003, several papers published in 2004 and 2005 were included), including those studies that involve low-level exposures where increases in temperature could not be measured or were not expected. New insights gained from improved experimental and numerical methods and a better understanding of the effects of acute and chronic RF electromagnetic field exposures of animals and humans are included. A lack of credible scientific and medical reports showing adverse health effects for RF exposures at or below corresponding exposure limits in past standards supports the protective nature of this standard. Above 100 kHz, the limits are designed to protect against adverse health effects resulting from tissue heating, the only established mechanism relating to adverse effects of exposure to RF energy at these frequencies. For the first time, guidance on the necessity of an RF exposure control program (e.g., recommendations in IEEE C95.7-2005 [17]) is included. The C95.1 standard consists of normative sections, including an overview of the document (scope, purpose, and introduction), references, definitions, and recommendations, as well as informative sections. The informative sections include seven annexes; the first three explain the revision process, summary of the literature, and rationale of the revision; the fourth provides examples of practical applications; and the last three annexes are glossary, literature database, and bibliography. Refer to the standard [5] for details, especially on the literature summary of about 1,300 peer-reviewed papers (Annex B) and the rationale (Annex C).

6.4.2.1 Recommendations The recommendations are expressed in terms of basic restrictions (BRs) and maximum permissible exposure (MPE) values (sometimes called reference levels or investigation levels). The BRs are limits on internal fields, SAR, and current density; the MPEs, derived from the BRs, are limits on external fields and on induced and contact currents. The recommendations are intended to apply to all human exposures except for exposure of patients by, or under the direction of, physicians and medical professionals.

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Basic Restrictions The whole-body-average BRs shown in Table 6.3 for frequencies between 100 kHz and 3 GHz protect against established adverse health effects associated with heating of the body during whole-body exposure. Consistent with the approach used in the prior standards and the ICNIRP guidelines, a traditional safety factor of 10 has been applied to the established SAR threshold of 4 W/kg for such effects, yielding an SAR of 0.4 W/kg averaged over the whole body. In the absence of an RF safety program, the BRs of the lower tier (action levels) may also be used for the general public. Applied to members of the general public, the lower tier provides more assurance that continuous, long-term exposure of all individuals in the population will be without risk of adverse effects. The BRs in terms of peak spatial-average SAR shown in Table 6.3 protect against excessive temperature rise in any part of the body that might result from localized or nonuniform exposure. As the frequency increases above 3 GHz, the power deposition becomes more superficial and SAR less meaningful. To account for the shallow penetration depth at the higher frequencies, the BRs are expressed in terms of incident power density and are identical to the derived limits (MPEs). Although exposure at or near these values may be accompanied by a slight sensation of warmth, this effect is not considered adverse. Maximum Permissible Exposure Values The derived limits (MPEs) in terms of equivalent power density, considered appropriate for all human exposure, are shown in Figure 6.2. (For detailed information on averaging time, refer to Table 6.4 and [5].) 6.4.2.2 RF Safety Programs Throughout the RF spectrum, the BRs and MPEs apply to exposure of people (i.e., compliance is determined by whether exposures of people to RF fields, currents, and Table 6.3 Basic Restrictions for Frequencies Between 100 kHz and 3 GHz

Whole-body exposure Localized exposure Localized exposure a

(Whole-body Average) (Local peak spatial-average) (Extremitiesb and pinnae)

Action Level SAR (W/kg)

Persons in Controlled Environments SAR (W/kg)

0.08 2a 4a

0.4 10a 20a

Averaged over any 10g of tissue (defined as a tissue volume in the shape of a cube). The extremities are the arms and legs distal from the elbows and knees, respectively. Source: [5].

b

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Figure 6.2 IEEE C95.1-2005 [5] MPEs for the upper and lower tiers in the frequency band 100 kHz to 300 GHz, as compared to reference levels in ICNIRP guidelines [13].

voltages exceed the applicable values). Where there may be access to RF fields, currents, and/or voltages that exceed the lower tier (action level) BRs and MPEs of IEEE C95.1-2005, an RF safety program such as detailed in IEEE Std C95.7-2005 [17] can be implemented to ensure that exposures do not exceed the MPEs or BRs for the upper tier (persons in a controlled environment). 6.4.3 Similarities and Differences Between the 1998 ICNIRP Guidelines and IEEE C95.1-2005 Table 6.4 compares various parameters of the 1998 ICNIRP guidelines with the corresponding parameters of C95.1-2005. This comparison indicates that while the two documents are similar, there are some differences between the two that suggests a need for continued harmonization efforts to achieve one global standard. 6.4.4 Regulations Based on Older Standards In the United States, the Federal Communications Commission (FCC), in 1996, adopted a combination of the IEEE C95.1-1991 and NCRP 1986 exposure criteria to regulate RF

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Table 6.4 Comparison of the 1998 ICNIRP Guidelines [13] with the IEEE C95.1-2005 Standard [5] over the Frequency Range Where the Predominant Interaction Mechanism Is Tissue Heating Parameter

ICNIRP

IEEE C95.1-2005

Frequency range Recognition of whole-body resonance Incorporation of dosimetry (SAR) Database of experimental literature Most significant biological endpoint

∼ 100 kHz to 300 GHz Yes

∼ 100 kHz to 300 GHz Yes

Yes

Yes

Large

Very large (∼ 1,300 citations)

Behavioral disruption (associated with ∼ 1°C core temperature rise) 1–4 W/kg

Behavioral disruption (associated with ∼ 1°C core temperature rise) ∼ 4 W/kg

0.4 W/kg (occupational) 0.08 W/kg (general public) 100 kHz to 10 GHz

0.4 W/kg (controlled environment) 0.08 W/kg (action level) 100 kHz to 3 GHz 10 W/kg (controlled environment) 2 W/kg (action level) 10g of tissue in the shape of a cube 6 minutes (controlled environments) 30 minutes (action level) 20 W/kg (extremities and pinnae) 4 W/kg (extremities and pinnae) 100 kHz < f ≤ 3 GHz 6 minutes ( f ≤ 3 GHz) then decreasing to 10 seconds at 300 GHz) 6 min (3 kHz ≤ f ≤ 1.34 MHz). E 2 and H 2 have different averaging times for 1.34 MHz < f ≤ 100 MHz but both are equal to 30 minutes at 100 MHz. For 100 MHz < f ≤ 5 GHz the averaging time is 30 minutes and then decreases to 10 seconds at 300 GHz. 90 mA (each foot) 45 mA (each foot) 100 kHz ≤ f ≤ 110 MHz

Whole-body-averaged SAR associated with behavioral disruption Limiting whole-body-averaged SAR —Applicable frequency range Peak spatial-average SAR (localized exposure) —Averaging volume —Averaging time

10 W/kg (occupational) 2 W/kg (general public) 10g of contiguous tissue 6 minutes (occupational) 6 minutes (general public)

Limits for extremities —Upper tier —Lower tier —Applicable frequency range Averaging time ( f > 100 kHz) —Upper tier —Lower tier

20 W/kg (limbs) 4 W/kg (limbs) 100 kHz < f ≤ 10 GHz

Induced and contact current limits —Upper tier —Lower tier —Applicable frequency range

6 minutes ( f ≤ 10 GHz) decreasing to 10 seconds at 300 GHz 6 minutes ( f ≤ 10 GHz) decreasing to 10 seconds at 300 GHz

40 mA (limb currents) 20 mA (limb current) 100 kHz ≤ f ≤ 110 MHz

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Table 6.4 (continued) Comparison of the 1998 ICNIRP Guidelines [13] with the IEEE C95.1-2005 Standard [5] over the Frequency Range Where the Predominant Interaction Mechanism Is Tissue Heating Parameter

ICNIRP

IEEE C95.1-2005

Special criterion for modulated fields Specific limits for high-peak, low-average-power pulses

No

No

Yes. Based on evoked auditory response (‘‘microwave hearing’’) Not specifically

Yes. Based on the stun-effect

RF safety program

Yes. IEEE C95.7-2005. The BRs and MPEs of the lower tier (action level) are linked to an RF safety program to mitigate against exposures that could exceed the BRs and MPEs of the upper tier

Source: [6].

exposures from transmitting equipment (including mobile communications) [18]. The basic restrictions for the whole body exposure is the same as those of ICNIRP and IEEE C95.1-2005, but the peak SAR is 1.6 and 8 W/kg averaged over any 1g of tissue for exposure in controlled environments (occupational exposure) and general-public exposure, respectively. The MPEs are shown in Table 6.5. Table 6.5 FCC Limits for Maximum Permissible Exposure (MPE) Frequency (MHz)

E Field (V/m)

H Field (A/m)

Power Density* (W/m 2 )

Exposure in Controlled Environments (Occupational) 0.3 to 3 614 1.63 3 to 30 1842/f 4.89/f 2 30 to 300 61.4/f 0.163 300 to 1,500 — — 1500 to 100,000 — —

1000 9000/f 2 10 f /30 50

Exposure in Uncontrolled Environments (General Population) 0.3 to 1.34 614 1.63 1.34 to 30 824/f 2.19/f 30 to 300 27.5 0.073 300 to 1,500 — — 1,500 to 100,000 — —

1000 1800/f 2 f /150 10

*Plane-wave equivalent power density. Source: [19].

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At the time this chapter was prepared, the FCC had not taken an action to issue a Notice of Proposed Rule Making for public comment in order to initiate the process for revising the above limits based on the new IEEE C95.1-2005 Standard or any other RF safety recommendations.

6.5 COMPLIANCE STANDARDS Several standards are used to ensure that various products (e.g., mobile phones and base stations) comply with contemporary safety standards, guidelines, and national regulations. For whole body exposures, compliance with the MPEs (reference levels) can be determined by measuring the incident fields using commercially available survey meters, or similar devices, following the protocols described in standards such as IEEE C95.3-2002 [20]. For near field exposures from specific devices, particularly handheld and portable devices, determination of the SAR is usually required. In 1996, IEEE Standards Coordinating Committee 34 (now IEEE ICES TC-34) began drafting a standard that specifies measurement protocols for certifying that mobile phones meet peak spatial-average SAR requirements. The result of this effort is IEEE 1528-2003 [7] and 1528a-2005 [21]. The European Committee for Electrotechnical Standardization (CENELEC) published a similar standard (EN50361) in July 2001 [22]. Because of minor differences between the two standards (e.g., the values for some tissue simulant parameters for certain frequencies), some products are required to be tested twice, once for the European and Australian markets and once for other parts of the world. This duplication emphasized the need for a globally harmonized test method. This issue was resolved with the publication of IEC 62209-1 [9] which is harmonized with IEEE 1528-2003 and replaces the CENELEC standard (EN50361).

6.5.1 Main Features of IEEE 1528-2003 (Including 1528a-2005) and IEC 62209-1 IEEE ICES TC-34 Subcommittee 2 and IEC TC-106 Project Team 62209 share common membership and goals for harmonized international standards for the measurement of peak SAR for mobile phones intended for use when placed next to the head. As a result, both the IEEE standards (1528-2003 and 1528a-2005) and IEC 62209-1 are technically identical. These standards specify measurement protocols designed to allow various laboratories to perform SAR measurements in a consistent manner. Included is a standard phantom model with specified size and shape, specific liquid compounds for tissue simulation, standard calibration techniques for E-field probes, and phone positioning requirements. Some of the salient features of the two standards are described below. For details of the SAR measurement protocol, the reader should refer to the original standards: IEEE 1528-2003 [7]; IEEE 1528a-2005 [21]; and IEC 62209-1 [9].

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SAR Measurement System Figure 6.3 shows a typical SAR measurement system used to perform SAR measurement in a head phantom. The system consists of an E-field probe, dc voltage amplifiers, highimpedance cables connecting the amplifier outputs to a personal computer, a controlling robot, a simulated tissue phantom, and a holder assembly for placing the phone with respect to the phantom. The robot holding the probe scans the entire exposed volume of the phantom in order to evaluate the three-dimensional field distribution. The entire system, including the E-field probe, is calibrated in a controlled laboratory environment in each tissue equivalent liquid at the appropriate operating frequency.

Figure 6.3 Mobile phone SAR measurement system showing robot-controlled electric field probe for SAR measurement in a head phantom exposed to a mobile phone at the left ear.

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Phantom Model To simulate the human head, both committees agreed to use the dielectric property values of Gabriel et al. [23]. Table 6.6 lists the dielectric properties of the equivalent head tissue at various frequencies based on a plane-wave analysis [24]. The dielectric constant is the average of all head tissues, and the conductivity is the larger of the 1g or 10g average calculated effective conductivity. Formulas for liquid head tissue phantoms for the various frequency bands are included in the IEEE and IEC standards. The dielectric constant and conductivity are required to be within 5% of the target values at the specified frequencies, excluding instrument error, and must be checked periodically to ensure compliance. The measured dielectric properties of the liquid at the device operating frequencies should be used in SAR calculations instead of the target values shown in Table 6.6. The U.S. Army head model (90% adult male head dimensions and shape) [25] was adopted by both committees from which a hairless specific anthropomorphic mannequin (SAM) of the head was constructed. Figure 6.4 shows the side view of SAM with ear structure and marking lines. The ear protrusion of the Army head model (28 mm) was reduced to 4 mm to simulate the compression of the ear during mobile phone use. This spacing brings the phone close to the head and provides results that are relevant to the exposure of the population with smaller ears, such as children. The SAM shell is made of fiberglass with a thickness no greater than 2 mm at the site of measurement, except at the ear. The relative dielectric constant of the shell is less than 5 and the conductivity less than 0.01 S/m. A 4-mm lossless spacer plus the 2-mm shell thickness at the ear canal is used to simulate the ear. All dimensions are specified in a CAD file. Right and left head models, obtained by bisecting the fiberglass SAM shell, are necessary because the asymmetric location of the antenna in many phones results in Table 6.6 Dielectric Properties of the Equivalent Head Tissue for Frequencies Between 300 and 3,000 MHz Frequency (MHz)

Relative Dielectric Constant (⑀ r )

Conductivity (␴ ) (S/m)

300 450 835 900 1,450 1,800 1,900 1,950 2,000 2,450 3,000

45.3 43.5 41.5 41.5 40.5 40.0 40.0 40.0 40.0 39.2 38.5

0.87 0.87 0.90 0.97 1.20 1.40 1.40 1.40 1.40 1.80 2.40

Source: [7, 9].

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Figure 6.4 Side view of the SAM phantom head showing the ear and marking lines BM and NF. Intersection RE is the ear reference point (ERP). (Source: [7].)

different SAR distributions on each side of the head. The models are filled with the correct liquid mixture simulating head tissue at the desired frequency. The liquid is 15 ± 0.5 cm in depth measured at the ear canal, which is approximately equivalent to the distance between the ears of the phantom. Measurement Procedures Figure 6.5 is a flow chart of the SAR testing protocol. E-field measurements are taken at a reference point where the fields are above the noise level (e.g., 10 mm above the ear reference point) to monitor power changes during the testing. This measurement is conducted after placing the mobile phone in operation with a fully charged battery. The inside surface of the SAM phantom immediately adjacent to the phone is scanned. If the peak occurs at the border of the area, the scan is repeated using an enlarged area when possible. A volumetric scan known as a zoom scan is then conducted at the location of the peak of the area scan, and the peak spatial-average SAR is calculated. These steps are repeated if the peak spatial-average SAR touches any side of the zoom scan volume. At the end of the zoom scan, the field is measured again at the initial power measurement reference

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Figure 6.5 Flow chart of SAR testing procedures. (Source: [9].)

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point. If the power has changed by more than 5%, it is recommended that the measurements be repeated. In Step 1, the above scan evaluation is conducted at the center frequency of the device under test at two positions on each side of the head with the antenna fully extended and with it retracted (if applicable), and all operational modes (i.e., AMPS, TDMA, CDMA, and so forth). The two positions are ‘‘cheek touch’’ and ‘‘15° tilt.’’ Figure 6.6 shows the two positions. The second position (15° tilt) is achieved by positioning the phone at the cheek touch position and then pivoting the device outwards by 15° with the top of the phone against the pinna. When positioning the phones against the head, the coordinates of two types of phones are as shown in Figure 6.7. Point A on the phone is positioned against the ear reference point on the SAM phantom, and the centerline of the phone (line AB) is lined up with the back-to-mouth (BM) line on the phantom. In Step 2, the same evaluation is repeated with the phone operating at a frequency at the high end of its frequency range and again at a frequency at the low end with the phone placed at the side of the phantom and the position that resulted in the largest peak spatial-average SAR. The last step is to examine the data and determine the maximum spatial peak SAR, which is the largest value found in Steps 1 and 2. The time needed for a complete test of a new multimode phone for compliance can be up to several weeks.

Figure 6.6 ‘‘Cheek’’ and ‘‘touch’’ positions of the mobile phone on the left side of the phantom. (Source: [7].)

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Figure 6.7 Handset vertical and horizontal reference lines and reference points A, B on two example device types. (Source: [7].)

6.5.2 Other Standards Related to Mobile Communication Many contemporary communication devices now use frequencies above 3 GHz, and therefore, IEEE ICES TC-34 is developing a second amendment to IEEE 1528 to extend the frequency range from 3 to 6 GHz. TC-34 is also developing standards for assessing compliance of mobile phones and radios using numerical techniques. These include a standard specifying the general requirements for using the finite difference time domain (FDTD) method for SAR calculations, a standard specifying the specific requirements for FDTD modeling of vehicle mounted antennas, and a standard for FDTD modeling of mobile phone exposure. Through a liaison arrangement with IEEE ICES, IEC Project Team 62209 is developing Part 2 of the 62209 standard for testing two-way radios, palmtop terminals, desktop terminals, body-worn devices including accessories, as well as multiple transmitters (30 MHz to 6 GHz). Another IEC Project Team 62232 is drafting a standard for characterizing the RF electromagnetic environment near base stations used for mobile radio communication.

6.6 DISCUSSION AND CONCLUSIONS Many government agencies throughout the world have adopted regulations that ensure devices used for mobile communications are safe. Such regulations are generally based on the basic restrictions and derived limits (reference levels, MPEs) found in consensus

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standards and guidelines that protect against adverse effects associated with exposure to RF energy. The collective credible evidence on which these standards and guidelines are based has not demonstrated that exposure to RF energy at levels at or below the basic restrictions and derived limits can affect biological systems in a manner that might lead to, or augment, any health effect. Moreover, both the ICNIRP guidelines and IEEE standards are living documents—once a standard is approved, work begins on assessing the evolving database of relevant scientific literature to ensure that the limits continue to be valid (i.e., the surveillance and evaluation of the RF bioeffects literature is continuous). If any new adverse effect is established that would require a change in the current limits, the standard can be promptly revised or amended to reflect these changes. Although whole-body-averaged and peak spatial-average SAR have been the accepted dosimetric quantities for almost three decades, replacing the latter with temperature increase was discussed during development of the 2005 IEEE standard and is being explored as a possibility for the next revision. The rationale for using temperature rather than peak SAR is based on the literature showing that adverse health effects of RF exposure are associated with significant temperature increases in the body. Dosimetry studies are now in progress to identify the relationship between temperature rise and peak spatial-average SAR for future consideration. In order to ensure that such devices comply with the safety standards and guidelines, compliance standards have been developed by international committees (e.g., IEEE ICES, IEC, and CENELEC). These product/compliance standards identify specific protocols to ensure that test methods used throughout the world are consistent. One set of such standards specifies uniform SAR test methods, which are utilized by mobile phone manufacturers to demonstrate that their products comply with the requirements of the safety standards. The authors have participated in the development of both the safety (exposure) and compliance (measurement) standards. During committee deliberations that led to IEEE C95.1-2005, the focus was on conservatism; during deliberations on the compliance standards, the focus was on precision. Worst-case assumptions were always considered. While it is always a good practice to make precise and accurate measurements, there is a trade-off when assessing compliance of a device with limits having large built-in safety margins. That is, whether or not a product meets a specified limit is a compliance issue— not a safety issue. An unrealistic focus on precision causes one to lose sight of the objective (i.e., ‘‘can’t see the forest for the trees’’). The objective should be agreement on a realistic compliance method and international harmonization. Harmonized standards provide triple wins to all involved. First, consumers gain the protection of an internationally recognized safety standard and have equal access to products and services that are available to consumers elsewhere in the world. Second, regulators gain the framework for a consistent approach to regulation that is in agreement with the recommendations of the WHO, the ITU, and the WTO. Third, industry gains by developing and manufacturing products to widely accepted international standards and, once tested for compliance, can make those products available around the world in a consistent and timely manner.

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REFERENCES [1] NCRP, ‘‘Radiofrequency Electromagnetic Fields—Properties, Quantities and Units, Biophysical Interaction, and Measurements,’’ NCRP Report No. 67, National Council on Radiation Protection and Measurements, Bethesda, MD, 1981. [2] Chou, C. K., et al., ‘‘Radio Frequency Electromagnetic Exposure: A Tutorial Review on Experimental Dosimetry,’’ Bioelectromagnetics, Vol. 17, 1996, pp. 195–208. [3] Guy, A. W., ‘‘Dosimetry Associated with Exposure to Non-Ionizing Radiation: Very Low Frequency to Microwaves,’’ Health Phys., Vol. 53, 1987, pp. 569–584. [4] USAS C95.1-1966, Safety Level of Electromagnetic Radiation with Respect to Personnel, United States of America Standards Institute. [5] IEEE C95.1-2005, ‘‘IEEE Standard for Safety Levels with Respect to Human Exposure to Radio Frequency Electromagnetic Fields, 3 kHz to 300 GHz.’’ [6] Petersen, R. C., ‘‘Radiofrequency/Microwave Safety Standards,’’ in RF Dosimetry Handbook, P. Chadwick, (project leader), 2007, Chapter 6, available at http://www.emfdosimetry.org/petersen/Radiofrequency_ Safety_Standards.html. [7] IEEE 1528-2003, ‘‘IEEE Recommended Practice for Determining the Peak Spatial-Average Specific Absorption Rate (SAR) in the Human Head from Wireless Communications Devices: Measurement Techniques.’’ [8] Osepchuk, J. M., and R. C. Petersen, ‘‘Safety and Environmental Issues,’’ in The RF and Microwave Handbook, M. Golio, (ed.), Boca Raton, FL: CRC Press LLC, 2007. [9] IEC 62209-1, ‘‘Human Exposure to Radio Frequency Fields from Hand-Held and Body-Mounted Wireless Communication Devices—Human Models, Instrumentation, and Procedures—Part 1: Procedure to Determine the Specific Absorption Rate (SAR) for Hand-Held Devices used in Close Proximity to the Ear (Frequency Range of 300 MHz to 3 GHz),’’ International Electrotechnical Commission, Geneva, 2005. [10] ANSI C95.1-1982, ‘‘American National Standard Safety Levels with Respect to Human Exposure to Radio Frequency Electromagnetic Fields, 300 kHz to 100 GHz.’’ [11] NCRP, ‘‘Biological Effects and Exposure Criteria for Radiofrequency Electromagnetic Fields,’’ Report 86, National Council on Radiation Protection and Measurements, Bethesda, MD, 1986. [12] IEEE C95.1-1991, ‘‘IEEE Standard for Safety Levels with Respect to Human Exposure to Radio Frequency Electromagnetic Fields, 3 kHz to 300 GHz.’’ [13] ICNIRP (International Commission on Non-Ionizing Radiation Protection), ‘‘Guidelines for Limiting Exposure to Time-Varying Electric, Magnetic, and Electromagnetic Fields (Up to 300 GHz),’’ Health Physics, Vol. 74, 1998, pp. 494–522. [14] Chou, C. K., A. W. Guy, and R. Galambos, ‘‘Auditory Perception of Radio-Frequency Electromagnetic Fields (80th Review and Tutorial Paper),’’ J. of Acoustic Society of America, Vol. 71, No. 6, 1982, pp. 1321–1334. [15] Elder, J. A., and C. K. Chou, ‘‘Auditory Response to Pulsed Radiofrequency Energy,’’ Bioelectromagnetics, Vol. 24, Supplement 6, 2003, pp. S162–S173. [16] Guy, A. W., and C. K. Chou, ‘‘Effects of High-Intensity Microwave Pulse Exposure on Rat Brain,’’ Rad. Sci., Vol. 17, No. 5S, 1982, pp. 169S–178S. [17] IEEE C95.7-2005, ‘‘Recommended Practice for Radio Frequency Safety Programs.’’ [18] Federal Communication Commission, 47 CFR Parts 1, 2, 15, 24 and 97, ‘‘Guidelines for Evaluating the Environmental Effects of Radiofrequency Radiation,’’ August 6, 1996. [19] Federal Communications Commission, Office of Engineering and Technology, OET Bulletin 65, Edition 97-01, Washington, D.C., August 1997. [20] IEEE C95.3-2002, ‘‘Recommended Practice for Measurements and Computations of Radio Frequency Electromagnetic Fields with Respect to Human Exposure to Such Fields, 100 kHz–300 GHz.’’

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[21] IEEE 1528a-2005, ‘‘IEEE Recommended Practice for Determining the Peak Spatial-Average Specific Absorption Rate (SAR) in the Human Head from Wireless Communications Devices: Measurement Techniques—Amendment 1: CAD File for Human Head Model (SAM Phantom).’’ [22] EN50361-2001, ‘‘Basic Standard for the Measurement of Specific Absorption Rate Related to Human Exposure to Electromagnetic Fields from Mobile Phones (300 MHz–3 GHz),’’ European Committee for Electrotechnical Standardisation, (CENELEC), Brussels. [23] Gabriel, S., R. W. Lau, and C. Gabriel, ‘‘The Dielectric Properties of Biological Tissues: 2. Measurement in the Frequency Range 10 Hz to 20 GHz,’’ Phys. Med. Biol., Vol. 41, No. 11, 1996, pp. 2251–2269. [24] Drossos, A., V. Santomaa, and N. Kuster, ‘‘The Dependence of Electromagnetic Energy Absorption Upon Human Head Tissue Composition in the Frequency Range of 300–3000 MHz,’’ IEEE Trans. on Microwave Theory and Techniques, Vol. 48, No. 11, 2000, pp. 1988–1995. [25] Gordon, C. C., et al., ‘‘1988 Anthropometric Survey of U.S. Army Personnel: Methods and Summary Statistics,’’ Technical Report NATICK/TR-89/044, U.S. Army Natick Research, Development and Engineering Center, Natick, MA, September 1989.

Chapter 7 Applications of Modern EM Computational Techniques: Antennas and Humans in Personal Communications Yahya Rahmat-Samii, K.W. Kim, M. Jensen, and Kyohei Fujimoto

7.1 INTRODUCTION Recent market evaluations and projections point to continuing growth in the popularity of personal terrestrial and satellite communications systems. Among the diversified components involved in the operation of these systems, handset units are perhaps the most visible part of the system that must be designed to satisfy user needs. Design issues range from the aesthetic look of the unit to health considerations and antenna performance. Depending on the user’s sophistication and operational needs, it is projected that a wide variety of designs will be used. Antennas play a paramount role in optimal design of wireless personal communication devices. Current evolution of wireless personal communications has necessitated a comprehensive understanding of electromagnetic (EM) interactions between handset antennas and the nearby human body. These human–antenna interactions influence the electromagnetic performance of the antenna by altering antenna input impedance, modifying the antenna radiation patterns and polarization state, absorbing antenna-delivered power, and so on. Also questions concerning health hazards have necessitated a more thorough evaluation and characterization of the specific absorption rate (SAR) in the human tissue. Recently, significant progress in understanding human–antenna interactions has been achieved due to advanced computational techniques as well as accurate near343

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field measurements. The reader is referred to [1–4] for a review of recent developments and a collection of many pertinent references that are not duplicated here. Popular numerical techniques currently used for EM interaction computations are the finite-difference time-domain (FDTD) method [1], the method of moments (MoM) [5], the eigenfunction expansion method (EEM) [6], and their hybridization as shown in Figure 7.1. Among them, the most popular technique is the FDTD method because of its computational flexibility in modeling complex antenna geometries and the nearby biological tissues. For example, the radiation and absorption characteristics of various antenna configurations—such as the monopole, side-mounted dual PIFA (planar inverted F antenna), and back-mounted PIFA (see Figure 7.2)—in the presence of the human head have been presented [1]. In this reference an anatomical human head is modeled based on actual magnetic resonance image (MRI) scans. In addition, significant work has been performed to evaluate power absorption in biological tissues using various antenna configurations [1–4]. Nevertheless, in certain situations the use of FDTD may not be practical and may not allow generation of parametric engineering data. In these situations, the EEM has

Figure 7.1 Various numerical approaches to study EM interactions between human and antennas.

Figure 7.2 Various antenna configurations on typical handsets (∼105 cc): (a) monopole; (b) side-mounted dual PIFA; and (c) back-mounted PIFA. Frequency of operation is 915 MHz.

345

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been used to efficiently evaluate the EM interaction between various antennas and the human head [6]. The EEM is based on the exact scattering solution of infinitesimal dipoles in the presence of a multilayered, lossy dielectric sphere. The EEM has facilitated systematic and parametric studies of EM interactions, resulting in useful engineering curves for antenna design. The results from the EEM can also be used to validate results computed by other numerical techniques. In wireless communication systems, the ideal antenna for the handheld unit will be low profile and unobtrusive to the user. In principle, it must provide a good impedance match and offer radiation characteristics, such as pattern and polarization, that meet the requirements of the particular communications system. Difficulties in achieving these objectives arise from the presence of an operator whose biological tissues perturb the antenna’s electromagnetic properties [7–10]. Investigations of these effects lend insight into new antenna designs that are less susceptible to human tissue influence. The practical considerations for the design of handset antennas are as follows: (1) impedance match and bandwidth, (2) biological tissue effect on the input impedance and radiation pattern, (3) antenna gain, (4) antenna polarization, (5) power absorption in human biological tissue, and (6) antenna diversity performance to minimize signal fading. This chapter focuses on the biological tissue effects on the input impedance and radiation patterns, the power absorption in the human biological tissue, and the role of antenna diversity. The results presented have been obtained using both the FDTD and EEM. After a brief review of both methods, representative calculated results are discussed and many engineering-oriented design charts and tables are presented. Some of the major observations developed in this chapter can be summarized as follows: 1. The casual manner in which users hold their handset units impacts the antenna design. 2. The proximity of the head and hand can considerably influence the radiation patterns, polarization states, impedance match, and efficiency of handset antennas. 3. On average, between 30% and 60% of the total power radiated by the antenna is absorbed in the biological tissues. 4. Penetration depth within the tissue strongly depends on the frequency. 5. An important factor in determining power absorption is the proximity of the ‘‘hot’’ current spots near the head and hand, an observation which suggests that novel antenna designs may help reduce the interaction. 6. The health hazard aspects of handset units are not yet fully understood, and research and data collection efforts are ongoing. 7. As lower powered units are introduced, the peak average SAR levels are reduced. 8. Even a moderately larger separation between the antenna and tissues can reduce SAR considerably. 9. Internal integrated antennas can become useful if they are kept away from the hand and head. 10. Diversity and directive antennas can play an important role in certain applications.

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11. When proper numerical models are used, most of the existing computational techniques predict similar results within variations on the order of 10%. 12. Antenna–human interactions must be evaluated in various configurations in order to obtain statistical averages of the overall performance for system link evaluations. 13. Elaborate measurement techniques have been developed for antenna–tissue interaction characterization. 7.2 DEFINITION OF DESIGN PARAMETERS FOR HANDSET ANTENNAS In this section, definitions of the important design parameters—specific absorption rate (SAR), antenna gain, input impedance, and so on—for handset antennas are briefly reviewed. 7.2.1 Absorbed Power and Specific Absorption Rate If the handset antenna is located in the vicinity of human biological tissue, some portion of the antenna delivered power is absorbed in the tissue. The total power absorbed in the lossy tissues can be defined as Pabs =

冕

1 ␴ | E |2 dV 2

(7.1)

V

where ␴ is the conductivity of the tissue, E is the electric field intensity, and V is the volume of the biological tissue. The radiated power to the far-field region is obtained as 1 Prad = Re 2

再冕

E × H* ⭈ nˆ dS

S

冎

(7.2)

where E and H are the electric and magnetic field intensities on a surface S completely enclosing the antenna and tissue, and nˆ is the outward unit vector normal to the surface. The efficiency of the antenna/tissue system can be defined as

␩a =

Prad P = rad Prad + Pabs Pdel

(7.3)

where Pdel is the antenna delivered power. The total antenna efficiency ␩ 0 is the product of the reflection efficiency (1 − | ⌫ |2 ), conduction efficiency, dielectric efficiency, and antenna–tissue efficiency.

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Specific absorption rate (SAR) is one of the most widely used parameters when discussing the health risk associated with electromagnetic power absorption. It is defined as SAR =

␴ | E |2 2␳

(7.4)

where ␳ is the material density. The ANSI/IEEE standard C95.1-1992 RF Safety Guideline [11] suggests that the 1-g averaged peak SAR should not exceed 1.6 W/kg and the wholebody averaged peak SAR should be less than 0.08 W/kg. These guidelines are applicable to uncontrolled situations and therefore must be satisfied for personal handsets. 7.2.2 Directivity and Gain The directivity of an antenna, D, is the ratio of the radiated power density p (␪ , ␾ ) to the radiated power density averaged in all directions, or D=

冕冕

(1/4␲ r 2 )

p (␪ , ␾ )

(7.5)

p (␪ ′, ␾ ′ )r 2 sin ␪ ′d␪ ′d␾ ′

The gain of an antenna, G, takes into account the total efficiency ␩ 0 of the antenna as well as the directivity. It is defined as G=

4␲ r 2p (␪ , ␾ ) = ␩0D Pin

(7.6)

7.2.3 Antenna Impedance and S 11 The antenna input impedance is defined as the ratio of the voltage (V ) to the current (I ) at the antenna feed point; that is, Z in =

V I

|

at antenna feed point

(7.7)

S 11 of the antenna is the reflection coefficient of the antenna at the feed point and is defined as S 11 =

Z in − Z 0 Z in + Z 0

where Z 0 is the characteristic impedance of the feeding transmission line.

(7.8)

349

7.3 FINITE-DIFFERENCE TIME-DOMAIN FORMULATION The FDTD algorithm can be derived from Maxwell’s time-domain integral equations given as

冕

冖

S

C

d ⑀ (r)E(r, t ) ⭈ dS = dt

H(r, t ) ⭈ dl −

冕

␴ (r)E(r, t ) ⭈ dS

(7.9)

S

冕

冖

S

C

d ␮ (r)H(r, t ) × dS = − E(r, t ) × dl dt

(7.10)

where E and H are the electric and magnetic field intensities and ⑀ , ␮ , and ␴ are the permittivity, permeability, and conductivity, respectively. By placing the field components on the ‘‘Yee’’ unit cell as shown in Figure 7.3 [12], these equations can be discretized to provide second-order accuracy [13]. For example, z components of the E and H fields are obtained from (7.9) and (7.10) as n+1 E z ,i , j , k

=

n ␣ z, i , j ,k E z, i, j , k

+ ␤ z, i , j ,k

冋

n + 1/2

n + 1/2

H y , i + 1, j, k − H y ,i , j ,k ⌬x

n + 1/2

−

n + 1/2

H x , i , j + 1, k − H x ,i , j , k ⌬y

册

(7.11) n + 1/2 H z, i, j, k

=

n − 1/2 H z, i, j , k

+ ␥ z, i , j ,k

冋

n

n

E x , i , j, k − E x , i, j − 1, k ⌬y

n

−

n

E y ,i , j , k − E y, i − 1, j ,k ⌬x

册

(7.12)

where

⑀ p , i , j, k ␣ p ,i , j , k =

⌬t

⑀ p , i , j ,k ⌬t

− +

␴ p , i, j, k 2

␴ p , i , j, k

, ␤ p , i, j, k =

1

⑀ p, i , j ,k ⌬t

2

+

␴ p, i , j ,k

, ␥ i, j, k =

⌬t

␮ p ,i , j ,k

2

(7.13)

and

⑀ p , i, j, k =

1 ⌬S

冕

⌬S

⑀ (r)nˆ ⭈ dS, ␴ p ,i , j , k =

1 ⌬S

冕

⌬S

␴ (r)nˆ ⭈ dS, ␮ p , i , j, k =

1 ⌬S

冕

⌬S

␮ (r)nˆ ⭈ dS (7.14)

Figure 7.3 Location of the electric and magnetic field vectors for the FDTD methodology: (a) on a cubic unit cell (Yee cell); and (b) on one face of unit cell.

350

351

where ⌬S represents the area of the cell face normal to the unit vector nˆ. Other field components can be obtained in a similar fashion [13]. In the FDTD calculation, all field values are initially set to zero. A source is introduced by setting a voltage at the antenna feed point and computing from it the source electric field [14]. The electric and magnetic field values are then calculated using a leapfrog scheme: The magnetic field is first calculated at a time t = n + 1/2, and the electric field is subsequently calculated at t = n + 1. Continuity of fields is naturally enforced at a dielectric boundary in this algorithm. However, at the conductor interface, the tangential E-field must be explicitly set to zero. At the outer boundary of the computation, Mur’s absorbing boundary conditions (second order) [15] or perfectly matched layer (PML) boundary conditions [16] are typically applied. Once the time-domain calculation is complete, frequency-domain quantities such as input impedance, patterns, and gain can be obtained using the fast Fourier transform (FFT). By carefully choosing the frequency content of the time-domain excitation, this approach can be used to obtain the antenna response over a broad frequency band. 7.4 EIGENFUNCTION EXPANSION METHOD 7.4.1 EEM Implementation The eigenfunction expansion method (EEM), which uses the dyadic Green’s function, is based on the exact scattering solution of infinitesimal dipoles. The scattered field is expressed as a series expansion of the spherical vector wave functions. Figure 7.4 illustrates the geometry of a multilayered sphere consisting of N regions with the corresponding

Figure 7.4 Geometry for EEM: A multilayered, lossy dielectric sphere with (N − 1) layers; N discrete regions can be defined to describe the scattered field in each region; a p is the radius of each layer in the concentric sphere.

352

constitutive parameters. For example, infinitesimal dipoles are assumed to be oriented in the x direction and located outside of the multilayered sphere. The total electric field can be expressed using the dyadic Green’s function [17] as E(r) = −j␻␮ 1

冕冕冕

G e( p 1) (r, r ′ ) ⭈ J(r ′ )dv ′

(7.15)

where G e( p 1) (r, r ′ ) is the dyadic Green’s function for each region and ␮ 1 is the permeability for region 1. In the superscript of G e(p 1) (r, r ′ ), p denotes for the p th region and 1 is used for the source region. For an infinitesimal electric dipole located at r ′ = (r 0 , ␪ 0 , ␾ 0 ) with a current moment c = Il xˆ , the electric field can be obtained from (7.15) as E(r) = −j␻␮ 1 Il ⭈ G e( p 1) (r, r ′ ) ⭈ xˆ

(7.16)

Using the dyadic Green’s function for the scattered field in region p (a p ≤ r ≤ a p − 1 ) where p = 1, 2, . . . , N, the scattered electric field can be obtained as E s( p ) (r) = −

⭈

冉

␻␮ 1 Il ⭈ ␤1 4␲

冉

冦冉

冊∑ ∑ ∞

n

n=1 m=0

(2 − ␦ m0 )

(2n + 1) (n − m )! n (n + 1) (n + m )!

(4)

(1)

o

o

A n , p M e mn ( ␤ p ) + C n, p M e mn ( ␤ p ) (4)

(1)

o

o

冊冋 冊冋

+ B n , p N e mn ( ␤ p ) + D n , p N e mn ( ␤ p )

(4)′

M e mn ( ␤ 1 ) ⭈ xˆ o

(4)′

(7.17)

册 册冧

N e mn ( ␤ 1 ) ⭈ xˆ o

(i)

(i )

o

o

where ␤ p = ␻ √␮ p ⑀ p and ␦ m0 is the Kronecker delta function. M e mn and N e mn are the even or odd spherical vector wave functions, which are solutions of the source-free vector wave equation ⵜ × ⵜ × E − ␤ 2E = 0 [17]. A n , N = B n ,N = 0 in the above equation to avoid infinite field at the origin. The expansion coefficients A n ,p , B n ,p , C n ,p , and D n , p in (7.17) are obtained by applying boundary conditions to the tangential electric and magnetic fields at the dielectric interfaces (r = a p ). 7.4.2 Hybridization of the EEM and MoM To accurately account for EM interactions between antennas and the multilayered sphere, the hybridization of the EEM and the method of moments (MoM) has been performed. Using this hybridization, the current distribution on the antenna can be determined efficiently. The unknowns are limited only to the surface of the antenna since the scattered dyadic Green’s function is provided by EEM.

353

The electric field integral equation (EFIE) for any antenna in the presence of a multilayered dielectric sphere can be written as E(r) = Ei(r) + Es(r) = Ei(r) − j␻␮

冕冕冕

G e(11) (r, r ′ ) ⭈ J(r ′ )dv ′

nˆ × E(r) = 0 on the surface of the antenna

(7.18a) (7.18b)

where Ei(r) is the incident field due to the localized source (modeled as a delta gap or a magnetic frill), Es(r) is the scattered field due to the induced current on the antenna, and G e(11) (r, r ′ ) is the dyadic Green’s function for the region outside the sphere (region 1). In this treatment, a thin wire dipole antenna oriented in the x -direction is considered. On the surface of the thin wire, tangential components of the electric field vanish such that E x = E ix + E sx = 0

(7.19)

The tangential field can approximately be evaluated along the center of the thin wire, while the induced current is confined to the surface of the wire (thin wire approximation). Then, one obtains −E xi = E sx = E xs, fs + E xs ,sob = −j␻␮

冕冕冕 冕冕冕

− j␻␮

xˆ ⭈ G e 0 (r, r ′ ) ⭈ xˆ Jx (r ′ )dv ′

(7.20)

(11) xˆ ⭈ G es (r, r ′ ) ⭈ xˆ Jx (r ′ )dv ′

where E xs , fs is the field due to induced current through free-space propagation, E xs, sob is the field due to scattering from the multilayered sphere, G e 0 (r, r ′ ) is the free-space dyadic (11) Green’s function, and G es (r, r ′ ) is the dyadic Green’s function for the scattered field that is provided by EEM. The above equation can be solved for the unknown current coefficients on the surface of the antenna by applying the MoM with pulse basis functions and point matching weighting functions. 7.5 RESULTS USING EEM 7.5.1 Human Head Model A six-layered, lossy dielectric sphere is used to simulate the biological tissues of a human head. Identified biological tissues include skin, fat, bone, dura, cerebrospinal fluid (CSF),

354

and brain. The electrical parameters (permittivity and conductivity) of the biological tissues at 900 MHz and 1.9 GHz are shown in Table 7.1. The electrical parameters of the tissues at 900 MHz are taken from [10], and at 1.9 GHz from [8]. These parameters can also be obtained using the multiple Cole-Cole dispersion equation and the corresponding parameters in [18]. 7.5.2 EM Interaction Characterizations First, we assume that a ␭ /2 antenna is located 2 cm away from the surface of the spherical head at 900 MHz. The gain patterns are shown in Figure 7.5(a). As can be seen in the figure, the far-field patterns are significantly modified by the presence of the spherical head. In the x-z plane, the peak gain with the spherical head is slightly enhanced (∼0.28 dB) in the ␪ = 0 direction, but is reduced by 2.28 dB toward the spherical head direction as compared with the free-space antenna gain. The unaveraged and 1-g averaged SAR distributions inside the spherical head along the z -axis are shown in Figure 7.5(b). With the unaveraged SAR distribution, the main peak SAR occurs at the cerebrospinal fluid (CSF) layer just outside of the brain region; the second peak occurs at the skin layer. With the averaged SAR distribution (averaging over a 1.1 × 1.1 × 1.1 cm3 cubical volume with eleven 1-mm cells each side of the cube), the peak values are much lower than those of the unaveraged SAR. For example, with 1W of delivered power, the peak values are 4.5 W/kg with the 1-g averaged SAR near the CSF layer and 11.2 W/kg with the unaveraged SAR. Also, we observe that the peak SAR locations are shifted, due to the averaging process, from those of the unaveraged SAR. The SAR peaks at the biological layers, which possess high conductivity such as the skin and CSF layers. Three-dimensional surface plots of the SAR distribution are shown in Figures 7.6(a) and (b) in the x-z plane and in the y-z plane, respectively. As can be seen in the figures, peak SAR lies along the z -axis. In the figures, it is interesting to observe small ripples Table 7.1 Electrical Parameters of a Six-Layered Head Model at 900 MHz and 1.9 GHz* Biological Tissues Skin Fat Bone Dura CSF Brain

900 MHz

1.9 GHz

Radius ap (cm)

⑀r

␴

⑀r

␴

Mass Density (10 3 kg/m 3 )

9.00 8.90 8.76 8.35 8.30 8.10

40.7 10.0 20.9 40.7 79.1 41.1

0.65 0.17 0.33 0.65 2.14 0.86

37.21 9.38 16.40 37.21 77.30 43.22

1.25 0.26 0.45 1.25 2.55 1.29

1.01 0.92 1.81 1.01 1.01 1.04

*a p is the radius, ⑀ r the permittivity, and ␴ the conductivity of each spherical layer.

355

Figure 7.5 (a) Far-field gain patterns of a ␭ /2 antenna with and without a six-layered spherical head. (b) SAR distribution—1-g averaged and unaveraged—along the z -axis. Operating frequency of antenna is 900 MHz. Abbreviations in part (b): BR (brain), C (CSF), D (dura), BN (bone), F (fat), and S (skin).

near the center of the brain region due to the focusing effect of the sphere. These ripples may be more conspicuous when the head–antenna separation distance is increased. Contour plots of the electric field intensity are shown in Figures 7.6(c) and (d). The numbers in the contour plots designate contour lines for 20 log10 | E |. As another example, EM interaction results are obtained at an operating frequency of 1.9 GHz. Figure 7.7(a) shows the gain patterns. As compared with the free-space antenna gain in the x-z plane, the peak field intensity for the spherical head is enhanced by 2.21 dB in the ␪ = 0 direction, but is reduced by 8.27 dB toward the spherical head

356

Figure 7.5 (continued).

direction. The SAR distribution along the z -axis is shown in Figure 7.7(b). The magnitudes of both 1-g averaged and unaveraged SAR peaks near or at the skin layer and CSF region are greater than those of the 900-MHz case [see Figure 7.5(b) and Table 7.2]. The reason may be the shorter ␭ /2 antenna length (7.9 cm) at 1.9 GHz as compared with that (16.7 cm) at 900 MHz. Also, the SAR value at the skin layer is greater than that at the CSF layer, which implies the importance of the skin layer at higher frequencies. With 1-g averaged SAR, the main SAR peak is near the CSF layer rather than near the skin layer. The total power absorption in the spherical head at 900 MHz is plotted as function of the head–antenna separation distance and the head size in Figure 7.8(a). As shown in the figure, the power absorption is strongly dependent on the head–antenna separation distance (d ). A significant portion of the antenna delivered power is absorbed in the spherical head (40% to 50% at d = 2 cm) when the head is located near the antenna, but becomes smaller at large distances (≤10% at d = 10 cm). In this case, it is observed that the absorbed power decays roughly as d −0.75 at close distances from the sphere. Also, the total power absorption variation in the spherical head at 1.9 GHz is shown in Figure 7.8(b). As compared with the 900-MHz case, the power absorption drops more rapidly when the antenna separation distance is small; that is, Pa ⬀ d −0.85. It is observed that the size of the head does not strongly influence the total power absorption at this frequency. The results of the EM interactions between half-wave antennas and a six-layered spherical head at 900 MHz and 1.9 GHz are summarized in Table 7.2.

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Figure 7.6 3-D surface plots of SAR distributions inside the spherical head (a) in the x-z plane and (b) in the y-z plane; and contour plots of the electric field distributions (log scale) (c) in the x-z plane and (d) in the y-z plane. Operating frequency of the ␭ /2 antenna is 900 MHz. The numbers in the contour plots designates contour lines for 20 log10 ( | E | ).

358

Figure 7.6 (continued).

7.5.3 Effects of Size of the Head Model: Adult and Child Head sizes vary widely among individuals. Recently, for a smaller size head (such as a child’s head), higher and larger in-depth penetration of the absorbed energy has been reported [8]. To investigate this phenomenon, a ␭ /2 dipole antenna at 900 MHz is assumed to be located 2 cm away from the spherical head. Also, it is assumed that the radius of

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Figure 7.7 (a) Far-field gain patterns of a ␭ /2 antenna with and without a six-layered spherical head. (b) SAR distribution—1-g averaged and unaveraged—along the z -axis. Operating frequency of antenna is 1.9 GHz. Abbreviations in part (b): BR (brain), C (CSF), D (dura), BN (bone), F (fat), S (Skin).

an adult head is 10 cm, and that of a child is 7 cm with a six-layered head model as described in Table 7.1. Figure 7.9(a) shows the comparison of SAR distributions along the z -axis. Clearly, the SAR in the interior region of the child head is greater even though the SAR at the outer surface of the brain region is roughly the same as that of the adult head. This is because the size of the child’s brain region is smaller than that of the adult head. Contour plots of the electric field distribution in the x-z plane are shown in Figures 7.9(b) and (c). As can be seen in the figures, the electric field penetrates more deeply into the child head than the adult head. However, according to Figure 7.8(a), the total

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Figure 7.7 (continued). Table 7.2 Summary of EM Interactions Between a Half-Wave Dipole Antenna and a Six-Layered Spherical Head at 900 MHz and 1.9 GHz*

␭/2 Antenna Power absorption (%) Peak SAR (W/kg) Unaveraged 1-g Averaged Gain (dBi) Directivity (dBi) Input impedance (⍀) without head with head

900 MHz

1.9 GHz

43.9%

35.3%

11.2 (at CSF layer) 4.5 (near CSF layer) 2.54 4.88

27.6 (at skin layer) 6.9 (near CSF layer) 4.46 6.32

99.9 + j 51.9 128.0 + j 51.9

99.6 + j 51.9 104.0 + j 30.0

*The antenna-delivered power is 1W. Antenna is located 2 cm away from the surface of the spherical head.

absorbed power with the child head (38.4%) is smaller than that of the adult head (47.6%) at d = 2 cm. 7.5.4 Comparison Between Homogeneous and Multilayered Spheres The differences in SAR distributions between the homogeneous and multilayered spherical heads are discussed in this section. For a homogeneous head, the electrical parameters of brain in Table 7.1 are used. Figure 7.10 compares 1-g averaged SAR distributions between

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Figure 7.8 Power absorption in the six-layered spherical head as function of the head–antenna separation distance and the size of the head at (a) 900 MHz and (b) 1.9 GHz. The power absorption drops roughly as d −0.75 at 900 MHz and d −0.85 at 1.9 GHz for small d.

the six-layered and homogeneous spheres together with unaveraged SAR distributions at 900 MHz. As can be seen, near the surface of the spherical head, the 1-g averaged SAR with the homogeneous sphere is much larger than that with the six-layered spherical head. Since SAR at the fat or bone layer with the six-layered spherical head is much smaller than that of the homogeneous sphere, the 1-g averaged SAR is lower near the skin layer. Therefore, the homogeneous sphere overestimates the SAR values near the surface of the spherical head. 7.5.5 Vertical Location of Antennas Because the vertical handset position will differ for different users, it is interesting to assess the influence of antenna vertical location on the observed interaction. Gain patterns

362

Figure 7.8 (continued).

Figure 7.9 (a) Comparison of SAR distributions in an adult head and a child head. Contour plots of the electric field (b) in an adult head and (c) in a child head. Operating frequency is 900 MHz. The numbers in the contour plots designate contour lines for 20 log10 ( | E | ).

363

Figure 7.9 (continued).

364

Figure 7.10 Comparison of 1-g averaged SAR distributions between multilayered and homogeneous spherical heads at 900 MHz. The antenna delivered power is 1W.

at 900 MHz without and with the head in the horizontal plane (y-z plane) are shown in Figure 7.11(a). In the figure, the gain patterns are obtained as the distance (x 1 ) between the center of the dipole antenna and the head center is varied. One can observe that, in the presence of the head, the antenna gain can be lowered as much as ∼5.7 dB (with x 1 = 0) as compared with the free-space dipole antenna gain. This reduction of antenna gain is seen along ␪ = 120 deg in Figure 7.11(a). This may be an important factor in evaluating the communication link budget. As x 1 increases, the gain converges to the free-space antenna gain. Similar changes are observed in the antenna gain in the x-z direction in Figure 7.11(b). The total absorbed power in the head is 43.9% with x 1 = 0 cm, 30.1% with x 1 = 5 cm, and 14.8% with x 1 = 10 cm. Antenna gain patterns in the horizontal plane (y-z plane) at 1.9 GHz without and with the head for several dipole positions are shown in Figure 7.12(a). In this figure, we can observe that the antenna gain is significantly reduced (∼13.7 dB with x 1 = 0 at ␪ =150°) when the spherical head is present as compared to the free-space antenna gain. The antenna gain again approaches to the free-space gain as x 1 increases. The gain pattern changes in the x-z plane are shown in Figure 7.12(b). Also, the total absorbed power decreases as x 1 increases: that is, 35.3% with x 1 = 0, 24.6% with x 1 = 5 cm, 13.3% with x 1 = 10 cm. 7.5.6 Comparison with EEM and FDTD In Figures 7.13(a) and (b), the gain patterns of a half-wave dipole antenna (in the x-z and y-z planes) using the FDTD method in the presence of a homogeneous spherical head are

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Figure 7.11 Gain patterns of a ␭ /2 antenna at 900 MHz with and without a six-layered spherical head as changing the antenna vertical location (x 1 ) (a) in the y-z plane and (b) in the x-z plane. The antenna delivered power is 1W.

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Figure 7.12 Gain patterns of a ␭ /2 antenna at 1.9 GHz with and without a six-layered spherical head as changing the antenna vertical location (x 1 ) (a) in the y-z plane and (b) in the x-z plane. The antenna delivered power is 1W.

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Figure 7.13 Comparison of FDTD and EEM results with a homogeneous spherical head (⑀ r = 41.0, ␴ =0.86) and a half-wave dipole antenna at 900 MHz: gain patterns (a) in the x-z plane and (b) in the y-z plane; and (c) total electric field magnitude (20 log ( | E | ) inside the spherical head along the z -axis. The antenna delivered power is set at 1W.

compared with those using the EEM. The relative permittivity (41.0) and conductivity (0.86 S/m) of the spherical head are taken as those of brain at 900 MHz. The diameter of the head is 17.3 cm and the antenna is located 2 cm away from the surface of the head. For the FDTD calculation, the size of the cubical grid cell is 0.02␭ 0 (free-space wavelength) or 0.128␭ d (dielectric wavelength). The gain patterns from both techniques agree very well, with only minor differences observed in the backward direction. Also, the electric field distributions inside the spherical head using both techniques are compared

368

Figure 7.13 (continued).

in Figure 7.13(c). As can be seen in the figure, field distributions obtained using both techniques agree well, even for the resonant field structures. Better agreement could be obtained with finer FDTD grid cells. 7.5.7 Anatomical Head Versus Spherical Head A simple spherical head can be used to predict the peak SAR in the anatomical head. The electromagnetic interaction computations with anatomical head models [see Figures 7.21(a) and (b)] using FDTD are described in detail in Section 7.1.6. Electrical parameters of the anatomical head model are given later in Table 7.5. In the following computations, a dipole antenna is assumed to be located 2 cm away from the outer edge of the ear. To compare with the anatomical head results, antenna interactions with a homogeneous spherical head with a diameter of 17.3 cm are analyzed using the hybrid EEM. Also, EM interactions with a homogeneous head with an anatomical head shape are analyzed using FDTD. The gain patterns in the horizontal plane (y-z plane) and the vertical plane (x-z plane) are shown in Figures 7.14(a) and (b). The gain patterns are similar even with different head configurations except for minor differences in the direction of the head. In evaluating the amount of power absorption when using the handset antenna, SAR is a useful and pertinent parameter. However, SAR values change with electrical conductivity of biological tissues, and SAR distribution may not be intuitively understood. In Figure 7.15, the electric field distributions with the different configurations of the head models—a homogeneous spherical head, an anatomical head with different tissues, and a homogeneous head with an anatomical head shape—are shown. An interesting observation is that the magnitudes of the electric field near the surface of the outer edge of the different head models are very similar. Total electric field inside the head is smoothly

369

Figure 7.14 Gain patterns of a ␭ /2 dipole antenna at 900 MHz in the presence of a homogeneous spherical head, anatomical head with different tissues, and a homogeneous head with an anatomical head shape: (a) in the y-z plane and (b) in the x-z plane.

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Figure 7.15 Electric field magnitude inside three different head configurations: a homogeneous spherical head, an anatomical head with various biological tissues, and a homogeneous head with an anatomical head shape. The antenna delivered power is 1W.

decreasing even at the interface between different biological tissues due to the tangential continuity of the electric field. The peak SAR value in the anatomical head can be roughly estimated by calculating the electric field using a simple homogeneous spherical head and multiplying by the electric conductivity of the biological tissue [see (7.4)]. 7.5.8 Directional Antennas An array of antennas with appropriate amplitude and phase will produce radiation patterns that reduce the EM radiation toward the human head, thus resulting in less power absorption in the head. Previously, omnidirectional antennas were conceived as better antennas for better signal reception for mobile telecommunication equipments (MTE). However, as cell areas decrease, directional antennas may have advantages over the omnidirectional ones since they reduce the absorption in the human head and therefore prolong battery life. As an example, an end-fire directional antenna array consisting of two ␭ /2 dipole antennas at 1.9 GHz is considered in Figure 7.16(a). The gain patterns with and without the six-layered spherical head in the y-z plane (horizontal plane) are compared in Figure 7.16(b). As compared with the free-space dipole antenna gain (2.23 dB), the antenna gain of the end-fire antenna is significantly reduced in the direction of the head (−7.53 dB), but enhanced in the opposite direction (∼5.61 dB). Note that, in the presence of the spherical head, the gain of a nondirectional dipole antenna at 1.9 GHz in the direction of the head is about −6.11 dB, and the gain in the opposite direction is 4.46 dB. With this

Figure 7.16 (a) Schematic of arrangement; and (b) far-field gain patterns of an end-fire array (with two half-wave dipoles) in the presence of the spherical head in the y-z plane. Operating frequency is 1.9 GHz.

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372

end-fire antenna array, however, the total power absorption is only 5.5% of the antenna delivered power, while the total power absorption with a single ␭ /2 dipole antenna at d = 2 cm is ∼34.8%. Various directional antenna configurations with additional antenna elements or with the different current amplitude/phase arrangements have also been investigated. 7.5.9 High-Frequency Effect To accommodate transmission of the vast amount of multimedia data in wireless communications, wideband operations at Ka-band (e.g., 30 GHz) are drawing considerable attention in both wireless terrestrial and satellite communications. At these high frequencies, the popular numerical techniques such as FDTD have severe limitations in EM interaction calculations due to huge computer storage requirements and excessive computation time (see Table 7.3). This is because the typical size of the adult head is 18 cm in diameter, which corresponds to ∼75 dielectric wavelengths in the head at 30 GHz. On the other hand, the hybrid EEM/moment method technique is ideally suited to critically assessing human–antenna interactions at these high frequencies [19]. Head Model As will be demonstrated, most of the EM interaction at high frequencies occurs near the exterior region of the head. Therefore, application of the exact anatomical head model will not appreciably change the outcome of the computations. The electrical parameters for the six biological tissues at 30 GHz (see Table 7.4) were calculated using the multiple Cole-Cole dispersion equation [18]. It is observed that the conductivities of the tissues are high (e.g., 27.1 S/m for skin), whereas the permittivities are relatively low as compared with those at lower frequencies (see Table 7.1). Dimensions of the layered head model are also listed in Table 7.4. The outer radius of the spherical head is 9 cm, and the thickness of the skin layer is 1 mm. Table 7.3 Required Number of Cells for the Computation of EM Interactions Using FDTD Calculation at Different Frequencies* Frequency

Head Diameter

Skin

Number of Cells

⑀r

␴ (S/m)

900 MHz 1.9 GHz 30 GHz

3.5␭ d 7.5␭ d 75.6␭ d

0.02␭ d 0.04␭ d 0.39␭ d

42,875 421,875 432 × 106

41.1 43.2 17.62

0.65 1.29 27.78

*␭ d is the dielectric wavelength. Number of cells for FDTD calculation is based on a cell size of 0.1␭ d . Skin layer ~ 1 mm; outer radius of the spherical head ~ 18 cm.

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Table 7.4 Electrical Parameters of a Six-Layered Head Model at 30 GHz* Biological Tissues Skin Fat Bone Dura CSF Brain

30 GHz Radius a p (cm)

⑀r

␴

Mass Density (10 3 kg/m 3 )

9.00 8.90 8.76 8.35 8.30 8.10

15.52 5.91 6.12 15.52 30.72 17.62

27.10 5.33 7.21 27.10 57.81 27.18

1.01 0.92 1.81 1.01 1.01 1.04

*a p is the radius, ⑀ r the permittivity, and ␴ the conductivity of each biological tissue.

Nondirectional Antennas First, we assume that the multilayered spherical head is irradiated by a half-wave dipole antenna located 2 cm away from the sphere surface at 30 GHz as shown in Figure 7.17(a). The antenna is oriented in the x -direction. Figure 7.17(a) shows the unaveraged SAR and 1-g averaged SAR distributions in the spherical head along the z -axis (where the maximum SAR occurs). In the figure, we observe that extremely high unaveraged peak SAR (457 W/kg per 1W of antenna delivered power) occurs at the thin skin layer (∼1 mm), and most power absorption is localized near the skin (≤1 mm). This high peak SAR results from the high conductivity of the skin layer at 30 GHz. Because the head is located in the far-field region of the half-wave antenna at this frequency, the peak electric field at the outer skin layer can be roughly estimated using a plane wave approximation as shown in Figure 7.18. The peak SAR value obtained by this method for 1W of antenna delivered power is 417 W/kg, which is very close to the accurate result using EEM (457 W/kg). With 1-g averaging over a specified volume (1.1 × 1.1 × 1.1 cm3 cubical volume with eleven 1-mm cells each side of the cube), the averaged peak SAR becomes much lower (10.6 W/kg), although significant averaged SAR values are spread to a broader region in the head. The total power absorption in this case is 14.7% of the antenna delivered power, which is much smaller than those at 900 MHz (43.9%) and 1.9 GHz (35.3%). The gain patterns of the half-wave dipole antenna in the y-z plane with and without the head are shown in Figure 7.17(b). It is observed that, in the presence of the head, the gain pattern of the half-wave antenna is significantly distorted and clearly loses its omnidirectionality in this plane. Directional Antennas The peak SAR may be significantly reduced by utilizing directional antennas. For example, we consider an end-fire configuration with two half-wave dipole antennas; one antenna

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Figure 7.17 Nondirectional antenna (a half-wave dipole antenna): (a) unaveraged and 1-g averaged SAR distributions along the z -axis in the six-layered spherical head; and (b) gain patterns in the y-z plane with and without the spherical head. The antenna delivered power is 1W. Operating frequency is 30 GHz.

Figure 7.18 At high frequencies (e.g., 30 GHz), peak electric field magnitude at the outer edge of the skin layer can be roughly estimated using plane wave approximation.

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376

is located 2 cm away from the head surface and the other separated by ␭ /4 (2.5 mm) with a −90-deg phase shift. The unaveraged and 1-g averaged SAR distributions are shown in Figure 7.19(a). [Note that the scales for this figure are different from those for Figure 7.17(a).] As shown in the figure, the unaveraged peak SAR reduces to 2.8 W/kg, and the 1-g averaged SAR reduces to 0.06 W/kg with 1W of the delivered power. Also, the total power absorption is dramatically reduced to ∼1% of the delivered power. The reason for these striking reductions of SAR and power absorption is that, at 30 GHz, the head is located in the far-field region of the antenna and the incident field due to the end-fire antenna array is very small in the direction of the head. At lower frequencies, the handset antennas are usually located in the reactive near-field region, and the directional antennas may not be as effective (see Section 7.5.8). The gain patterns of the directional antenna in the y-z plane with and without the head are shown in Figure 7.19(b). In this case, the gain pattern of the directional antenna does not change appreciably in the presence of the head except for small changes in the direction of the head. 7.6 RESULTS USING THE FDTD METHOD The results from the EEM outlined in the previous section provide a wide variety of engineering data that can be used to assess the behavior of many different antenna–tissue configurations. In all of these results, however, the antenna was limited to simple geometries such as a dipole radiator. While theoretically any antenna structure could be modeled using the EEM in conjunction with the MOM as discussed, modeling requirements of complicated handsets often become very difficult using this approach. Therefore, this section focuses on using the FDTD method to demonstrate the electromagnetic behavior of realistic handset configurations operating in the presence of human tissues. The computations will concentrate on the three handset configurations depicted in Figure 7.2. 7.6.1 Tissue Models The derivations provided in Section 7.3 illustrate that modeling of anatomical features within the FDTD framework can be accomplished by assigning appropriate values for the permittivity and conductivity at each spatial grid cell. The human hand is simply modeled as a layer of bone surrounded by a layer of muscle that covers three sides of the handset as depicted in Figure 7.20. An anatomical head model based on MRI scans of a human head has also been constructed. Figure 7.21(a) shows the full head model with the hand and the handset, and Figure 7.21(b) illustrates a midsagittal cross section of the head. The cell size for the computations is 3.28 mm. Clearly, more refined cell sizes can be used if necessary. The tissues utilized in the models are listed in Table 7.5. Figures 7.22(a) and (b) show two different views of the computational model along with relevant dimensions. It is noteworthy that in order to allow accurate modeling of realistic operator/handset configurations, it is important to allow rotation of the handset. This is

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Figure 7.19 Directional antenna (the end-fire antenna): (a) unaveraged and 1-g averaged SAR distributions along the z -axis in the six-layered spherical head; and (b) gain patterns in the y-z plane with and without the spherical head; z direction points away from the head. The antenna delivered power is 1W. Operating frequency is 30 GHz.

378

Figure 7.20 Model of the human hand used in the FDTD simulation of antennas on a handset.

accomplished by rotating the position of each cell in the head model in the y-z plane about the head center and reconstructing the grid based on these rotated tissue locations as implied in Figure 7.22(a). 7.6.2 Input Impedance and the Importance of the Hand Position A first investigation into the influence of biological tissue on handset-mounted antennas involves a study of the effect of the operator on the antenna input impedance behavior. This study was conducted for each handset in Figure 7.2 by first computing the antenna reflection coefficient | S 11 | for each configuration (assuming a 50⍀ transmission line) with no tissue and comparing the result to that for the handset with the tissue. For each case, the hand was introduced first and moved to different vertical positions along the handset (denoted by the parameter d as shown in Figure 7.22). The impedance results for the monopole antenna as a function of frequency are shown in Figure 7.23. As can be seen, the introduction of the hand slightly shifts the antenna resonant frequency. However, the results indicate that the impedance behavior is relatively insensitive to the absolute hand position. The addition of the head also produces a shift in the antenna resonant behavior due to the loading by the high-permittivity tissue. It is interesting to contrast these results with those from the side-mounted PIFA antennas. Reflection coefficient curves for this configuration are provided in Figure 7.24 for several

Figure 7.21 (a) Head model with hand and the plastic covered handset; and (b) a sagittal cut of the discrete head model at the head center.

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380

Table 7.5 Relative Permittivity, Conductivity, and Density of the Tissues in the Hand and Head Near 900 MHz Tissue Bone Skin/fat Muscle Brain Humour Lens Cornea

Permittivity

Conductivity (S/m)

Density (× 10 3 kg/m 3 )

8.0 34.5 58.5 55.0 73.0 44.5 52.0

0.105 0.60 1.21 1.23 1.97 0.80 1.85

1.85 1.10 1.04 1.03 1.01 1.05 1.02

Figure 7.22 (a) Side and (b) rear views of the FDTD head/hand/handset model showing dimensions.

different hand positions. In this case, because of the element locations, the operator may very easily mask part or all of the antenna with his hand while using the device. Most significant is the high impedance mismatch that occurs when the hand begins to mask the antenna. This mismatch will result in decreased communication range due to the decreased amount of signal power transmitted or received through the antenna. These results illustrate the importance of minimizing antenna masking through proper antenna placement. The dots in the figure indicate experimentally obtained results for the antenna when the hand is absent and when the hand is at d = 6.56 cm. This comparison shows good correlation between the experimentally and computationally obtained results. The curves in Figure 7.25 represent | S 11 | with the hand at d = 7.21 cm and the head at b = 1.97 cm. Two different models of the head have been used in this example. The curve labeled ‘‘sphere’’ corresponds to the case of a homogeneous spherical ball of muscle (⑀ r = 58.5, ␴ = 1.21 S/m) with a radius of 9 cm. The second head is the anatomical inhomogeneous head model. As can be seen, for this particular configuration the choice

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Figure 7.23 | S 11 | versus frequency for the monopole of Figure 7.2(a) with no tissue, with the hand at two locations, and with the head and hand (b = 1.97 cm, d = 5.90 cm).

Figure 7.24 Computed value of | S 11 | for the side-mounted PIFA on the handset without the hand and with the hand for three different values of d. Measured values appear for the configurations with no hand and with the hand at d = 6.56 cm.

of models exercises little influence on the antenna input impedance. This insensitivity occurs because the input impedance is a reasonably local phenomenon that is influenced most significantly by structures in the near vicinity of the antenna rather than objects such as the head, which are displaced somewhat from the antenna feed point. Once again, the measured data given in Figure 7.25 show that the FDTD accurately predicts the effects of the human operator on the antenna performance.

382

Figure 7.25 | S 11 | versus frequency for the side-mounted PIFA on the handset with the head and the hand (b = 1.97 cm, d = 7.21 cm). Computed results obtained using both head models are compared with measured data.

Figure 7.26 illustrates the | S 11 | behavior of the back-mounted PIFA, where once again the different curves illustrate the performance when no tissue is present, when the hand is at four positions, and when the hand (d = 4.26 cm) and the head (b = 1.97 cm) are included. These results again demonstrate the change in resonance frequency and resonant resistance with changing tissue locations. For this configuration, once the hand comes above the step in the conducting chassis, significant degradation to the antenna performance begins to occur.

Figure 7.26 | S 11 | versus frequency for the back-mounted PIFA on the handset for different handset/tissue configurations. When the head is included, b = 1.97 cm and d = 4.26 cm.

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7.6.3 Gain Patterns As expected, the presence of the operator tissue can have a pronounced effect on the radiation properties of the antenna because it impacts the total radiated power, the shape of the pattern, and the polarization characteristics of the radiated field. As an example, consider the gain patterns shown in Figure 7.27 for the monopole antenna on the handset at 915 MHz. For these computations, the handset is rotated 60° from upright in the y-z plane. Results for both the handset alone and the handset in the presence of the tissue are provided (b = 1.97 cm, d = 5.90 cm). As can be seen, the peak gain, pattern shape, and radiated field polarization are all impacted by the presence of the tissue. Figure 7.28 shows similar patterns for the back-mounted PIFA for the same tissue configuration and handset rotation. It is interesting that in this case, while the tissue does influence the radiation pattern, it appears that less gain loss occurs for this geometry as compared to that for the monopole antenna. Figure 7.29 provides a thorough study of the pattern characteristics for the sidemounted PIFA configuration. In this set of plots, the patterns in the principal vertical planes are shown for no tissue, for the homogeneous spherical head model, and for the inhomogeneous head model (b = 1.97 cm, d = 7.21 cm). In this case, it is apparent that while there is some difference in the results obtained from the two different head models, these differences are not drastic. This provides confidence in the ability to approximately assess the effects of tissue on antenna radiation characteristics using simplified tissue models.

Figure 7.27 Gain patterns for the monopole antenna on the handset at 915 MHz without and with the head and hand (b = 1.97 cm, d = 5.90 cm). The antennas are rotated 60° from upright.

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Figure 7.28 Gain patterns for the back-mounted PIFA on the handset at 915 MHz without and with the head and hand (b = 1.97 cm, d = 5.90 cm). The antennas are rotated 60° from upright.

7.6.4 Near Fields and SAR Figure 7.30 compares the field variation around the handset, head, and hand at 915 MHz for the side-mounted PIFA (both inhomogeneous and spherical head models), the monopole, and the back-mounted PIFA. Figure 7.31 presents the SAR distribution for the same configurations. For each computation, d = 7.21 cm, b = 1.97 cm, and the handset is upright for simplicity in data presentation. The data plane is located at the center of the head/handset combination, and the plots are viewed from the −y -direction (see Figure 7.22). The values in Figure 7.30 represent the total squared magnitude of the peak electric field per watt of total power delivered to the antenna. Similarly, the values in Figure 7.31 represent the SAR averaged over 1g of tissue and normalized to the power delivered to the antenna. Table 7.6 provides a numerical comparison of the peak SAR values occurring in the head and hand for each of the configurations in Figures 7.30 and 7.31. The two sets of data correspond to the configurations where the handset is rotated 60° and the handset is upright with respect to the head. As can be seen, each of the antenna structures results in very similar values of the peak SAR for both handset orientations with the exception of the side-mounted PIFA geometry. In this case, the rotated handset places the antenna nearly in contact with the ear tissue, resulting in a somewhat higher SAR value as compared to the upright handset. For all of the configurations, the peak SAR in the head occurs either in the ear tissue or in the skin/fat layer in the antenna vicinity. Table 7.6 also

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Figure 7.29 Computed patterns at 915 MHz normalized to the antenna gain for the side-mounted PIFA on the handset rotated 60° from upright. Results are shown for the handset alone, with the spherical head model, and with the inhomogeneous head model (b = 1.97 cm, d = 7.21 cm).

386

Figure 7.30 Computed normalized near-field distribution at 915 MHz in a plane through the center of the head with d = 7.21 cm, b = 1.97 cm. The configurations are the side-mounted PIFA, the sidemounted PIFA with the spherical head, the monopole, and the back-mounted PIFA.

provides the SAR averaged over the entire head, which as expected is considerably lower than the peak SAR levels. A very interesting phenomenon is observed for the back-mounted PIFA, which has a peak SAR in the head that is considerably reduced in comparison to the values for the other antennas. This occurs because the conducting handset chassis lies between the antenna and the head, providing some degree of shielding from exposure. These numbers and plots also show that the spherical and inhomogeneous head models predict slightly different absorption and SAR characteristics. It is important to note that the numbers

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Figure 7.31 Computed normalized SAR distribution at 915 MHz in a plane through the center of the head with d = 7.21 cm, b = 1.97 cm. The configurations are the side-mounted PIFA, the sidemounted PIFA with the spherical head, the monopole, and the back-mounted PIFA.

provided here are consistent with previously reported measured and computed results [20–23]. Comparisons such as these are very useful for determining the suitability of different radiators for personal communications applications. Table 7.6 also presents the fraction of power absorbed in the head and hand for the different antenna topologies, along with the radiation efficiency ␩ a of the configuration. As can be seen from these results, the large amount of power dissipation considerably reduces the antenna gain, with the efficiencies dropping below 50% for most cases. This gain loss is an extremely important issue that should be considered when planning link budgets for wireless communications systems involving handheld subscriber units.

Pdel

0.472 0.317 0.470 0.518 0.465 0.344 0.517

0.184 0.324 0.225

␩a

0.169 0.301 0.318 0.222

Pdel

hand

P abs

*All data are computed at 915 MHz and 1W delivered power. † With homogeneous spherical head (⑀r = 58.5, ␴ = 1.21).

Handset rotated 60 deg with respect to head Monopole 0.359 Side-mounted PIFA 0.382 0.212 Side-mounted PIFA† Back-mounted PIFA 0.260 Handset upright with respect to head Monopole 0.351 Side-mounted PIFA 0.332 Back-mounted PIFA 0.258

Configuration

P abs

head

2.06 2.07 0.90

1.97 3.81 3.14 1.32

Pdel

head

SAR max

2.43 4.91 3.53

2.29 4.54 5.42 3.58

Pdel

hand

SAR max

0.0856 0.0809 0.0629

0.0875 0.0931 0.1003 0.0634

Pdel

head

SAR ave

Table 7.6 Computed Normalized Power Absorption and Peak SAR (mW/g/W) in the Head and Hand, Average SAR (mW/g/W) in the Head, and Radiation Efficiency for the Different Handset/Body Configurations Shown in Figure 7.31*

388

389

The preceding results for radiation efficiency and peak SAR in the head have all been provided for a given separation between the head and the handset (b = 1.97 cm). However, it is interesting to examine the effect of this distance on these parameters. Figure 7.32 presents the variation of the antenna efficiency and peak SAR (1W delivered power) in the head versus the distance b for the monopole and the back-mounted PIFA configurations with the handset upright. As might be expected, the radiation efficiency increases with distance, while the peak SAR decreases in a nearly exponential fashion. 7.7 ASSESSMENT OF DUAL-ANTENNA HANDSET DIVERSITY PERFORMANCE The majority of handsets currently in use for mobile communications systems utilize only a single radiating element. However, there is increasing interest in the placement of two antennas on the handset to allow signal diversity combining for multipath fading mitigation [24, 25]. In this case, it is interesting to assess the effect of the tissue on the diversity performance of the dual-antenna configuration [26–28]. One difficulty in predicting this diversity performance is its dependence on the propagation environment in which the handset operates. While formulations exist for assessing diversity gain given envelope correlation coefficients in a Rayleigh distributed fading scenario, they are only approximate in nature and rely on the different antennas having matched radiation characteristics. Therefore, a complete characterization of antenna diversity performance must rely on computational as well as experimental studies. The goal of this section is to outline computational and experimental procedures and results for assessing the diversity performance of dual-antenna handsets operating in typical indoor propagation environments.

Figure 7.32 Antenna efficiency ␩ a (%) and peak SAR (Pdel = 1W) in the head for the monopole and backmounted PIFA versus the distance between the head and handset. The handset is in an upright position.

390

7.7.1 Dual-Antenna Handset Geometries The two different handset geometries shown in Figure 7.33 have been used in the computational and experimental observations in this study. The first configuration, shown in Figure 7.33(a), consists of two quarter-wavelength monopole antennas, with one mounted on the handset top and another lying on a hinged mouthpiece at the bottom of the handset. The second geometry, shown in Figures 7.33(b) and (c), utilizes the same mouthpiece-mounted monopole structure, but replaces the top-mounted antenna with the back-mounted PIFA. For simplicity of simulation and consistency in the data collection, the mouthpiecemounted antennas are positioned to be perpendicular to the handset chassis, as shown in Figure 7.33. The handset bodies have been constructed as aluminum and copper casings and are modeled as perfect conductors in the numerical simulations. 7.7.2 Simulated Assessment of Diversity Performance The FDTD simulation technique allows computation of antenna parameters that can serve as key indicators of diversity performance in a multipath fading environment. The two parameters chosen for this study are the antenna mean effective gain (MEG) and antenna correlation coefficient (␳ e ). To compute these parameters, the FDTD methodology is

Figure 7.33 Two dual-antenna diversity handset configurations: (a) monopole/mouthpiece monopole: (b) PIFA/mouthpiece monopole; and (c) front side of handset in part (b).

391

utilized to compute the antenna vector power gain pattern G (␪ , ␾ ), which has ␪ˆ and ␾ˆ vector components G ␪ and G ␾ , respectively. The MEG G e is then computed using the expression [29] Ge =

Prec 1 = S␪ + S␾ 1 + X

冕冕 2␲

0

␲

[XG ␪ (␪ , ␾ )P␪ (␪ , ␾ ) + G ␾ (␪ , ␾ )P␾ (␪ , ␾ )] sin ␪ d␪ d␾

0

(7.21) where S ␪ and S ␾ represent the average power contained in the ␪ˆ and ␾ˆ components of the incident field (with respect to the antenna coordinate system), respectively, and Prec refers to the average power received by the antenna along a random route of the handset. The term X = S ␪ /S ␾ is referred to as the cross-polarization discrimination ratio (XPR) and is dependent on the particular propagation environment of interest. Values P␪ and P␾ represent the angular density of plane waves polarized in the ␪ˆ and ␾ˆ directions in the environment. In this study, they are approximated using the expression [29]

冋

A␰ (␪ − m␰ )2 P ␰ (␪ , ␾ ) = exp − 2␲ 2␴ 2␰

册

(7.22)

where ␰ = ␪ or ␾ and m␰ and ␴␰ denote the mean and standard deviation of the elevation angle for the ␰ polarization. In this work, a value of m␰ = ␲ /2 (horizontal plane) will be assumed for all computations. Also, based on the findings in [24,25] that incident multipath signals arrive at elevation angles less than 40°, we will use ␴␪ = ␴␾ = 40°. The coefficient A␰ is determined from the property

冕 冕 2␲

0

␲

P␰ (␪ , ␾ ) sin ␪ d␪ d␾ = 1

(7.23)

0

The envelope correlation coefficient is a metric for assessing the temporal correlation of the signal envelopes at the two antenna terminals. Unfortunately, no simple formulation exists for obtaining this parameter directly based on the antenna gain characteristics and the average parameters of the multipath field. Therefore, a commonly used alternative approach that evaluates the signal correlation coefficient ␳ s will be employed in this study. The envelope correlation coefficient can then be approximated using the relation [30]

␳ e ≈ | ␳ s |2

(7.24)

The details of this approach are provided in [14]. It has been shown that this relation is relatively accurate for urban environments where the multipath received signal is Rayleigh

392

distributed. Therefore, it is interesting to compare the results of this simple approach to values obtained from the experimental measurements outlined below for the indoor environments assessed in this work. 7.7.3 Experimental Assessment of Diversity Performance The platform used to acquire measurements of different propagation and antenna diversity characteristics is shown schematically in Figure 7.34. The transmitter consists of a HewlettPackard HP 8657A sweep oscillator at 915 MHz, which is connected to a vertically oriented balun-fed dipole antenna. The receiving subsystem consists of two independent branches, allowing simultaneous measurement of signals received on the two antennas. Each antenna signal is routed through a low-noise amplifier providing 23 dB of gain into a Hewlett-Packard HP 8590B spectrum analyzer operating in zero-span mode at the transmission frequency. The output voltage from the analyzer is sampled using a twochannel analog-to-digital converter (ADC), which offers a sample rate of 400 Hz per channel. During the measurement campaign, the system was calibrated by putting a known power level into each channel and properly weighting the ADC output to accurately represent the power level. A test performed by sweeping the input power from −40 to −70 dBm showed the system power reading to be linear to within 0.25 dB. The data for this study were collected in two different buildings on a university campus. The first building has steel-reinforced concrete structural walls and cinder-block/ brick partition walls. Two different measurement sites were used in this building. For site 1, the transmitter and receiver were placed about 50m apart on the building fourth floor. For site 2, the transmitter was placed on the third floor with the receiver (approximately 40m away) on the fourth floor. The second building contains steel-reinforced concrete structural walls and drywall/steel-stud partition walls. Only one measurement site (site 3) was used in this building, with the transmitter and receiver on the same floor and separated by a distance of approximately 30m. The measurements at each site were performed by moving the transmitter over a circular path with a 3-m diameter. This pattern was chosen because (1) it localizes the

Figure 7.34 Data acquisition platform consisting of two-branch receiver and remote transmitter.

393

antenna position such that the mean signal envelope remains relatively constant over the measurement, and (2) it lends repeatability to the measurements to allow comparison between operation with and without human tissue. In each case, the handset was moved at a rate of approximately 0.5 m/s. In measurements involving human operators, data were collected for two different subjects (adult males). Simple measurements of XPR using two orthogonally oriented dipole antennas yielded values of X 1 = X 2 = 5 dB at sites 1 and 2 and X 3 = 1.5 dB at site 3. The first evaluation of the antennas shown in Figure 7.33 consists of measuring the MEG for each element with and without the presence of the operator. In the measurement setup, the handset under test was mounted to a Styrofoam cross-member attached to a Styrofoam mast, approximately 2 m above the ground, affixed to the cart carrying the receiver subsystem. The handset was oriented at 60° from vertical in order to allow comparisons between performance with and without the tissue. One of the handset antennas was connected to the receiver, while the second was terminated in a 50⍀ load. The second receiver branch was connected to a vertically polarized half-wave dipole antenna positioned 50 cm from the handset on the Styrofoam cross-member. Following completion of measurements for this configuration, the handset was removed from the mast and held by an operator to the right ear again at an angle of 60°. The operator walked in front of the cart (approximately 50 cm from the cart-mounted dipole antenna) as the measurement was performed. From the acquired data, the MEG was computed using the formula [31]

MEG =

P AUT

(7.25)

Pd

where P AUT and P d represent the time-average power received by the antenna under test and the dipole, respectively. It should be noted that in the FDTD simulations, the computed MEG is normalized by the computed MEG of a dipole antenna to allow comparisons between measured and simulated data. To measure the envelope correlation coefficient, the two handset antennas were simultaneously connected to the receiver system. The handset (rotated 60°) was either mounted on the mast or held by the user. If V 1 and V 2 represent voltage envelopes collected at antennas 1 and 2, respectively, the envelope correlation coefficient is given as

␳e =

E 冋冠V 1 − V 1 冡冠V 2 − V 2 冡册 E 冋冠V 1 − V 1 冡 册E 冋冠V 2 − V 2 冡

√

2

where V i represents a time average of voltage envelope V i .

册

2

(7.26)

394

7.7.4 Results Table 7.7 provides a summary of the computed and measured MEG values for each of the different handset antennas in the three environments under consideration. The results show reasonable agreement between the two sets of data. The discrepancies most likely arise from incorrect modeling assumptions in the MEG simulation model. Additionally, the values provided in Table 7.7 are similar to values reported in the literature for antenna/ handset configurations of similar construction [26, 31, 32]. The important finding of Table 7.7 is the uniformity of MEG for antennas on a single handset. With few exceptions, the values for different antennas on a handset remain within 1.5 dB and are generally within 0.5 dB. The similarity of these values is essential for achievement of high diversity gain. Additionally, the data show that the tissue reduces the MEG by between 3 and 6 dB. Finally, it is noteworthy that the widest variability in MEG occurs at site 2, most likely due to the multipath structure observed there. Table 7.8 shows measured and computed values of ␳ e for the two different handsets at the three different measurement sites. Three different observations can be made concerning these data. First, both measurements and simulations show that correlation coefficients for each handset are very low (generally less than 0.2). This implies that both handsets will offer reasonable diversity performance. Second, while simulation shows that the operator tissue always reduces ␳ e , measurements indicate that the operator can actually increase the envelope correlation, perhaps because of strong mutual antenna interactions induced by the high-permittivity tissues in the head. Finally, while the simulation approach precludes the possibility of negative correlation coefficients, Table 7.8 indicates that measured values of ␳ e can be less than zero. This occurrence of negative correlation Table 7.7 Measured and Simulated Values of MEG for Individual Handset Antennas With and Without the User Present Handset 1

Site 1

No user User

Site 2

No user User

Site 3

No user User

Measured Simulated Measured Simulated Measured Simulated Measured Simulated Measured Simulated Measured Simulated

Handset 2

Monopole

Mouthpiece

PIFA

Mouthpiece

−2.32 −3.20 −5.44 −7.63 −1.27 −3.20 −7.17 −7.63 −0.82 −1.70 −4.89 −5.92

−2.90 −3.26 −5.67 −7.60 −2.35 −3.26 −5.46 −7.60 −0.99 −1.81 −4.10 −5.60

−2.66 −3.47 −6.55 −6.71 −2.00 −3.47 −7.06 −6.71 −1.81 −2.12 −5.61 −5.36

−2.01 −4.24 −5.83 −7.52 −0.89 −4.24 −7.33 −7.52 −1.61 −2.41 −5.95 −5.64

395

Table 7.8 Measured and Simulated Values of Envelope Correlation Coefficients for the Handset Branches With and Without the User Present

Site 1

No user User

Site 2

No user User

Site 3

No user User

Measured Simulated Measured Simulated Measured Simulated Measured Simulated Measured Simulated Measured Simulated

Handset 1

Handset 2

0.1262 0.0377 0.0758 0.0006 0.2866 0.0377 0.0908 0.0006 0.0965 0.0614 0.2719 0.0015

−0.0361 0.1329 0.1246 0.0008 0.0138 0.1329 0.0521 0.0008 0.1147 0.1641 0.2979 0.0043

coefficients implies the possibility of even higher diversity gain than what is predicted by standard theoretical approaches which assume that ␳ e > 0. However, it is notable that only one negative value appears, and its magnitude is relatively small. It is conceivable that this negative value would become positive if more data were used in the computation. The preceding results indicate that the antennas of Figure 7.33 possess similar MEG values and low values of ␳ e , suggesting that they should offer good diversity performance. It is interesting to assess this performance by directly measuring the diversity gain for each handset in the different environments. This is accomplished by taking the experimental data collected for determination of ␳ e and forming signal-to-noise ratio (SNR) histograms for each antenna individually and then for a diversity combined system. In this case, only selection combining is used such that the sequence used for the diversity histogram is obtained by comparing the SNR reading at the two antennas for each sample and choosing the largest value. Diversity gain is then obtained by computing the cumulative density function (CDF) from each histogram. The gain is defined as the difference in SNR between the single-branch (highest of the two branch signals is chosen) and combined signals at a given probability level. Table 7.9 illustrates the diversity performance of the two handsets at three different reliability levels. The values represent averages of three different measurements for each site. The theoretical results are taken from curves in [25] for zero branch correlation. The ‘‘average’’ values are obtained by averaging over all measurement sites and both users. As can be seen, selection combining offers a gain of 7–10 dB at the 99% level. It is noteworthy that while the tissue tends to lower the antenna MEG, it has little impact on the diversity gain of the system. Finally, the results indicate that the theoretical gains obtained from ideal branches are reasonably close to what has been observed in these measurements. Discrepancies in these values may arise from the nonequal effective antenna

396

Table 7.9 Averaged Diversity Gain Values for Both Handsets at Three Reliability Levels Handset 1

Site 1

No user User 1 User 2 Site 2 No user User 1 User 2 Site 3 No user User 1 User 2 Average No user User Theory ( ␳ e = 0)

Handset 2

90%

95%

99%

90%

95%

99%

3.8 4.7 4.3 4.4 4.4 4.0 5.0 4.6 4.3 4.4 4.4 5.6

5.0 5.6 5.3 5.6 5.4 5.2 6.4 5.5 5.9 5.6 5.4 6.7

7.9 6.6 5.9 8.4 5.4 6.4 9.8 9.1 11.2 8.7 7.1 10.0

5.6 4.1 4.9 5.1 4.4 4.3 5.0 5.7 3.8 5.2 4.9 5.6

7.6 4.7 6.8 6.0 4.7 4.7 6.4 6.9 5.2 6.7 5.8 6.7

9.8 5.6 10.0 8.9 4.5 5.3 9.8 11.5 9.4 9.5 8.1 10.0

gains as well as the nonzero envelope correlation coefficients characterizing the handset antennas. REFERENCES [1] Jensen, M. A., and Y. Rahmat-Samii, ‘‘EM Interaction of Handset Antennas and a Human in Personal Communications,’’ Proc. IEEE, Vol. 83, No. 1, pp. 7–17, January 1995. [2] Rahmat-Samii, Y., and W. L. Stutzman, (eds.), ‘‘Special Issue on Wireless Communications,’’ IEEE Trans. Ant. Propagat., Vol. 46, June 1998. [3] Rosen, A., and A. Vander Vorst, (eds.), ‘‘Special Issue on Medical Application and Biological Effects of RF/Microwaves,’’ IEEE Trans. on Microwave Theory and Tech., Vol. 44, October 1996. [4] Gandhi, O. P., ‘‘FDTD in Bioelectromagnetics: Safety Assessment and Medical Applications,’’ in Computational Electromagnetics: The Finite Difference Time Domain Method, A. Taflove, (ed.), Norwood, MA: Artech House, 1995. [5] Colburn, J. S., and Y. Rahmat-Samii, ‘‘Electromagnetic Scattering and Radiation Involving Dielectric Objects,’’ J. Electromagnetic Waves and Applications, Vol. 9, No. 10, pp. 1249–1277, 1995. [6] Kim, K. W., and Y. Rahmat-Samii, ‘‘Antennas and Humans in Personal Communications: An Engineering Approach to the Interaction Evaluation,’’ Proc. IEEE Engineering in Medicine and Biology Society, Chicago, October 1997, pp. 2488–2491. [7] Kim, K. W., and Y. Rahmat-Samii, ‘‘EM Interactions Between Handheld Antennas and Human: Anatomical Head vs. Multi-layered Spherical Head,’’ IEEE-APS Conference on Antennas and Propagation for Wireless Communications, Waltham, MA, November 2–4, 1998, pp. 69–72. [8] Gandhi, O. P., G. Lazzi, and C. M. Furse, ‘‘Electromagnetic Absorption in the Human Head and Neck for Mobile Telephones at 835 and 1900 MHz,’’ IEEE Trans. Microwave Theory Tech., Vol. 44, pp. 1884–1897, October 1996. [9] Okoniewski, M., and M. A. Stuchly, ‘‘A Study of the Handset and Human Body Interaction,’’ IEEE Trans. Microwave Theory Tech., Vol. 44, pp. 1855–1864, October 1996. [10] Hombach, V., et al., ‘‘The Dependence of EM Energy Absorption upon Human Head Modeling at 900 MHz,’’ IEEE Trans. Microwave Theory Tech., Vol. 44, pp. 1865–1873, October 1996.

397

[11] ANSI/IEEE C95.1-1992, American National Standard—Safety Levels with Respect to Exposure to Radio Frequency Electromagnetic Fields, 3 kHz to 300 MHz. New York: IEEE. [12] Yee, K. S., ‘‘Numerical Solution of Initial Boundary Value Problems Involving Maxwell’s Equations in Isotropic Media,’’ IEEE Trans. Antennas Propagat., Vol. AP-14, pp. 302–307, May 1966. [13] Taflove, A., Computational Electromagnetics: The Finite-Difference Time-Domain Method, Norwood, MA: Artech House, 1995. [14] Jensen, M. A., and Y. Rahmat-Samii, ‘‘Performance Analysis of Antennas for Hand-Held Transceivers Using FDTD,’’ IEEE Trans. Antennas Propagat., Vol. 42, pp. 1106–1113, August 1994. [15] Mur, G., ‘‘Absorbing Boundary Conditions for the Finite-Difference Approximation of the Time-Domain Electromagnetic Field Equations,’’ IEEE Trans. Electromagnetic Compatibility, Vol. 23, pp. 377–382, 1981. [16] Berenger, J.-P., ‘‘A Perfectly Matched Layer for the Absorption of Electromagnetic Waves,’’ J. Computational Physics, Vol. 114, pp. 185–200, 1994. [17] Tai, Chen-To, Dyadic Green Functions in Electromagnetic Theory, 2nd ed., IEEE Press Series on Electromagnetic Waves, New York: IEEE Press, 1994. [18] Gabriel, S., R. W. Lau, and C. Gabriel, ‘‘The Dielectric Properties of Biological Tissues: III. Parametric Models for the Dielectric Spectrum of Tissues,’’ Phys. Med. Biol., Vol. 41, pp. 2271–2293, November 1996. [19] Kim K. W., and Y. Rahmat-Samii, ‘‘Handset Antennas and Humans at Ka-Band: The Importance of Directional Antennas,’’ IEEE Trans. Antennas Propagat., pp. 949–50, June 1998. [20] Toftga˚rd, J., S. N. Hornsleth, and J. B. Andersen, ‘‘Effects on Portable Antennas of the Presence of a Person,’’ IEEE Trans. Antennas Propagat., Vol. 41, pp. 739–746, June 1993. [21] Dimbylow, P. J., ‘‘FDTD Calculations of the SAR for a Dipole Closely Coupled to the Head at 900 MHz and 1.9 GHz,’’ Phys. Med. Biol., Vol. 38, pp. 361–368, February 1993. [22] Chuang, H. R., ‘‘Human Operator Coupling Effects on Radiation Characteristics of a Portable Communication Dipole Antenna,’’ IEEE Trans. Antennas Propagat., Vol. 42, pp. 556–560, April 1994. [23] Mumford, R., Q. Balzano, and T. Taga, ‘‘Land Mobile Antenna Systems II: Pagers, Portable Phones, and Safety,’’ Ch. 4 of Mobile Antenna Systems Handbook, K. Fujimoto and J. R. James, (eds.), Norwood, MA: Artech House, 1994. [24] Lee, W. C. Y., Mobile Communications Engineering, New York: John Wiley & Sons, 1982. [25] Jakes, W. C., Jr., Microwave Mobile Communications, New York: John Wiley & Sons, 1974. [26] Pedersen G. F., and S. Skjaerris, ‘‘Influence on Antenna Diversity for a Handheld Phone by the Presence of a Person,’’ Proc. 1997 47th IEEE Vehicular Technology Conference, Phoenix, AZ, May 4–7, 1997, Vol. 3, pp. 1768–1772. [27] Green, B. M., and M. A. Jensen, ‘‘Diversity Performance of Personal Communications Handset Antennas Near Operator Tissue,’’ 1997 IEEE AP-S Intl. Symp. Digest, Montreal, Canada, July 13–18, 1997, Vol. 2, pp. 1182–1185. [28] Colburn, J. S., et al., ‘‘Evaluation of Personal Communications Dual-Antenna Handset Diversity Performance,’’ IEEE Trans. Vehicular Technology, Vol. 47, pp. 737–746, August 1998. [29] Taga, T., ‘‘Analysis for Mean Effective Gain of Mobile Antennas in Land Mobile Radio Environments,’’ IEEE Trans. Vehicular Technology, Vol. 39, pp. 117–131, May 1990. [30] Pierce, J. N., and S. Stein, ‘‘Multiple Diversity with Nonindependent Fading,’’ IRE Proc., pp. 89–104, January 1960. [31] Murase, M., Y. Tanaka, and H. Arai, ‘‘Propagation and Antenna Measurements Using Antenna Switching and Random Field Measurements,’’ IEEE Trans. Vehicular Technology, Vol. 43, pp. 537–541, August 1994. [32] Arai, H., N. Igi, and H. Hanaoka, ‘‘Antenna-Gain Measurement of Handheld Terminals at 900 MHz,’’ IEEE Trans. Vehicular Technology, Vol. 46, pp. 537–543, August 1997. [33] Johnson, R. C., Antenna Engineering Handbook, 3rd ed., New York: McGraw-Hill, 1980.

Chapter 8 Digital TV Antennas for Land Vehicles Kunitoshi Nishikawa, Hideo Iizuka, and Kyohei Fujimoto

Japanese digital television services and the demonstration of the mobile reception in Tokyo area are presented in this chapter. Then, automobile antennas for digital television reception are presented. The antennas were designed for such parameters as limited installation spaces, omnidirectional radiation pattern, and so on, and include not only prototype antennas but also antennas on the market. 8.1 RECEPTION SYSTEMS 8.1.1 Digital Television Services in Japan Digital television (TV) services have been available in Europe and North America since September 1998. In Japan, the services were started in three large cities, Tokyo, Osaka, and Nagoya, in December 2003, and are spreading over a large area. Frequency bandwidth is assigned from 470 to 710 MHz, and horizontal polarization is utilized. The Japanese terrestrial digital broadcasting protocol is designated as Integrated Service Digital Broadcasting-Terrestrial (ISDB-T), and Band Segmented Transmission-Orthogonal Frequency Division Multiplexing (BST-OFDM) has been adopted in ISDB-T to simultaneously provide a variety of services over a single channel [1]. The bandwidth of each channel is about 6 MHz, and is comprised of 13 segments; 12 out of the 13 segments are currently used for broadcasting of high definition television (HDTV) for the home environment, with the remainder being used for mobile terminals [2]. 399

400

Broadcasting to mobile terminals employs quadrature phase shift keying (QPSK) as a modulation method. Since this modulation method features noise resistance, reception can easily be achieved even in automobiles. However, because the bit rate that can be transmitted in one segment is low, the quality of the images is not sufficiently high for larger monitors to be installed in automobiles, although it is sufficient for watching on the small screens of mobile phones. On the other hand, 64-quadrature amplitude modulation (64-QAM) has been adopted for HDTV broadcasting to the home using the 12 available segments. Naturally, the noise resistance of this method is considerably weaker than single segmentation broadcasting. Thus, stable reception of broadcasts to automobiles is difficult while they are moving [3]. Nonetheless, if it could be realized, there is a possibility that it could change the role of television sets in automobiles. For example, HDTV images of terrestrial digital broadcasting displayed on a large screen positioned in the rear seat would definitely be a main media for entertainment. Demand for the service would be very high, not only for personal use in vehicles, but also for taxies, buses, or trains. Under these circumstances, mobile receivers are being developed so that viewers can enjoy HDTV programs anywhere they may go [4, 5]. 8.1.2 Problems of Mobile Reception Up to now, it has been a big challenge to make mobile receivers that can display HDTV programs in a car. The first reason is that signal-to-noise ratio in mobile reception is much lower than that in fixed reception. As shown in Figure 8.1, compared with an antenna for fixed reception in a typical household, the antenna for mobile reception is lower and, as a result, the received electric field strength is lower. According to ITU-R Recommendation P.1546 [6], the correction value for received electric field strengths in the UHF band for reception at 10m and 1.5m above ground is 16 dB. Moreover, experimental tests [7] using a Yagi antenna set at 10m above the ground and a standard dipole antenna set at 1.2m have shown that the average difference in the received electric field is 11.2 dB with a standard deviation of 3.4 dB. The second reason is that mobile reception in the car is strongly affected by multipath fading. Figure 8.2 shows an example of the measured variation in electric field strength during mobile reception. The figure shows that electric field strength instantaneously drops many times during a short traveling distance of 10m. 8.1.3 Diversity Reception Methods There are mainly two methods for diversity reception of OFDM signals. The first one is pre-FFT diversity reception. Figure 8.3 shows the basic algorithm of pre-FFT diversity reception. The incoming signals are the individual IF signal corresponding to each antenna, and the outgoing signal is the combined IF signal. First, digitized IF signals from each antenna are downconverted to baseband (BB) signals with a bandwidth of 6 MHz. Each

401

Figure 8.1 Radio wave environment for mobile reception. (From: [4].  2007 IEEE. Reprinted with permission.)

Figure 8.2 Example of field strength variation in mobile reception. (From: [4].  2007 IEEE. Reprinted with permission.)

BB signal then is divided by band segmentation filters into three subband signals with bandwidths of about 2 MHz (low, mid, and high bands). The BB signals in each subband are then weighted and combined into one. The weighting coefficient is used to control both the phase and the amplitude in order to maximize the signal-to-noise ratio (S/N) of the combined signal. The coefficient can be determined by cross-correlation computation between the BB signal corresponding to each antenna and the combined signal. Next, the combined signals for each subband are combined into one signal, with a bandwidth of 6 MHz. Finally, the combined BB signal is upconverted to an IF signal and transferred

402

Figure 8.3 Block diagram of pre-FFT diversity reception system. (From: [4].  2007 IEEE. Reprinted with permission.)

to an OFDM demodulator. This method can precisely control the beam in a multipath environment, because it divides the original 6-MHz bandwidth into narrower bandwidths and precisely controls the weighting for each band. The second method is post-FFT diversity reception. In the receiver, maximal ratio combining (MRC) is performed after an FFT operation on each branch signal. Figure 8.4 shows the block diagram of the post-FFT diversity reception system for an OFDM channel. There are four antenna branches for combining the received signals. Assuming that each signal is statistically uncorrelated, combining the signals would enhance the carrier-tonoise ratio (CNR) of the OFDM channel. In order to maximize the CNR, MRC is performed after an FFT operation for the signals of each branch. The weighting factor for each carrier is derived from the frequency response, which is calculated from the scattered pilot signal of each branch signal. 8.1.4 Demonstration Figure 8.5 shows an outline of the demonstration system of pre-FFT diversity reception. The antenna was made of transparent film, which did not obstruct the driver’s view. Two elements were attached to each of the front, rear, and side windows of the bus, totaling eight elements. The two elements on each side were set close to each other in order to

403

Figure 8.4 Block diagram of post-FFT diversity reception system. (From: [4].  2007 IEEE. Reprinted with permission.)

Figure 8.5 High-speed mobile reception system. (From: [4].  2007 IEEE. Reprinted with permission.)

404

improve the reception characteristics by compensating Doppler effects during high-speed traveling [8]. The radio waves received by these antennas were input into a signal processor for Doppler compensation and directivity control. Mobile reception characteristics were measured while traveling at high speed on an expressway in Japan. A bus equipped with the reception system was used in this experiment. Figure 8.6 illustrates the reception results, superimposed on a map of the Tokyo area where the experimental course was located. The image reception rates at each point (calculated in terms of TS packet errors) are shown, with the reception conditions being plotted on the map along the course of travel. Except in tunnels, most of the measured locations in downtown Tokyo had good reception characteristics during high-speed driving. Departing from Tokyo Tower and driving southwest on the expressway, reception characteristics gradually deteriorated until it became impossible beyond the Yokohama/Machida Interchange. The boundary lines for the fixed reception area, provided by the Association for Promotion of Digital Broadcasting (D-PA), are also shown in the figure. The lines have been plotted on the map used for this experiment for comparison with the mobile reception area of the demonstration. It is clear from the measurement that the high-speed mobile reception system has a potential for receiving terrestrial digital HDTV over almost the same area that D-PA quoted as the fixed reception area. In addition, since the experiments conducted on actual roads were limited, evaluations were also conducted in the laboratory. These bench experiments indicate that the reception system can achieve stable reception under simulated circumstances of traveling at 180 km/hr.

Figure 8.6 Experiment results of high-speed mobile reception. (From: [4].  2007 IEEE. Reprinted with permission.)

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The reception system has been also installed into road trains, as shown in Figure 8.7, and sufficient performance has been confirmed. 8.2 DIGITAL TELEVISION ANTENNAS 8.2.1 Quarter Glass Antenna for a Van Figure 8.8(a) shows the geometry of a quarter glass antenna for a digital TV [9]. The antenna has been developed by modifying a batwing antenna [10]. Since batwing antennas have a broad bandwidth, but are too large to be installed in vehicle, the development focused on size reduction. The antenna is normal to ground and fed by a coaxial cable. The antenna height H was reduced by bending the wires in that direction without narrowing the bandwidth. The height of the antenna is approximately half of the original antenna height. The prototype antenna with two antenna elements for field experiments using a passenger van is shown in Figure 8.8(b). Two antenna elements and a mesh ground plane were printed on the inside of the rear side glass window of the passenger van. The

Figure 8.7 Reception system installed into road trains.

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Figure 8.8 Configurations of quarter glass antennas for digital TV reception: (a) antenna element; and (b) prototype antennas. (From: [9].  2007 IEICE. Reprinted with permission.)

dielectric constant of the glass was 6.7, and the loss tangent was 0.008. The thickness of the glass was about 5 mm. The element was printed using silver paste, and the mesh ground plane was connected electromagnetically to the vehicle body upon installation of the window. The conductivity of the silver paste was about 2 × 106 S/m, which is lower than that of copper (5.8 × 107 S/m). Antenna gain loss was confirmed to be less than 0.3 dB within the frequency band by comparison with the measured results of an antenna composed of copper. The radiation pattern of the antenna installed on the passenger van was measured in the vertical plane on an outdoor test range. Figure 8.9 shows the radiation pattern of the antenna installed on the left side of the passenger van at 485 MHz. The solid line represents E ␾ , and the dotted line represents E ␪ . The co-polarization of DTV is E ␾ , which is horizontal. As expected, the elevation pattern peak of E ␾ was horizontal. The main polarization, E ␾ is stronger than E ␪ by 10 dB in the −x direction. Figure 8.10 shows the measured reflection coefficient of the antenna installed on the passenger van. The antenna has a reflection coefficient | S 11 | < −10 dB between 470 and 710 MHz.

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Figure 8.9 Measured radiation patterns of quarter glass antenna (vertical plane, 485 MHz). (From: [9].  2007 IEICE. Reprinted with permission.)

Figure 8.10 Measured reflection coefficient of quarter glass antenna. (From: [9].  2007 IEICE. Reprinted with permission.)

8.2.2 Thin Antenna A thin antenna was developed [11]. Figure 8.11(a) shows the configuration of a stubloaded folded dipole. The antenna has a pair of lines as stubs inside a folded dipole. The antenna has a width W of 20 mm and a length L of 240 mm. The width w l of the lines is set at 1 mm. The frequency bandwidth can be controlled mainly by the lengths L for the lower frequencies and l s for the higher frequencies. Figure 8.11(b) shows a prototype antenna printed on polyethylene terephthalate (PET) film with a thickness of 0.125 mm.

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Figure 8.11 Configuration of folded dipole antenna for digital TV reception: (a) wire model; and (b) prototype. (From: [11].  2007 IEEE. Reprinted with permission.)

The RF circuit connected at the feed point of the antenna consists of a balun, a filter, and a low noise amplifier (LNA). The LNA contributes to improving the noise figure of the RF subsystem, reducing the effect of loss from the RF cables between an antenna on a window glass or inside the spoiler, and a receiver inside the vehicle. The LNA has a series inductor so that the input impedance for the minimum noise may be adjusted to around 50⍀. The filter was inserted between the LNA and balun to avoid saturation of the LNA. The balun consists of a high pass filter, low pass filter, and T junction, and both filters have a fifth-order Butterworth function to cover the frequency range from 470 to 710 MHz. The balun transforms impedance from 200⍀ for the balanced port to 50⍀ for the unbalanced port. In terms of C/N, impedance matching can be achieved in the prototype antenna with the RF circuit. Figure 8.12 shows the measured voltage standing wave ratio (VSWR) of the stubloaded folded dipole antenna as a solid line. The VSWR is less than 2.6 over the frequency band from 470 to 710 MHz. The measured result for a folded dipole antenna without stubs is represented as a dot-dashed line for comparison. The comparison indicates that the stubs work in the higher frequency band. Numerical investigation was also carried out by the commercial simulator ‘‘FEKO’’ based on the method of moments. The calculated result represented by a dotted line agrees with the measured result. 8.2.3 Omnidirectional Pattern Synthesis Technique for a Car Installation positions of on-glass antennas are generally limited to the top of the front and rear windows for a sedan-type car, from the point of view for antenna performance,

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Figure 8.12 VSWR of folded dipole antenna. — — — : stub loaded (Mea.), – ⭈ – ⭈ – : without stub (Mea.), - - - - : stub loaded (Cal.). (From: [11].  2007 IEEE. Reprinted with permission.)

maintaining the driver’s view, and so on. The received signal level is unacceptably weak to the sides of the car when conventional loop antennas or dipole antennas are installed, because those antennas mainly have gain to the front and rear directions. A method of omnidirectional pattern synthesis was developed using four antennas. Figure 8.13(a) shows the configuration of a modified H-shaped antenna [12]. Two wires a-b-c-d-e and f-g-h-i-j are symmetrically placed around the center m of the antenna and connected by the wire c-m-h. The parts a-b-c and f-g-h of the two wires are longer than the parts c-d-e and h-i-j. A control system for combining received signals is shown in Figure 8.13(b). The received signals of the four antennas are downconverted, weighted, and combined. The weight vector for each signal is controlled, based on the MRC method. The four antennas mounted at the top of the front and rear windows are also depicted. A design concept of the coverage in the horizontal plane is summarized in Table 8.1. The mechanism of the antenna is described in Figure 8.13(a) and Table 8.1. The antenna has three resonant modes. Each resonant mode is excited when a part of the antenna has the length of a half resonant wavelength. The frequency band for digital terrestrial services is divided into three bands. The wire a-b-c-m-h-g-f is resonated at the low frequency, which is series resonance. Parallel resonance occurs at the middle frequency. The two wires a-b-c-d-e and f-g-h-i-j are resonated. Series resonance occurs again at the high frequency, in which the wire e-d-c-m-h-i-j is resonated. The radiation pattern of a single element is rotated clockwise with increasing frequency. The minimum level of relative amplitude increases when the maximum radiation is directed toward the x-axis direction. The four antennas are symmetrically placed around the x-axis and y-axis directions in Table 8.1, which is assumed to be equivalent to installation at the top of the front and rear windows. The symmetrical arrangement allows coverage of 360°. Figure 8.14(a–d) show typical measured radiation patterns of the four prototype antennas at 530 MHz. Each prototype antenna had gain to the x′-axis direction as well as to the y′-axis direction because the four antennas in free space had inclined figure-8

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Figure 8.13 Modified H-shaped antenna and control system for combining received signals for digital TV reception: (a) antenna element; and (b) control system. (From: [12].  2007 IEEE. Reprinted with permission.)

radiation patterns. The peak plot combined with the radiation patterns in Figure 8.14(a–d) is shown in Figure 8.14(e). A near omnidirectional pattern was achieved in the ␾ ′ plane. 8.2.4 Antennas Currently on the Market 8.2.4.1 Bended Folded Dipole and Monopole Antennas currently on the market are presented in this section. Figure 8.15(a, b) show a windshield-corner antenna installed in a car as well as the radiation pattern [13]. This is a kind of folded dipole antenna, and it is squarely bended for installation near the corner of the windshield. The antenna mainly has gain to the front direction. Figure 8.16(a, b) show a rear antenna and its radiation pattern [13]. A bended monopole antenna is attached on the rear window, where the defogger works as a ground plane. The antenna mainly

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Table 8.1 Concept Covering 360° in X-Y Plane Using Four Antennas Across Frequency Band for Digital TV Services

Source: [12].

has gain to the rear direction. Four antenna elements (two elements are on the front windshield and the other two elements are on the rear window) are used for reducing Doppler effect. The digital TV receiving system is shown in Figure 8.17. 8.2.4.2 Folded Monopole and Helical Rod Another set of antenna elements are presented in Figures 8.18 and 8.19 with radiation patterns. The front antenna is a folded monopole in Figure 8.18(b), and it is installed near the corner of the windshield in Figure 8.18(c); the rear antenna is a helical rod antenna in Figure 8.19(b). The rear antennas are installed near both side edges of the rear window in Figure 8.19(c). Radiation patterns in Figures 8.18(a) and 8.19(a) were measured at 470 MHz, inserting LNA with a gain of 12 dB. The front antenna mainly has gain to the sides of the car. The rear-left antenna covers not only the rear-left but also the rear-right and front-left, although the gain is lower than that of the front antenna element.

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Figure 8.14 Measured radiation patterns in ␾ ′ plane at 530 MHz. Modified H-shaped antennas were mounted at the top of the front and rear windows of car. Peak plot of combined pattern based on MRC method is also presented. (■ : ␪ ′ = 70°; ▲ : ␪ ′ = 80°; 䊉 : ␪ ′ = 90°). (a) Front left; (b) front right; (c) rear left; (d) rear right; and (e) peak plot. (From: [12].  2007 IEEE. Reprinted with permission.)

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Figure 8.15 Windshield-corner antenna (front-left): (a) radiation pattern; and (b) installed antenna. (From: [13].  2007 Fujitsu Ten. Reprinted with permission.)

Figure 8.16 Monopole on rear window (rear-right): (a) radiation pattern; and (b) installed antenna. (From: [13].  2007 Fujitsu Ten. Reprinted with permission.)

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Figure 8.17 Configuration of antenna-switching control. (From: [13].  2007 Fujitsu Ten. Reprinted with permission.)

Figure 8.18 Windshield monopole antenna (front-left): (a) radiation pattern (LNA was inserted); (b) antenna configuration; and (c) installed antenna. (Courtesy of Harada Ind. Co. Ltd.)

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Figure 8.19 Helical rod antenna (rear-left): (a) radiation pattern (LNA was inserted); (b) antenna configuration; and (c) installed antenna. (Courtesy of Harada Ind. Co. Ltd.)

REFERENCES [1] ARIB, ‘‘Receiver for Digital Broadcasting – ARIB STD-B21, 4.2,’’ October 16, 2003. [2] Sasaki, M., ‘‘Technologies and Services of Digital Broadcasting (12); Terrestrial Digital Television Broadcasting,’’ Broadcast Technology, No. 20, August 2004, NHK STRL, Tokyo, Japan, pp. 14–19, http:// www.nhk.or.jp/strl/english/index.html. [3] Takada, M., S. Kimura, and S. Moriyama, ‘‘Mobile Reception Performance for Digital Broadcasting System ISDB-T,’’ J. ITE, Vol. 54, No. 11, 2000, pp. 1590–1597. [4] Itoh, N., and K. Tsuchida, ‘‘HDTV Mobile Reception in Automobiles,’’ IEEE Proc., Vol. 94, No. 1, January 2006, pp. 274–280. [5] Sanda, K., et al., ‘‘Adaptive Beam Steering Reception System for ISDB-T Based on Pre-FFT Diversity Technique,’’ IEEE Trans. Consum. Electron., Vol. 52, No. 2, May 2006, pp. 327–335. [6] ITU-R Recommendation P.1546-1, ‘‘Method for Point-to-Area Predictions for Terrestrial Services in the Frequency Range 30 MHz to 3000 MHz,’’ April 2003. [7] Taniguchi, T., et al., ‘‘Portable Reception Experiments of Digital Terrestrial Television Broadcasting— The Comparison with Reception Characteristics Under Fixed Receptions,’’ ITE, Vol. 26, No. 53, BCS200236, July 2002, pp. 33–36 (in Japanese). [8] Itoh, N., ‘‘Receiving Over the Air HDTV Broadcasts in a Moving Car,’’ Nikkei Electronics, No. 856, September 15, 2003, pp. 156–164, and No. 857, September 29, 2003, pp. 152–157 (in Japanese). [9] Matsuzawa, S., K. Sato, and K. Nishikawa, ‘‘Study of On-Glass Mobile Antennas for Digital Terrestrial Television,’’ IEICE Trans. Commun., Vol. E88-B, No. 7, July 2005, pp. 3094–3096. [10] Johnson, R. C., and H. Jasik, Antenna Engineering Handbook, 2nd ed., New York: McGraw-Hill, 1984. [11] Iizuka, H., et al., ‘‘Stub-Loaded Folded Dipole Antenna for Digital Terrestrial TV Reception,’’ IEEE Antennas and Wireless Propag. Lett., Vol. 5, 2006, pp. 260–261. [12] Iizuka, H., et al., ‘‘Modified H-Shaped Antenna for Automotive Digital Terrestrial Reception,’’ IEEE Trans. Anetnnas Propag., Vol. 53, No. 8, August 2005, pp. 2542–2548. [13] Takayama, K., et al., ‘‘Development of Terrestrial Digital TV Broadcasting Receiver,’’ Fujitsu Ten Technical Report, Vol. 24, No. 1, 2006, pp. 43–51.

Chapter 9 Antennas for the Bullet Train Yoshiyuki Chatani

In this chapter, the bullet train antenna system in Japan is introduced. In Section 9.2, an outline of the LCX (leaky coaxial cable) communication system of Japan’s bullet train is described. In Section 9.3, the structure and principle of operation of the leaky coaxial cable is described, and two types of train antenna are shown. 9.1 INTRODUCTION Leaky coaxial cables are used for the train communication systems of Tokaido [TokyoOsaka (553 km)], Sanyou [Osaka-Fukuoka (624 km)], Tohoku [Tokyo-Hachinohe (632 km)], and Joetsu [Tokyo-Niigata (334 km)] Shinkansen [Japan Railways (JR) new bullet trains] [1–5]. They are coaxial cables whose outer conductors have slots radiating a part of the transmitted energy [6–9]. (An example is shown in Figure 9.1.) LCX cables were first developed for train communication systems in a tunnel for Tokaido and Sanyo Shinkansen. The advantages of LCX cable systems compared to radio communication systems were then realized. They offer more stable and better train communications compared to radio communication systems because LCX cables radiate weak electromagnetic energy and the environmental effects on their radiation characteristics are small [10, 11]. Once LCX cable is installed, their radiation characteristics are small [10, 11]. One LCX cable is installed along each railway; thus there are two LCX cables along the railway for inbound and outbound train communications, as shown in Figure 9.2. Trains have slot type or patch type antennas at the side of the body that transmits and receives electromagnetic energy to and from the LCX cables. 417

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Figure 9.1 LCX cable structure. (Source: [2].  IEICE.)

Figure 9.2 Outline of an LCX cable communication system. (Source: [2].  IEICE.)

9.2 TRAIN RADIO COMMUNICATION SYSTEMS The LCX communication system of Tohoku/Joetsu Shinkansen has been renewed from an analog system to a digital system in 2002 [12, 13]. There are 22 voice channels and 15 data channels. Train operation commands are sent through direct channels between

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the central command station in Tokyo and the train driver’s seat. Business commands are also sent through direct channels between the central command station and train conductors. Two public telephones in a train are connected to NTT public telephones all over Japan through JR’s telephone exchange networks. ⌸/4-shift QPSK modulation is adopted for data channels which are used for data network control and monitoring train running conditions in order to deal with an emergency. The data rate of data channels is 9.6 Kbps (12 ch) or 64 Kbps (3 ch). The bandwidths are 900 and 700 kHz from base station to trains and from trains to base station, respectively. The advantages of the LCX system are stable channel quality, efficient channel utilization, and small environmental effects. The last two advantages are, as mentioned, due to weak radiation from the LCX slots. Each base station is located at each train station and covers one service area (average 20 km). The LCX relay system is shown in Figure 9.3. When there is a communication problem in one LCX route, a route is changed to the other LCX at the relay station to prevent the trouble from spreading to other places. There is a maximum of 20 relay stations between the two train stations and four kinds of LCX cables with different coupling loss: #488 (loss = 75 dB), 487 (loss = 65 dB), 486 (loss = 55 dB), and #485 (loss = 50 dB), between a train antenna and an LCX cable. They are connected properly to reduce the received signal level change (less than 10 dB with respect to the center level) between the two relay stations, as shown in Figure 9.4. The maximum distance between the two relay stations is 1.5 km. Each relay station has a 400-MHz amplifier with 42-dB gain and the output is about 1W. The radio communication system of Tokaido/Sanyou Shinkansen is still an analog system, but will be renewed to a digital system in 2009. 9.3 ANTENNA SYSTEMS 9.3.1 LCX Cable The LCX cable shown in Figure 9.5 consists of an array of three slots with different inclination angles. The separation of two sets of three slots is P. They are designed to work in a 400-MHz band.

Figure 9.3 LCX relay system. (Source: [2].  IEICE.)

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Figure 9.4 LCX cable combination for signal attenuation grading. (Source: [3].  IEICE.)

Figure 9.5 LCX slot configuration. (Source: [3].  IEICE.)

LCX cables and train antennas are located as shown in Figure 9.6. There are two LCX cables along the railway and two train antennas on each side of the train. The LCX cables at the left side of the train toward the direction of travel (trains keep to the left in Japan) are usually used for communication from train to base station. When there is a fault, the communication channels are changed to the right-side cable. On the other hand, both sides of the LCX are used for communication from base station to train in order to improve radio network quality.

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Figure 9.6 LCX cable and train antenna location. (Source: [3].  IEICE.)

The vertical radiation pattern of LCX cables has a broad half-power beam width, and a measured vertical radiation pattern is shown in Figure 9.7. The maximum radiation of LCX cables in the horizontal direction occurs slightly backward (i.e., toward a transmitter direction), and the peak radiations of the LCX cables for inbound and outbound trains are slightly different. 9.3.2 Train Antenna As stated above, the maximum radiation angle of LCX cable changes where the LCX cable is laid on the inbound side or outbound side of the transmitter. Thus, the radiation

Figure 9.7 Vertical radiation pattern of an LCX cable. (Source: [2].  IEICE.)

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pattern of the train antenna has two peaks corresponding to the maximum radiation angle of the LCX cables as shown in Figure 9.8(a). On the other hand, a train antenna must have a broad half-power beam width in vertical plane [Figure 9.8(b)] corresponding to variation of LCX cable height and the inclination of the train on a curve. The half-power beam width of the vertical plane of the train antenna is about 110° and the gain of the antenna is about 7 dBi. Two types of the train antenna are used for Shinkansen. One is a slot array type antenna used in Tohoku and Joetsu Shinkansen, and the other is a short-patch array type antenna used in Tokaido and Sanyou Shinkansen. The slot array type antenna consists of four slot antenna units and one splitter/ combiner unit, as shown in Figure 9.9. Each slot antenna unit and splitter/combiner unit is connected by coaxial cables. Each slot antenna unit consists of a folded dipole and a matching section for matching at two different frequencies: 412 MHz (transmitter) and 452 MHz (receiver) [Figure 9.10]. There are two sets of slot array antenna on each side of a front car for space diversity receiving. The short-patch array type antenna consists of four transmit antenna elements, four receiving antenna elements, and two splitters/combiners. Each antenna element is a 1/4 wavelength short-patch antenna, as shown in Figure 9.11 for broad vertical pattern. Eight antenna elements, two splitters/combiners, and connecting cables are built in to one chassis, and the short-patch type antenna is more downsized than slot array type antenna (Figures 9.12 and 9.13).

Figure 9.8 Train antenna radiation pattern: (a) horizontal pattern; and (b) vertical pattern. (Source: [2].  IEICE.)

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Figure 9.9 Radiation element structure of a slot type train antenna. (Source: [3].  IEICE.)

Figure 9.10 Slot type train antenna configuration. (Source: [3].  IEICE.)

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Figure 9.11 Short-patch type train antenna configuration.

Figure 9.12 Radiation element structure of a short-patch type train antenna.

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Figure 9.13 Picture of train antenna for Tokaido-Shinkansen.

REFERENCES [1] Kishimoto, T., and S. Sasaki, ‘‘Train Telephone System,’’ Proc. IECEJ, Vol. 63, No. 2, February 1980, pp. 128–133. [2] Kishimoto, T., and S. Sasaki, ‘‘LCX Communication Systems,’’ IECEJ, Tokyo, 1982. [3] Watanabe, H., ‘‘Electronic Control and Communication System of Shinkansen,’’ IECEJ, Tokyo, 1982. [4] Hayashi, Y., ‘‘Train Radio System of Shinkansen,’’ Proc. IECEJ, Vol. 65, No. 5, May 1982, pp. 541–543. [5] Taguchi, K., et al., ‘‘Recent Train Radio Communication Systems,’’ Mitsubishi Electric Co. Technical Report, Vol. 61, No. 2, February 1987, pp. 33–37. [6] Amemiya, Y., ‘‘Surface Wave Coaxial Cable,’’ Proc. IECEJ, Vol. 48, No. 12, April 1965, pp. 131–142. [7] Mikoshiba, K., Y. Nurita, and S. Okada, ‘‘Radiation from a Coaxial Cable and Its Application to a Leaky Coaxial Cable,’’ Trans. IECEJ, Vol. 51-B, No.10, October 1968, pp. 499–505. [8] Yoshida, K., ‘‘New Communication Systems by Leaky Coaxial Cables,’’ Proc, IECEJ, Vol. 55, No. 5, May 1972, pp. 655–663. [9] Cree, D. J., and L. J. Giles, ‘‘Practical Performance of Radiating Cables,’’ Radio & Electronic Engineering, Vol. 45, No. 5, May 1975, pp. 215–223. [10] Mikoshiba, Y., S. Okada, and S. Aoki, ‘‘Near Electromagnetic Fields Around a Leaky Coaxial Cable,’’ Trans. IECEJ, Vol. 54-B, No. 12, December 1971, pp. 789–796. [11] Delogne, P., Leaky Feeders and Surface Radio Communication, London, U.K.: Peter Peregrius, Ltd., 1982. [12] Atsuzawa, M., Y. Nishimura, and K. Yoshida, ‘‘Development of Digital Train Radio Communication System of Tohoku-Joetsu Shinkansen,’’ JR EAST Technical Review, No. 5, August 2003, pp. 59–64. [13] Shigeru, F., et al., ‘‘Tohoku-Joetsu Shinkansen Digital Train Radio Communication System,’’ Mitsubishi Electric Co. Technical Report, Vol. 78, No. 2, February 2004, pp. 36–40.

Chapter 10 Antennas for ITS Hiroyuki Ohmine

This chapter presents antennas for Intelligent Transportation Systems (ITS), especially a 5.8-GHz active system of dedicated short range communication (DSRC), which is the key to the development of ITS. A special design method that illuminates a specified area on a lane at a tollgate and its design applications are described. Additional field strengths, including reflections and diffractions from some obstacles, are also indicated for the use of practical application. Some advanced topics, such as applications of DSRC of a nonstop parking access control system, bun location system, and probe system, are also addressed. 10.1 GENERAL Intelligent Transportation Systems have been developed as a solution for various traffic problems, including accidents, traffic jams, and exhaust emissions by the introduction of electronics and communication and information systems, and various research and development work has been promoted in many countries of the world. Through the abrupt progress of information and communication technology (ICT), vehicles are also equipped with ICT. In Japan, the total accumulated number of car navigation units has exceeded 26 million as of the the time of this writing. Among these, 18 million units have been equipped with the Vehicle Information and Communication System (VICS). Further, the number of on-board equipment for electronic toll collection (ETC) in the market had exceeded 19 million units by September 2007. A 5.8-GHz active system of dedicated short range communication is treated as the key to the deployment of ITS and has been 427

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introduced for ETC as the primary technology. This DSRC can allow a massive volume of data for high-speed interactive communication. These recent trends are regarded with the understanding that the vehicle is not merely a means of transportation but is a ubiquitous mobility space. At first, the ARIB STD-T55 [1] standard was established only for the ETC system, which is the most typical application of road-to-vehicle communication using the ASK method. ARIB STD-T75 [2] was enhanced with the addition of the QPSK format to enable higher bit rate transmission, which can be applied to a wider service variation, including information download. This DSRC system was authorized by the International Telecommunication Union-Radiocommunication Sector (ITU-R) as the international standard in May 2000. These technologies became available for multipurpose services through revision of the Japanese ministerial ordinance related with the radio law in April 2001. The DSRC applications [3, 4] as shown in Figure 10.1 are expected to include parking control, information services at parking areas SA/PA and Michi-no-Eki (roadside rest area), downloading services of updated navigation maps, and music and video at convenience stores and other places, which are offered by both public and private vendors. In this chapter, antennas using DSRC systems (including the ETC system) are described as a current key component of ITS. In designing antennas, special considerations are required individually to satisfy the specific requirements for the systems. For instance, with ETC systems, the radiation pattern should illuminate a specified area on the lane at the tollgate. Such fundamental antenna design for DSRC and its applications is presented.

Figure 10.1 DSRC applications.

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10.2 ANTENNA DESIGN 10.2.1 Communication Beam Coverage Careful consideration of the design is needed for the required antenna beam coverage for DSRC systems, including the ETC system. In the design of antenna beam coverage, not only covering the required coverage, but also considering interference to another lane or another DSRC system are needed. This section describes requirements of antenna beam coverage of ETC tollgate and general DSRC systems. ETC is an automatic toll collecting system that enables vehicles to pass through a tollgate without stopping for toll payment. The toll collection is processed automatically by means of wireless communications between a vehicle and the roadside equipment (RSU) placed at tollgate facilities. Drivers benefit from the ETC system because they are not required to stop at the tollgate. Figure 10.2 shows the outline of the ETC system. The communication is exchanged through the RSU installed on each lane at the tollgate, and with the on-board unit (OBU) in the vehicle to collect the toll automatically. This system must be equipped with the functions to retry communication to prevent communication errors and to identify one vehicle from a continuous stream of subsequent ones. This also

Figure 10.2 ETC system.

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improves services for the drivers and reduces the costs of the toll management. The ETC system is composed of the wireless communication system and the toll processing system. The tollgate antenna is designed to illuminate the communication area uniformly and to have low sidelobes to prevent interference between the adjacent lanes. This antenna pattern must be precisely designed because interference causes the problem of toll collection error. The tollgate antenna is settled over about 5m from the lane. The communication coverage area is specified for an area 4m long and 3m wide at an elevation of 1m over the lane, as illustrated in Figure 10.3. Circular polarization is used because it can decrease the interference caused by reflected waves which have inverse polarization. Right-hand circular polarization is used in the Japanese system, while left-hand circular polarization is used in U.S. and European systems, and interference between two different systems is eliminated intrinsically by different polarization. The specifications of the ETC system with regard to RSU and OBU are described in Table 10.1. General DRSC on a road system is shown in Figure 10.4. The communication system consists of the wireless system between the vehicle and the roadside, its control system, and the data processing system pertaining to the toll processing. In this kind of application, spot-area communications are performed primarily between the roadside equipment and the vehicles. In locating DSRC beacons, careful design for the layout of beacons to avoid interference is needed because severe interference would occur among self-system and/or different systems depending on conditions of the location. Also, consideration is needed for the distortion of the communication area due to reflections and diffractions from obstacles such as sidewalls, buildings, road surface, and vehicles. However, specific and high-class

Figure 10.3 Coverage area.

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Table 10.1 ETC Specification

Frequency Modulation Data rate Transmit power Sensitivity level Nonresponding level

RSU

OBU

5.795 or 5.805 GHz ASK 1.024 Mbps < 10 mW < −65 dBm < −75 dBm

5.835 or 5.845 GHz ASK 1.024 Mbps < 10 mW < −60.5 dBm < −70.5 dBm

Figure 10.4 DSRC communications.

beam shaping antenna design such as the ETC gate may be difficult, because of the antenna size, space to locate, and the cost.

10.2.2 Antenna Fundamental Design For design methods that realize the required beam coverage, several methods are used to synthesize the arbitrary shaped coverage, such as the Remez algorithm, the least square, the Fourier transform method, and the Woodward-Lawson method [5, 6]. Here, the WoodwardLawson method, which is known as practical use, and its application examples are described. Though a single antenna element has a limited radiation pattern, with the use of an array of several antenna elements, it is possible to change the radiation patterns and beam width. In general, the array patterns are controlled by the choice of the element, such as a dipole, a horn, and a patch, and by the geometry of the array and the excitation (amplitude and phase) of each element.

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Here, the antenna arrays are restricted to linear arrays where the radiated fields depend only on the ␪ coordinate, as shown in Figure 10.5. The electric field in far zone is represented by: E ␪ (␪ ) ≈ E ␪ 1 ⭈ AF

(10.1)

where E ␪ 1 is the electric field produced by the element at the origin of the coordinate system and AF is known as the array factor and is given by N

AF =

∑

n =1

C n e −j(n − 1)␺

(10.2)

where ␺ = kd cos ␪ + ␤ n . C n , ␤ n represent amplitude and phase of the n th element with respect to the origin, respectively. The process to obtain the position and excitation of the elements in order to get some desired radiation patterns is known as the synthesis of an array. Woodward-Lawson Method This is a beam-shaping method based on the sampling of the desired AF. For each sampling, a set of uniform amplitude and progressive phase excitation signals are calculated. Such excitation produces a sinc (x)-like pattern with the major lobe just on the direction of the sample points, as shown in Figure 10.6. Minor lobes and nulls are also produced, but

Figure 10.5 Linear array.

433

Figure 10.6 Sample points.

sampling points are chosen so that the contribution in the sampled direction comes from one set of excitation only. The pattern function of each sample point can be written as [5] sin f m (␪ ) = a m

冋

N−1 kd (cos ␪ − cos ␪ m ) 2

(N − 1) sin

冋

册

册

1 kd (cos ␪ − cos ␪ m ) 2

(10.3)

Excitation of the array is obtained by adding all partial excitation signals. The total array factor can be written as a superposition of M terms each of (10.3).

M

AF =

∑

m =1

冋

册

N−1 kd (cos ␪ − cos ␪ m ) 2 am 1 (N − 1) sin kd (cos ␪ − cos ␪ m ) 2 sin

冋

册

The angles of sample points are selected as [5]

␪ m = cos−1

冉

m␭ (N − 1) d

冊

(10.4)

The normalized current excitation of each element is given by [5] In =

1 N

M

∑

m =1

a m e −jkd(N − 1) cos ␪ m

(10.5)

For example, the number of sampling points M is set to 11, as shown in Figure 10.6.

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The angles of the sample points are given according to (10.4), and the excitation coefficients at the sample points are selected from Table 10.2 for the desired shaping beam. Figure 10.7 shows the demonstration of each pattern of each sample point from (10.4) and Table 10.2. At each sampling point, all the composing functions are zero, except the one sample function. Thus the value of the desired pattern at each sampling point is determined by only a value of one sample function. Table 10.2 Angles and Excitation Coefficients at Sample Points (Number of Elements N: 11, Element Distance: 0.5 Wave Length) m

␪ m (Degree)

am

−5 −4 −3 −2 −1 0 1 2 3 4 5

180 143.1 126.9 113.6 101.54 90 78.5 66.4 53.1 36.9 0

0 0 0 1 1 1 1 1 0 0 0

Figure 10.7 Component of each function.

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As a result, the synthesized radiation pattern obtained from composing each sample function is shown in Figure 10.8, where it is compared with the desired pattern. A good reconstruction is indicated. Design Example The design examples of pencil beam, sector beam, and cosecant beam antenna patterns using the Woodward-Lawson method are described next. Figure 10.9 shows the pencil beam radiation pattern with varying the number of elements of a uniform array. Beam width is adjusted by selecting the number of elements. If low sidelobe is required, excitation of amplitude, such as Taylor distribution, will be considered for the design [6]. The design examples of the sector beam antenna pattern are indicated in Figure 10.10, with varying the number of elements. The amplitude and phase of each element by the Woodward-Lawson method is shown in Table 10.3. It is found that shapes are improved by increasing the number of elements. Another design example of the cosecant beam is indicated in Figure 10.11, and the amplitude and phase of each element by the Woodward-Lawson method is shown in Table 10.4. 10.2.3 Microstrip Antenna Design A thin and lightweight microstrip antenna (MSA) can be easily produced at low cost in mass production because the MSA is formed by etching one surface of dielectric substrate

Figure 10.8 Synthesized beams.

436

Figure 10.9 Pencil beam example.

covered with copper foil on both surfaces, as shown in Figure 10.12. These features of the MSA are suitable for DSRC antenna requirements, and the MSA is one candidate for elements of several antenna applications. In this section, the fundamental design of the MSA for DSRC application, including the ETC tollgate antenna, is described. The resonant frequency f of circular MSA is obtained by [7, 8] f=

冋

ae = a 1 +

299.8 ⭈ x 11 2␲ a e √⑀ r

2t ␲ a⑀ r

冉

ln

(GHz)

冊册

␲a + 1.7726 2t

(10.6) 1/2

(mm)

(10.7)

where x 11 = 1.8411 (TM 11 mode). In the design of an MSA array, the desired amplitude and the phase of each element is fed by the feed circuit. Two types of feed design are described in Figure 10.13. Figure 10.13(a) shows the two-layer configuration so that it has features that eliminate the

437

Figure 10.10 Sector beam example.

Table 10.3 Amplitude and Phase of Each Element (Element Spacing: 0.65␭ ) Element No.

Amplitude (dB)

Phase (Degree)

±1 ±2 ±3 ±4 ±5 ±6

0.0 −11.5 −15.0 −22.5 −23.0 −27.5

0 0 180 180 0 0

radiation from the feed line to radiation direction and has enough space for the feed line, while the configuration is complicated due to two-layer substrate and pin feed connection. On the other hand, the one-layer configuration in Figure 10.13(b) has features of a simple configuration and can achieve low cost production. However, it has the possibility that radiation from the feed line affects the radiation distortion. Further deviation from the desired amplitude and the phase will arise due to the coupling of the feed lines that are arranged in close space because the space for the feeder circuit is limited by the coplanar feed system. As a result, the layout of the feed circuit should be properly designed. There are two ways of generating circular polarization. One is two-point feed with 90° out-of-phase as shown in Figure 10.14(a), and the other is one-point feed with detuning

438

Figure 10.11 Cosecant pattern synthesized (10 elements).

Table 10.4 Amplitude and Phase of Each Element (Element Spacing 0.5␭ ) Element No.

Amplitude (dB)

Phase (Degree)

1 2 3 4 5 6 7 8 9 10

−2.0 −0.2 0.0 −2.7 −10.0 −12.8 −8.0 −10.0 −17.0 −6.8

45 −55 −160 90 0 −15 −100 170 110 100

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Figure 10.12 Configuration of circular MSA.

Figure 10.13 (a, b) Feed line configuration.

orthogonal two modes as shown in Figure 10.14(b, c). Two-point feed has wider band width, while a large space is needed for the layout of the feed circuit. On the other hand, one-point feed configuration can achieve a very simple configuration, while frequency performance of the axial ratio is relatively narrow. Since the MSA is manufactured by etching the printed circuit board, its center frequency may deviate from the design value under the following conditions in mass production:

440

Figure 10.14 (a–c) Generating circular polarization.

1. Irregular size of the antenna element; 2. Error of the dielectric constant of the printed circuit board. Therefore, broadening the bandwidth design is required when considering mass production. The frequency response of the axial ratio of the one-point feed MSA is shown in Figure 10.15 [9]. It is found that the bandwidth of the axial ratio is not so wide. As for the technique to improve the axial ratio, the paired-elements array consisting of two circular polarization antenna elements, which are arranged at a left angle in the space and phase difference by 90° to achieve the wider frequency band, has been proposed as shown in Figure 10.16 [7]. This paired element arrangement can also have the feature, which

Figure 10.15 Frequency response of the axial ratio.

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Figure 10.16 Paired-element configuration.

makes the impedance wider at about twice the bandwidth compared to a single-element configuration in the typical case. In the typical case, twice of bandwidth can be enlarged by applying the paired-element configuration. Figure 10.17 shows the circularly polarized radiation pattern of one-element MSA. For a comparison, the calculated radiation patterns with/without considering the ground plane are also indicated in the figure. Some distortion can be seen due to the ground plane effect and almost the same tendency was obtained between calculated and measured radiation patterns. Next, an example design of the sector beam radiation pattern produced by 64-element MSA is depicted in Figure 10.18. The design conditions of amplitude and phase of 8 elements are the same as in Table 10.3. Also, a comparison of the designed and measured sector beam radiation pattern of 64-element MSA is shown in Figure 10.19. The circularly polarized circular MSA with one-point feed circuit and one-layer coplanar feed circuit are adopted to reduce the cost and to achieve simple configuration. This figure shows that the 3-dB beam width is over 32° and the sidelobe level is around −30 dB. 10.2.4 Communication Coverage The electric power field strength is determined from the transmitting power, the antenna radiation pattern of the beacon, and span loss due to different distances from the beacon and communication points, as shown in Figure 10.20. The field strengths including span loss are calculated in Figure 10.21(a, b), the loss budget of which is cited from Table 10.1. Figure 10.20(a) shows the field strength toward the vehicle driving direction, and Figure 10.20(b) shows that of the vehicle lateral direction.

442

Figure 10.17 Radiation pattern of one-element MSA.

The communication area is calculated from the field strength to which the vehicle can respond. 10.2.5 Multiple Reflections The radiation patterns are affected by the reflections and diffractions from the tollgate, the road wall, the road surface, and the bodies of vehicles in the actual field as shown in Figure 10.22. These multiple reflections should be considered when designing the actual antenna field. When the dimensions of the reflected object are large compared to the wavelength, high-frequency asymptotic techniques can be used to analyze many otherwise not mathematically tractable problems. One such technique is the geometrical theory of diffraction (GTD), which is an extension of the classical geometrical optics (GO; direct and reflected rays), and it overcomes some of the limitations of GO by introducing a diffracted field. Total field E t can be obtained from sum of direct field E i, reflected field E r, and diffracted field E d. Et = Ei + Er + Ed

(10.8)

The initial value of the field on a diffracted ray is determined from the incident field with the aid of an approximate diffraction coefficient (which, in general, is a dyadic

443

Figure 10.18 Radiation pattern of the sector beam.

for electromagnetic fields). When the source is located a distance s′ from the point of diffraction and the observations are made at a distance s from it, as shown in Figure 10.23, the diffracted field can be written, in general, as [10] E d (s) = E i (Q) D

√

s′ e −jks s (s′ + s)

(10.9)

where E i (Q) is the incident field at the point of diffraction, and D is the diffraction coefficient (usually a dyadic). 10.3 FIELD STRENGTH IN COMMUNICATION AREA 10.3.1 Multiple Reflections from Canopies The tollgate has special structures, such as canopies for shelter against rain and pillars to support them. The measured field distribution of an actual tollgate turned out to be different

444

Figure 10.19 Sector beam radiation pattern.

from that of the test site. This difference was assumed to have been caused by the multiplereflection of waves between the road surface and the canopy and other obstacles. Figure 10.24 shows how the analysis model represents the multiple reflections between the road surface and the canopy. Figure 10.25 shows analysis results with/without multiple reflections. From this figure, it is found that field strengths are changed from multiple reflections and multiple reflections propagate backwards against the driving direction, which causes interference to other vehicles. Figure 10.26 shows the measurement results of the field strength of the ETC communication area, and it was found that the backwards level of the ETC communication area is increased due to multiple reflections and that this tendency of an increasing level corresponding to the simulation results as shown in Figure 10.25. The detailed configuration of the canopy is incorporated into the analysis to find the reflection point, and more precisely to find how to mitigate the multiple reflections from canopies, as shown in Figure 10.27. Reflected points are specified precisely through this analysis. 10.3.2 Mitigation Using an Absorber at the ETC Gate One method of mitigating negative effects from multiple reflections is to attach an absorbing material to reflection points. To obtain the requirements for the absorber perfor-

445

Figure 10.20 Communication coverage area.

Figure 10.21 Field strengths: (a) vehicle driving direction; and (b) vehicle lateral direction.

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Figure 10.22 Multiple reflections.

Figure 10.23 Diffraction from the wedge.

Figure 10.24 Analysis model for multiple reflection.

mance, field strength simulation with different angles of incidence has been done with the condition that the receiving power of the vehicle terminal is less than −75 dBm and corresponds to nonreaction level. The obtained results of the requirement of absorbed power with different angles of incidence are depicted in Table 10.5 [11]. Further, the actual absorber should meet the following requirements: lightweight, strong enough to bear wind pressure, and a thickness of several millimeters or less [12].

447

Figure 10.25 Analysis result with/without multiple reflections.

Figure 10.26 Measurement result of field strength.

448

Figure 10.27 Reflection points at the canopy.

Table 10.5 Requirement for Absorber Performance Angle of Incidence 0.0° 45.0° 47.5° 50.0°

≤ < <
0.7␭ . The infinite ground plane gives the broadest beam, which approaches −6 dB at the horizontal plane. These results show that the assumption of an infinite ground plane in approximate analysis of MSAs will have a serious effect on the correct prediction of the radiation patterns, particularly for angular ranges beyond 45° off the main beam axis. For a rectangular microstrip patch with a finite ground plane, Huang [61] presented radiation patterns calculated on the basis of slot theory and the uniform GTD. Both the E- and H-plane radiation patterns of a rectangular microstrip patch, whose geometry is illustrated in Figure 11.54, are shown in Figure 11.55 with the measured results. The

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Figure 11.49 Dual-frequency quadrifilar helical antennas: (a) enclosed type; and (b) piggyback type.

antenna dimensions in inches are given in the figure, and the operating frequency is 2.295 GHz (wavelength = 5.146 inches). The agreement between the measurement and the calculation is quite good. From the figure, it can be seen that the gain reduction for the low elevation angles is relatively large, particularly in the H-plane. These characteristics seem to be similar to those of the circular microstrip patches; therefore, for GPS application, especially for its operation in the low elevation angle, a QHA may be more preferable than an MSA. 11.6 MULTIBAND ANTENNAS FOR FUTURE GPS/ITS SERVICES 11.6.1 Slot Ring Multiband Antenna for Future Dual Bands (L1 , L2 ) GPS Recently, wireless technologies have advanced significantly. In particular, advances in the field of personal wireless technologies such as cellular phones, ITS, and GPS are remarkable. For instance, presently, one frequency band at L1 band (with a center frequency of 1.57542 GHz) is assigned for GPS. In the near future, the L2 band (with a center frequency of 1.2276 GHz) will be added so that two-frequency operation will be initiated. The measurement error, currently of the order of 10m, will be improved to about 10 cm, and the service in wider areas is expected [62]. One of the techniques important to further advances for these services is reduction of the size and profile of antennas. In addition, multiband operation covering many frequency ranges with one antenna [63–66] and simplification of the antenna feed are indispensable. However, research on antennas to take account of these requirements simultaneously has just begun [67]. To the authors’ knowledge, the only research from

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Figure 11.50 Amplitude and phase patterns of the enclosed-type QHA (the L 1 volute is inside the L 2 volute) at L 1 (– –), at L 2 (- - - -), and of a single QHA (—). (From: [55].  1990 IEEE. Reprinted with permission.)

such a point of view is [68]. This type of antenna requires studies of size, structural complexity, and the need for a matching circuit. The present authors have noted the issues described above and have studied a multiband antenna with one simple feed [69–74]. In [74], an elliptical double slot ring antenna with a single feed is proposed. It is reported that this antenna can achieve excellent return loss and axial ratio characteristics at three frequency bands of the next generation GPS (L1 , L2 , and L5 (with a center frequency of 1.17645 GHz). However, since this antenna has bidirectional radiation characteristics,

509

Figure 11.51 Amplitude and phase patterns of the piggyback-type QHA (the L 1 volute is inside the L 2 volute) at L 1 (– –), at L 2 (- - - -), and of a single QHA (—). (From: [55].  1990 IEEE. Reprinted with permission.)

some provision is needed if a unidirectional radiation pattern is required. In general, the slot antenna has a number of advantages such as a thin profile, ease of fabrication, and ease of integration with electronic components. Unlike the patch antenna, however, there are cases in which bidirectional radiation takes place, as in [74]. To make the radiation pattern of the slot antenna unidirectional, there is a method, as an example, to place a metallic reflector at a certain fixed location away from the antenna, as reported in [75]. When a metallic reflector is used, its fixed location is ␭ /4 of the desired frequency. For

510

Figure 11.52 Circular microstrip antenna with a finite ground plane. (From: [60].  1986 IEEE. Reprinted with permission.)

instance, at the center frequency 1.57542 GHz of the L1 band of GPS, ␭ /4 is 47.57 mm, so that the antenna becomes rather thick. Upon the above circumstance, in this section, a new type of the slot ring multiband antenna for dual bands (L1 , L2 ) GPS operation is described The targeted frequency ranges are the L1 band (1.5742 ± 0.0001023 GHz), presently used by GPS, and the L2 band (1.2276 ± 0.0001023 GHz), expected to be introduced shortly (with satellites launched after the end of 2006). The target characteristics in both bands are return loss less than −10 dB, an axial ratio of 3 dB (right-handed circular polarization), and an F/B of the radiation characteristics of more than 5 dB (0.56). 11.6.1.1 Antenna Structure The dual frequency circularly polarized slot antenna is a single feed multiband antenna with small size and a low profile, intended to provide the characteristics described above. The structure of the antenna and the coordinate system are presented in Figure 11.56. As shown in the figure, a slot is cut in such a way that the minor axis of the inner elliptical patch (with a major axis of M j and minor axis of M i ) is oriented at 45° with respect to the Y-axis relative to the center of the outer circle (with a radius of O r ). Further, inside the elliptical patch, a circular slot is cut with an outer circle (with a radius of R o ) and in inner circle (with a radius of R i ) and concave perturbations (P h × Pw ) at 45° inclined with respect to the Y-axis, attempting to produce right handed circular polarization at two

511

Figure 11.53 The radiation patterns of a circular patch with different ground plane radius g. (From: [60].  1986 IEEE. Reprinted with permission.)

512

Figure 11.54 Rectangular MSA with a finite ground plane. (From: [61].  1983 IEEE. Reprinted with permission.

frequencies. This antenna is fed by a stub placed on the underside of the substrate and a ground plane. The stub is placed at a location offset (S p ) in the −X direction from the center while the stub length is L m . Also, the stub width is Wm = 6.8 mm, the substrate height is H d = 2.4 mm, the substrate length is Wx = Wy = 100 mm, and the relative permittivity of the substrate is ⑀ r = 2.55. As an example, a square reflector with dimensions of R b × R b = 120.0 × 120.0 mm is placed by 8.0 mm from the underside of the antenna. As described previously, the undesired reflection to the backside can be suppressed if a reflector is placed ␭ /4 behind it; the antenna turns out to be rather thick. Therefore, in this antenna, the reflector used to suppress backside radiation is placed at H 8 = 8.0 mm (about 0.33␭ ) as an example. However, as the influence of this reflector is extremely strong, two types of slots, a cross-slot and a ring-slot, are cut on the reflector. The former is for L1 band and the latter is for L2 band. 11.6.1.2 Design After adjustment, it is found that by setting each structural parameter as given in Table 11.9, a return loss of less than −10 dB and an axial ratio of less than 3 dB (right-handed circular polarization), and an F/B of the radiation characteristics exceeding 5 dB can be realized in the two frequency ranges corresponding to L1 and L2 bands, as shown in Figure 11.57. The gain at the center frequencies for both bands are listed in Table 11.10. In comparison to the case without a reflector, the gain is increased by 2.2 and 2.35 dB, respectively, at 1.575 GHz (L1 ) and 1.227 GHz (L2 ). Also, Figure 11.58 shows the entire view, a plan view of the tested antenna, and a plan view of the reflector. The used substrate was PTFE (⑀ r ∼ 2.55). The spacer between

513

Figure 11.55 (a) E-plane and (b) H-plane radiation patterns of a rectangular MSA. (A = 2.126 in, B = 1.488 inches, e = 10.5 inches, h = 14.0 inches, substrate = 0.125 inch, ⑀ r = 2.55, frequency = 2.295 GHz). (From: [61].  1983 IEEE. Reprinted with permission.)

514

Figure 11.56 Antenna configuration.

Table 11.9 Structural Parameters Parameters

Size (mm)

Radius Major axis ellipse Minor axis ellipse Radius Perturbation Stub length Stub offset Stub width Substrate length Substrate height Substrate-reflector distance Circular slit radius Cross slit length Cross slit width Reflector length

O r : 40.8 M j : 32.8 M i : 31.6 R o : 24.4 P h × Pw : 15.8 × 2.6 L m : 27.2 S p : 5.6 Wm : 6.8 Wx , Wy : 100, 100 H d : 2.4 Hs : 8 C o , C i : 32.4, 30.0 C l : 56.0 C w : 11.2 R b : 120.0

515

Figure 11.57 Calculated antenna characteristics: (a) axial ratio, (b) return loss characteristics, (c) radiation patterns at L 1 band, and (d) radiation patterns at L 2 band.

516

Table 11.10 Gain at the Center Frequencies for Both Bands Frequency (GHz)

Gain (dBi)

1.575 (L1 ) 1.227 (L2 )

6.64 7.09

Figure 11.58 Experimental antenna model: (a) entire view, (b) plan view of the antenna, and (c) plan view of the reflector.

517

the antenna and the reflector was acrylic. Figure 11.59 shows an example of the measured return loss and radiation patterns (in the X-Z plane); from the figure, it is seen that the return loss is less than −10 dB in both bands (L1 and L2 ) and that overall the results are similar to the results of the analysis. With regard to the radiation characteristics, the F/B of the E␾ component at 1.575 GHz does not completely satisfy the target value of 5 dB (0.56). Overall, the results have the same tendency as those of the analysis. 11.6.2 Microstrip Multiband Antennas for GPS, VICS, and DSRC Recently, applications for communications systems used in vehicles have greatly increased. For instance, GPS, vehicle information and communication system (VICS), and electronic toll collection (ETC), have spread widely in Japan. In addition, IMT-2000, wireless LAN (WLAN), and dedicated short range communication system (DSRC) are under consideration to install in a car. These communications systems require different types of antennas because the antenna characteristics are completely different for each communication system. Therefore, a lot of antennas are needed for cars. One solution to this problem is to use a multiband operation antenna [76]. Many techniques for multiple operation antennas are reported so far. Most of these antennas are concerned with the microstrip antenna. One of the techniques used is aperture-coupled stacked microstrip antenna [77, 78]. The aperture-coupled microstrip antenna has some advantages such as wide bandwidth and noncontacting feed transition. However, the feeding structure of this type of microstrip antenna is complicated in fabrication, in addition to the increase of antenna noise temperature due to feeding circuit loss. This antenna noise temperature degrades the receiving performance of satellite communications. A single layer and single feed microstrip antenna is also proposed [79]. This antenna has switchable slots and radiates a circularly polarized wave. The disadvantage of this antenna lies in the complicated feeding structure which employs either pin diodes or MEMS switches to realize dual band operation [80]. These switches also cause high antenna noise temperature owing to feed loss. A more effective design is proposed in [81] where a shorted annular patch antenna is demonstrated. One example of a triple-band antenna for GPS, VICS, and DSRC applications is introduced in this section. Some specifications for GPS, VICS, and DSRC are listed in Table 11.11. Concerning the specification of GPS, it is described in the previous section. Special features of the antenna for this application lie in the polarization characteristics of the antenna. 11.6.2.1 Antenna Structure As listed in Table 11.11 the target systems require linear and circular polarization characteristics, in addition to the frequency differences. To count the problem, the technique of combining a circular microstrip patch antenna with loop antennas is used because the both antennas easily produce circular and linear polarized waves. Figure 11.60 shows the

518

Figure 11.59 Measured results: (a) return loss; (b) radiation pattern at L 1 ; and (c) radiation characteristics at L 2 .

519

Table 11.11 Specifications for GPS, VICS, and DSRC

Frequency Return loss Antenna gain Polarization Axial ratio

GPS

VICS

DSRC

1.57542 GHz ± 1.023 MHz < −10 dB > 0 dBi Right-handed circular < 3 dB

2.4997 GHz ± 42.5 kHz < −10 dB > 0 dBi Linear N/A

5.775–5.845 GHz < −10 dB < 10 dBi Right-handed circular < 3 dB

Figure 11.60 Structure of triple-band patch antenna.

structure of the triple-frequency band operation microstrip patch antenna. This antenna consists of a dielectric substrate and a ground plane. One patch and two loop antennas are etched on one side of the substrate so as to be coincident with each antenna center. An original feed point is located at the patch antenna, and the two loop antennas are fed through arms which are pulled out from the patch and an inner loop antennas, respectively. 11.6.2.2 Principal of Operation The microstrip patch antenna is for DSRC application. Therefore, the antenna is designed so as to radiate a circularly polarized wave using perturbation technology. To perturb the patch, reactively loaded elements are attached at the edge of the patch. The loop antenna located between the patch and outer loop antennas radiates a linearly polarized wave. To produce the linearly polarized wave, the circumferential length of the loop is almost the

520

same as a one-wavelength loop antenna. The feed position and the arm configuration from the patch are properly chosen to match the input impedance of the loop. The outer loop produces a circularly polarized wave. To radiate the loop circular polarization easily, a reactively loaded element is attached to the loop. The feed point of the loop is selected adequately considering the output point of the inner loop and the arm configuration. 11.6.2.3 Characteristics of Triple-Band Antenna Figure 11.61 shows the fabricated triple-band operation antenna. The thickness of the substrate is 4 mm and relative permittivity is 2.55. The loss tangent of the dielectric substrate, tan ␦ , is 0.001. In this case, the size of this antenna is the diameter of 100 mm. Figure 11.62 shows the calculated and measured results of the return loss characteristics of the antenna. Good return loss characteristics less than −15 dB are obtained in each frequency band of GPS, VICS, and DSRC. Also, the calculated results agree well with measured results. Particularly, in the DSRC band, 5.8 GHz, excellent return loss characteristics are achieved over the wide frequency range from 5.3 to 6 GHz. The reason why such wide frequency characteristics are obtained seem to be that the two loop antennas act as if they were parasitic elements. Figure 11.63 shows both calculated and measured results for frequency response of the return loss and axial ratio characteristics in the 5.8-GHz band. In this figure, it is found that although the wideband characteristics are obtained for return loss, axial ratio characteristics are narrow bandwidth. Some examples

Figure 11.61 Fabricated antenna.

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Figure 11.62 Return loss characteristics.

Figure 11.63 Return loss and axial ratio in the 5.8-GHz band.

of the calculated radiation patterns are shown in Figure 11.64 for GPS, VICS, and DSRC applications comparing with the measured patterns. The antenna gain is almost 5.6, 6.2, and 5.4 dBi in 1.5, 2.5, and 5.8-GHz bands, respectively, in both measured and calculated results.

522

Figure 11.64 Example of radiation patterns: (a) 1.5-GHz band (GPS), (b) 2.5-GHz band (VICS), and (c) 5.8-GHz band (DSRC).

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11.7 SATELLITE CONSTELLATION SYSTEMS AND ANTENNA REQUIREMENTS In the 1980s, there were significant developments in the creation of LEO/MEO satellite constellation systems, as noted in the introduction to this chapter, and the practical realization of such systems has been commercially demonstrated. Global navigation systems such as GPS and GLONASS are, of course, well established, but these are relatively low information rate systems. The challenge to satellite communication systems designers has been immense, as have the projected development and manufacturing costs [82, 83]. Antenna designers, in particular, have introduced state-of-the-art electronically controlled spot-beam arrays to satisfy the downlink demands, and the design of compact antenna elements for mobile handset terminals has emerged as a critical design issue [84]. In this section, the main satellite constellation systems and their system parameters are outlined to highlight the antenna requirements of significance. 11.7.1 Constellation Systems and Demands on Antenna Design Iridium, Globalstar, and ICO are the main satellite constellation systems that are designed to provide personal voice communication services. A summary of some of their parameters is described in Table 11.12. These systems have received much attention in both the technical literature [82, 85–87] and general interest literature. Other LEO satellite constellation systems include Orbcomm, which offers a service for short messages at low cost. The deployment of low-altitude satellites in conjunction with spot-beam antennas leads to realizable link budgets that are compatible with the use of compact low-gain handset antennas with low uplink RF transmit power, as shown in Table 11.12, and low path delays. These advantages are accompanied by several propagation and spectral problems. For instance, Doppler shifts ranging from 36 kHz at 1.6 GHz and 55 kHz at 2.5 GHz have to be corrected with due regard to the relative direction of flight. Shadowing by buildings is more severe for the LEO and MEO systems than for the GEO ones and generates polarization-sensitive reflection and diffraction that manifests itself in multipath effects. 11.7.1.1 Downlink Array Design The photograph of the Iridium satellite in Figure 11.65 shows the outline shape and layout of the spot-beam phased array antennas. Full details of the Iridium array, referred to as the main mission antenna (MMA), have been published [88] and confirm the state-ofthe-art design that has been realized. The MMA consists of three fully active phased array panels, each one producing 16 fixed simultaneous beams, to give a total of 48 beams per satellite. While physical

524

Table 11.12 Satellite PCS Systems Iridium

Globalstar

ICO

Thuraya

Orbit Orbit altitude (km) Number of satellites + spare Orbit planes and Inclination Multiple access Target market

LEO 780 66 + 6

LEO 1,414 48 + 8

MEO 10,355 10 + 2

GEO 35,786 1+1

6 circular 86.5° TDMA-FDMA Global roamers

2 circular 45° TDMA-FDMA Global roamers

Spot beams per satellite Uplink freq. (GHz) Downlink freq. (GHz) User Terminal RF power (W) G/T (dB/K)

48

6 circular 52° CDMA-FDMA Cellular fill-in and fixed 16

163

0° TDMA-FDMA Cellular fill-in and fixed 256

1.616–1.626 1.616–1.626

1.610–1.621 2.483–2.495

1.980–2.010 2.170–2.200

1.626–1.660 1.525–1.559

0.45 −23.0

0.5 −22.0

0.625 −23.8

Source: [82, 85].

Figure 11.65 Iridium satellite showing the phased array antenna panels. (Courtesy of Iridium.)

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space requirements are not as critical as in the case of the user terminal antenna, the MMA design is constrained to satisfy a stringent specification demanding the simultaneous radiation of multiple carriers into multiple beams with high efficiency and linearity, within a lightweight, space-qualified, temperature-controlled structure. A seamless hand-off process enables users to be transferred from beam to beam and from satellite to satellite to achieve the best link margin at all times. Each array panel consists of more than 100 lightweight patch radiators each of which is connected to a transmit/receive (T/R) module powered by a regulator system embodying redundancy. The beamforming architecture is provided by two orthogonal stacks of Buttler matrices and a power divider. Intermodulation products are maintained at least 20 dB below the desired far-field carrier level by employing extensive prediction calculations in the design phase. 11.7.1.2 Generic Handset Antenna Constraints Personal handset antennas appear to be relatively simple devices, yet their design is constrained by many factors. For personal satellite terminal antennas, the design is much more critical and even a 1-dB loss or pattern distortion can render the system nonviable. There is, however, a very strong incentive to optimize handset antenna design because any increase in the link budget margin is achieved at much lower cost than that incurred in further modifying the satellite terminal antenna. The design of personal handset antennas for satellite operation is discussed in [84] and the most important issues are: • • • • •

Electrically small antenna behavior [89] is important for small-diameter radiators. An antenna function independent of the phone body is preferred. Requirements of input impedance matching over the prescribed bandwidth exist. The radiation pattern polarization and beamshape characteristics need to be maintained over the required bandwidth and in proximity to user. Noise temperature must be minimized to achieve the G/T specification.

However, main research has identified the QHA and its variants as the best radiating structure for LEO and MEO systems, and also in some cases GEO systems, where the margins are inherently small [90–93]. A particular advantage of the QHA is the increase in gain due to its hemispherical radiation pattern, which in turn reduces noise pickup. The fundamental relationship between antenna gain and size is well known and for electrically small ground planes the latter is usually part of the radiation structure and contributes to the overall radiator size as previously explained [84]. Given an efficient balun, the QHA is, apart from back-radiation impinging on the case, largely isolated from the handset case ground plane, thus decoupling hand effects to some extent. As such, the QHA needs to be supported some distance from the handset case to achieve isolation from the latter, in which case power loss in the hands is less of a problem than for cellular handset antennas. It thus remains to be seen if it becomes current practice to operate

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handset QHAs at some distance from both the handset case and head to achieve a clearer view of the satellite and less body perturbation, as opposed to connecting the QHA close to the handset case and sacrificing antenna pattern performance. Other recent research achievements include the use of ceramics to reduce the inherent bulky QHA size [91, 92], and smart QHAs [93], which offer an intelligent diversity system that has the potential to reduce multipath effects without the need for additional antennas. Ultimately it will be the user who will decide the size of the antenna that he or she is prepared to carry around in exchange for a global roamer service [84]. An appreciation of the system concepts for LEO/MEO/GEO satellites and the influence on the antenna design together with some recent practical examples of satcom handset antennas, are presented in the next section. 11.7.2 Handset Antennas for Satellite Systems 11.7.2.1 Basic System Concept Geostationary Earth Orbiting (GEO) systems have the satellite in a ‘‘fixed’’ position relative to the Earth, thus enabling a fixed terminal antenna to be steadily pointing toward the satellite, as is done for TV satellites (DBS). The big path loss caused by the distance (35,786 km) has to be compensated for by a big antenna at the satellite (implying high antenna gain) and perhaps high power (generated by big solar panels). In the newer satellite systems, multilobe antennas are used to combine coverage and capacity and as much as 25% to 30% of the circumference of the Earth can be covered. Widelobe terminal antennas are used in recent systems but for somewhat earlier systems bigger terminal antennas are used (such as the cover of a lap top) to meet the power budget requirement. GEO systems are typically supported by telecom companies in the countries below the satellites, and the system concept is fairly simple except for the fact that the satellite is very big: 12m antennas are in use and 25m antennas are being studied for future use. One disadvantage when using GEO systems is that the user has to accept a considerable latency due to the distance (0.25-second time delay one way from one phone on the ground to the other). Low earth orbit satellite systems are the other extreme, with satellites at 700 to 1,400 km, giving some 30 dB less path loss due to distance (as compared to GEOs), enabling much smaller antennas to be used in the satellite and also enable a design for better power budget margin. To keep contact continuous, including a certain redundancy, a considerable number of satellites must be used, such as 66 in the Iridium system, and an advanced handover scheme is necessary. A LEO system includes transmission to many ground stations and possibly also intersatellite connections. Widelobe and rather small antennas are used at the terminal. The number of satellites must allow two to three satellites to be visible at any time and outdoor location. This kind of system offers a negligible latency (delay)

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and enables a bigger capacity than a GEO system using the same bandwidth. A typical LEO satellite (Iridium) is fairly small due both to the small antenna and to the small solar panels, sufficient to fill the moderate power needs. 11.7.2.2 Frequencies Most systems are used or intended for telephone and low rate data and operate in a narrow frequency band within the range 1.4 to 2.5 GHz. This choice of frequency is (besides being an international frequency allocation) favorable for this kind of systems for three reasons: 1. Noise from the Earth (manmade noise) is high at low frequencies (< 100 MHz) but steadily decreasing at higher frequency. At 1 GHz it is lower than thermal noise. 2. Total noise from the sky is low in the 1 to 3-GHz range but increases at higher frequencies. 3. Effect of rain is negligible but will increase at higher frequencies. The choice of frequencies can also be illustrated by the propagation attenuation in free space (Friis law), which can be written in three ways to illustrate the influence of the different parameters: Pr /P t = G t G r (␭ /4␲ r )2 = A t G r /(4␲ r 2 ) = A t A r /(␭ r )2

(11.2)

Alternatively, antenna gain G and antenna effective area A are used to characterize the antenna connected by the basic relation G = 4␲ A /␭ 2. For a big-dish antenna, the area A corresponds to the geometrical surface of the dish. The first formulation implies that at low frequencies, where only low-gain antennas are possible, the power budget will improve if a low frequency can be used (utilized in early communication radio). The third formulation implies that at limited antenna area (radio links of all kinds) the power budget will be better if a high frequency is used (but also at the expense of the need for accurate directional aiming of the antennas and increased weather sensitivity). The second formulation is applicable to the actual kinds of communication systems where the receiver antenna gain G r is low but the transmitter antenna size A t is critical (cost-limited when it comes to 10m to 12m antennas in the sky). In this case, the choice of frequency is not critical for the power budget alone. Another obvious consequence from the second formulation is that the area A t of the satellite antenna must be bigger at bigger distances r, and that can be expressed as a required antenna diameter proportional to the distance to the Earth. A GEO satellite may have an antenna diameter of 10m to 12m, whereas a LEO satellite may have a 25 times smaller antenna diameter (≈ 35,786/1,400) to fulfill the same power budget. This is unnecessarily small, so a LEO system may use

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smaller power and still have a better power budget margin by using somewhat bigger antennas than the absolute minimum size mentioned above. There are also LEO systems for low rate data, only they operate at a much lower frequency, such as 150 MHz, and they also use low-gain terminal antennas (typically a quarter-wavelength whip antenna). The low frequency enables very small low-cost satellites with little power to be used and still fulfill the power budget. The low frequency greatly improves the power budget assuming low-gain antennas but bandwidth is also very limited as compared to the telephone systems mentioned above. The system cost is much lower than that of a full telephone system and the typical use is to send short messages (less than 200 bytes) for fleet management, emergency service, telemetry in remote areas, and so on. One such operating system is Orbcomm. In most cases two frequency bands (Rx and Tx) are used, and on the systems to date they are narrow compared to terrestrial cellular bands. In some systems, they are widely separated, which sets a number of special requirements on the antennas to keep their size small and still combine double antenna functions in one unit. 11.7.2.3 Antenna Types Whip antennas are used in terminals for satellite systems using VHF frequencies and those typically used on lorries. The obvious disadvantages with the linearly polarized whip is, of course, the 3-dB loss due to polarization and less suppression of wrongly polarized waves created by ground reflection. For the long wavelength, however, the simple whip implies big practical advantages and a part of the 3-dB polarization loss is regained by the highly simplified antenna structure. Quadrifilar antennas (see Section 11.5.2) have emerged as the preferred type for handsets based on a number of good virtues: • • •

Rather constant and well-polarized pattern over angles like ⌰ = 0°–70° or ⌰ = 0°–80°; Low gain for ⌰ > 90° giving low antenna noise temperature; Reasonable geometry if compared to other antennas giving a similar pattern.

Patch antennas are widely used for GPS but not for pocket satellite phones. The coverage for this latter use is rather bad because the radiation at low angles has a poor axial ratio simply because the horizontal polarization implies that the electrical fields are ‘‘shorted’’ by the ground plane, which is a functional part of the patch antenna (even if small). Furthermore, a hypothetical patch with sufficient telephone bandwidth generally has to be considerably bigger than a GPS patch, making it quite difficult to integrate in a phone. Roof-mounted patch antennas for cars are available for narrowband systems. Note that the GPS system is very different from a phone system and that the use of patch antennas is restricted to such GPS receivers where a handy shape is worth more than maximum performance. Many GPS receivers use a QHA antenna.

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Patch antennas using a higher mode are known for possible use on aircraft as true low-profile antennas having hemispherical coverage. They do not fit in the size of a handheld terminal and are complicated. 11.7.2.4 General Properties of the QHA The basic QHA [94] consists of four helical wires or ribbons on a cylinder having the radius a. Note that any number of wires greater than two can be used with the same function. This may be a motive for the term NHA rather than QHA but only N = 4 has reached widespread use. The helical ribbons are fed from a phasing network producing a phase progression of 90° between each wire (or 360/N in the NHA case). For N = 4, the phasing network is readily designed by standard 90° and 180° hybrid components of the lumped or distributed type. The radiation from the QHA geometry can be seen as a combination of vertical and horizontal polarization having a 90° phase difference. The function of the radiation can be understood, in a simplified way, as the radiation from a combination of four vertical monopoles together with a circular loop in the middle of the monopoles. The four monopoles radiate vertical polarization as an ordinary monopole but with a 360° phase rotation. Like any vertical monopole, the radiation toward zenith is zero. A vertical loop, on the other hand, gives a circular polarization toward zenith and horizontal polarization 90° away from zenith. Due to the 90° phasing and the small radius, the vertically polarized radiation from the four helices will be much less than the corresponding radiation from a single monopole carrying the same current. The combination of pitch angle and radius determines the relative amplitude of these two horizontal and vertical polarizations and by a suitable choice, good circularity can be obtained in all directions (at least above the horizon). The condition for circular polarization is approximately ka tan (␣ ) ∼ 1 where k = 2␲ /␭ , a = radius, and ␣ = the pitch angle (90° corresponds to vertical wires). For a typical phone antenna k is around 32 to 52 m−1 (1.5 to 2.5 GHz) and a = 6 to 8 mm, which gives a pitch angle around 75°. Note that each visualized horizontal plane slice of the QHA fulfills the same condition, and the length of the QHA is thus not critical for the polarization part of the radiation properties. On the other hand, the radiation pattern as well as the feeding impedance are more strongly dependent on the length. Some GPS receivers use a quarter-wave QHA while the typical phone antenna is 0.5 to 0.8 wavelength long. The radiation resistance (as referred to the maximum current somewhere along the helices) will be small due to the small radius of the virtual loop and the counteracting currents of the virtual vertical dipoles. This makes the narrow QHA (i.e., ka ≈ 0.2–0.35) a narrowband device. The typical QHA is a resonant antenna so the typical length can be thought of as ␭ /2 with an open end or ␭ /4 with a shorted end. In contrast to a straight single-wire dipole, the pattern for QHA having the length ␭ will not have any zero at ⌰ = 90° due to the combined

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effect of the twisting and the phase progression. The resonant structure simplifies the matching to a desired impedance (50⍀, and so forth), but is not important for the radiation properties. Radiation properties will be similar for another length, but then a condition for a reasonable input impedance may be a tuning coil or capacitor at the end, very analogous to single linear whip antenna. On a typical satellite phone system, the optimization of the gain is very important for making the minimum gain within the desired coverage as big as possible. For fine adjustments of the antenna pattern, the length is important and it turns out that the optimum hemispherical pattern generally occurs when the length is between 0.5 and 0.8 wavelengths. If the QHA is very short (␭ /4 or shorter), the radiation downward will increase, which will deteriorate the noise temperature of the antenna as well as the antenna gain. Quarter-wavelength QHAs are frequently used for GPS receivers. A QHA that is too long will, on the other hand, have a minimum or even a ‘‘hole’’ at zenith direction. Because QHAs are resonant antennas, the feeding can be made at any point along the helix, but a feeding at the bottom is simplest and most common. The bandwidth is mainly limited by the diameter, as the radiation resistance (and thus the bandwidth) will decrease proportional to the second power of the circumference expressed in wavelengths. A diameter of 12 to 16 mm thus seems to be close to the lower limit depending on the type of system. Polarization is determined by the direction of the helices and the phase order of feeding. Screw direction and phase order must coincide or the antenna will radiate in the wrong direction (i.e., downward). The feeding network can be done in many ways and for a wide bandwidth (of the feeding network alone) a combination of two 180° hybrids (sometimes called baluns) and one 90° hybrid or one 180° hybrid and 90° hybrids are used. Because of the two bands, the need for bandwidth of the feeding network may be considerable bigger than for each of the antenna bands separately. Distributed networks are used for the same purpose and tend to give lower loss at the expense of a bigger physical space. For very narrowband QHAs (mainly GPS antennas) a self-phasing QHA has been used. The four helices then can be considered as two pairs of twisted loops each having slightly different resonant frequencies giving a phasing of ±45° in each pair within a narrow frequency band. In the self-phasing case, it is important to distinguish between impedance bandwidth (which may be wide) and axial ratio bandwidth, which is very small (typically some 0.1%). Most GPS patches are self-phased by a very slight rectangular shape and a single but slightly off-centered feed. Figure 11.66 shows a typical phasing network using one 90° hybrid and two 180° hybrids (balun). In this case, the reflection in the QHA as well as possibly wrongly polarized signals received (including ground noise) will be absorbed by the termination connected to the 90° hybrid. The possibility of printing the lines on one side of the film substrate only can favor other network solutions. The widely separated uplink and downlink bands are hard to obtain in the original QHA concept using four wires on a small-diameter core, but by manufacturing the QHA as a printed pattern on a flexible film rolled on a cylinder, virtually any pattern can be printed with good accuracy. Due to the small diameter

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Figure 11.66 Phasing networks showing connections to QHA lines.

of this kind of QHA, the total bandwidth is small anyway, but it can be distributed over the two bands by modifying the pattern. Adding capacitance between the printed conductors is one method that has proven to be useful to achieve a sufficient two-band matching, and a meander-shaped pattern is another. Due to the requirements for gain and polarization purity, the design is much more determined by the radiation pattern requirements than by the impedance bandwidth, which over a limited band is easier to adjust by additional lumped or distributed elements. Two parallel helical and slightly different printed patterns (i.e., 2 × 4 conductors) have also been used to improve the Rx/Tx band coverage, and it is especially good in those cases where the Rx and Tx bands are close to each other but too separated to be treated as one band. The printed helical pattern is the radiating structure, but there will normally be some nonhelical and more or less undesired radiating parts as well, of which the feeding structure is one. A number of intentionally modified helical printed patterns have been proposed including a short vertical wire connected to the helices. A very important property of the QHA is the good isolation from the ground plane (such as on the roof of a car), and another useful property is the symmetry that can isolate the QHA from a cellular whip. 11.7.2.5 Examples of QHAs Antenna for the Iridium System Iridium can use a rather narrowband antenna because it uses the same frequency for uplink and downlink (1,616 to 1,626.5 MHz). The power budget has a good margin (said to be

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16 dB) to simplify use where the coverage is not perfect and to enable fairly low power at good connections (0.1W to 3.5W). Iridium was the first satellite telephone system with real handsets (operating late 1998). First and second generation handsets for Iridium are shown in Figure 11.67. The Iridium system was closed in 2000 for economical reasons. However, this system has restarted services by another company and continues to operate now. In the second generation of Iridium handsets (Figure 11.67), a QHA element has been used, because it has shown higher antenna performance than the first-stage antenna. Figure 11.68 shows the mechanical outline of such an Iridium antenna using an antenna element that is 100 mm long (active length 77 mm) and having a 17-mm diameter

Figure 11.67 Iridium handsets: (a) first generation; and (b) second generation. (Courtesy of Kyocera Corp., Japan.)

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Figure 11.68 Mechanical layout of Iridium handset antenna in Figure 11.67(b). (Courtesy of Allgon.)

(including radome). The antenna has a swivel joint so it can be used in a vertical position regardless of whether the phone is used with the right or left ear. A basic QHA with four plain helices is used to cover the rather narrow frequency band. The intended copolar and the cross-polarized antenna patterns are shown in Figure

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11.69 and illustrate the desired near-rectangular pattern. The input VSWR is shown in Figure 11.70, illustrating a fairly wideband behavior even if the optimized pattern and VSWR performance is obtainable in a narrowband only.

Antenna for Globalstar System The Globalstar system was put in service in 1999. It uses two widely separated frequency bands for uplink and downlink (1.6 and 2.5 GHz), which create a design challenge. A meander pattern along the helices together with double capacitive loadings is used to achieve this. Figure 11.71 shows a mechanical layout. The radome diameter is 15 mm and the total antenna length is 110 mm, of which 77 mm are active. Below the active part of the antenna, the phasing network is mounted on a small PCB together with the LNA to avoid the losses of any unnecessary piece of cable. The antenna is foldable so it can be used in a vertical position. The antenna pattern is shown in Figure 11.72 for Rx and Tx. The difference in shape depends on the fact that the physical length is used for both bands and the higher Rx band has a slight minimum at zenith.

Figure 11.69 Elevation radiation patterns (LHCP and RHCP) of Iridium antenna shown in Figure 11.68. (Courtesy of Allgon.)

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Figure 11.70 VSWR of Iridium antenna shown in Figure 11.68. (Courtesy of Allgon.)

Antenna for Thuraya System A number of GEO systems for handheld terminals such as the Thuraya system are in various stages of planning around the world. The active part is 100 mm long with a diameter of 14 mm. It uses the same 4 × 2 helix structure and is shown in Figure 11.73. Figure 11.74 shows a full Thuraya handset where it can be noted that the extended antenna is drawn straight out from the phone body without a swivel. The Thuraya antenna also includes a GSM antenna (see Figure 11.73), which utilizes the difference in symmetry properties between a QHA and a common monopole or axial helix. By the coaxial arrangement, the coupling and the associated losses will be very small. In this case the phone also includes a GPS patch. Because the polarization in this case is different for GPS (RHCP) and Thuraya (LHCP), it is not possible to use the same QHA (which otherwise would have been possible because the frequencies are fairly close).

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Figure 11.71 Mechanical layout of Globalstar antenna. (Courtesy of Allgon.)

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Figure 11.72 Elevation radiation patterns (Tx and Rx with lower gain at zenith) of Globalstar antenna shown in Figure 11.71. (Courtesy of Allgon.)

Figure 11.73 Mechanical layout of Thuraya handset antenna. (Courtesy of Allgon.)

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Figure 11.74 Thuraya handset phone. (Courtesy of Thuraya.)

REFERENCES [1] 2nd Mobile Satellite Conf., Ottawa, Canada, June 1990. [2] A series of MSAT-X Quarterly, Jet Propulsion Laboratory, No. 1 (1984) to No. 25 (1990). [3] Hamamoto, N., S. Ohmori, and K. Rondo, ‘‘Results on CRL’s Mobile Satellite Communication Experiments Using ETS-V Satellite,’’ Space Communications, 1990, pp. 483–493. [4] ‘‘PROSAT Phase I Report,’’ European Space Agency, ESA STR-216, May 1986. [5] Pisaruck, M., et al., ‘‘Connexion by BoeingSM–Broadband Satellite Services for Aircraft,’’ IEICE Technical Report, SAT2002-70, October 2002, pp. 105–109. [6] Gedney, R. T., and R. J. Scherfler, ‘‘Advanced Communications Technology Satellite (ACTS),’’ Int. Conf. Communications ’89 (ICC ’89 ), 1989, pp. 1566–1577. [7] Takeuchi, M., et al., ‘‘Experimental Advanced Mobile Satellite Communications System in MM-wave and Ka-Band Using Japan’s COMETS,’’ IEEE GLOBECOM ’92, Orlando, FL, December 1992, pp. 443–446. [8] http://www.jaxa.jp/projects/sat/ets8/index_e.html. [9] ‘‘Environmental Conditions and Test Procedures for Airborne Electronics, Electrical Equipment and Instruments,’’ Radio Technical Commission for Aeronautics, RTCA Do-160E, 2005. [10] Blake, L. V., Antennas, Dedham, MA: Artech House, 1984, Chapters 8 and 9.

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[11] Bell, D., et al., ‘‘L’MSAT-X Antennas: Noise Temperature and Mobile Receiver Grr,’’ MSAT-X Quarterly, No. 16, 1988, pp. 12–17. [12] CCIR, ‘‘Propagation Data for Maritime Mobile-Satellite Systems for Frequencies Above 100MHz,’’ Rep. 884-2, ITU, 1990. [13] Beckmann, R., and A. Spizzichino, The Scattering of Electromagnetic Waves from Rough Surfaces, New York: Pergamon Press, 1963. [14] Karasawa, Y., and T. Shiokawa, ‘‘Characteristics of L-Band Multipath Fading Due to Sea Surface Reflection,’’ IEEE Trans. Ant. Propag., Vol. AP-32, No. 6, 1984, pp. 618–623. [15] Sandrin, W., and D. J. Fang, ‘‘Multipath Fading Characterization of L-Band Maritime Mobile Satellite Links,’’ COMSAT Tech. Rev., Vol. 16, No. 2, 1986, pp. 319–338. [16] Karasawa, Y., and T. Shiokawa, ‘‘A Simple Prediction Method for L-Band Multipath Fading in Rough Sea Conditions,’’ IEEE Trans. Communications, Vol. Com-36, No. 10, 1988, pp. 1098–1104. [17] MSAT-X Technical Brochure, ‘‘Low-Cost Omni-Directional Vehicle Antennas for Mobile Satellite Communications,’’ Jet Propulsion Laboratory. [18] Adams, A. T., et al, ‘‘The Quadri-Filar Helix Antenna,’’ IEEE Trans. Antennas and Propag., Vol. AP-22, March 1974, pp. 173–l78. [19] Taira, S., M. Tanaka, and S. Ohmori, ‘‘High Gain Airborne Antennas for Satellite Communications,’’ IEEE Trans. Aerospace and Electronic Systems, Vol. 27, No. 2, March 1991, pp. 354–360. [20] Nakano, H., et al., ‘‘Singly-Fed Patch Antenna Radiating a Circularly Polarized Conical Beam,’’ Electronic Letters, Vol. 26, No. 10, May 1990, pp. 638–640. [21] Yasunaga, M., et al., ‘‘Phased Array Antennas for Aeronautical Satellite Communications,’’ 5th ICAP, March 1987, p. 1.47. [22] Howell, J. Q., ‘‘Microstrip Antennas,’’ IEEE Trans. Antennas Propagat., Vol. 23, January 1975, pp. 90–93. [23] Shiokawa, T., et al., ‘‘Microstrip Array Antenna for Aeronautical Satellite Communications,’’ Technical Report IEICE, A-P86-60, 1986, pp. 23–27. [24] Shiokawa, T., et al., ‘‘Cross Slot Array Antenna for Aeronautical Satellite Communications,’’ Technical Report IEICE, A-P86-59, 1986, pp. 17–21. [25] Ehrenspeck, H. W., ‘‘The Short Backfire Antenna, A New Type of Directional Line Source,’’ Proc. IRE, Vol. 48, January 1960, pp. 109–110. [26] Dod, L. R., ‘‘Experimental Measurements of the Short Backfire Antenna,’’ Rep. S-525-66-480, Goddard Space Flight Center, Greenbelt, MD, October 1966. [27] Takeuchi, K., M. Yasunaga, and T. Shiokawa, ‘‘Radiation Characteristics of Short Backfire Antenna Applicable to Mobile Communications,’’ IEICE Trans., Vol. E74, No. 10, October 1991. [28] Shiokawa, T., and Y. Karasawa, ‘‘Array Antenna Composed of 4 Short Axial-Mode Helical Antennas,’’ IECE of Japan, Vol. J65-B, No. 10, 1982, p. 1267. [29] Kumar, A., Fixed and Mobile Terminal Antennas, Norwood, MA: Artech House, 1991, Chapter 5. [30] MSAT-X Technical Brochure, ‘‘Low-Cost Omni-Directional Vehicle Antennas for Mobile Satellite Communications,’’ Jet Propulsion Laboratory. [31] Derneryd, A. G., ‘‘Analysis of the Microstrip Disk Antenna Element,’’ IEEE Trans. Antennas Propagat., Vol. AP-27, No. 5, September 1979, pp. 660–664. [32] http://www.inmarsat.com/. [33] http://www.deltawave.comm.com/. [34] http://www.jrc.co.jp/. [35] Noro, T., and Y. Kazama, ‘‘Helical Array Antennas with High Efficiency, Wide Frequency Bandwidth and Simple Structure,’’ IEEE AP-S International Symposium 2004, Vol. 1, June 2004, pp. 1050–1053. [36] Noro, T., and Y. Kazama, ‘‘A Novel Wideband Circular Polarization Microstrip Antenna—Combination of Different Shaped Antenna Element,’’ IEEE AP-S International Symposium 2005, Vol. 3A, July 2005, pp. 467–470. [37] Shiokawa, T., and Y. Karasawa, ‘‘Compact Antenna Systems for INMARSAT Standard-B Ship Earth Stations,’’ IEE 3rd Int. Conf. on Satellite Systems for Mobile Communication and Navigation, London, June 1983.

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[38] Ohmori, S., et al., ‘‘An Improvement in Electrical Characteristics of a Short Backfire Antenna,’’ IEEE Trans. Antennas Propagat., Vol. AP-31, No. 4, July 1983, pp. 644–646. [39] Stricland, P. C., ‘‘Low Cost, Electrically Steered Phased Array for General Aviation,’’ Proc. 2nd lnt. Mobile Satellite Conf., Ottawa, Ontario, 1990, pp. 169–171. [40] Ohmori, S., S. Taira, and A. Austin, ‘‘Tracking Error of Phased Array Antenna,’’ IEEE Trans. Antennas Propagat., Vol. AP-39, No. 1, January 1991, pp. 80–82. [41] James, J. R., P. S. Hall, and C. Wood, Microstrip Antennas-Theory and Design, IEEE Electromagnetic Wave Series 12, London, U.K.: Peter Peregrinus, 1981. [42] Long, S. A., ‘‘A Dual-Frequency Stacked Circular-Disk Antenna,’’ IEEE Trans. Antennas Propagat., Vol. AP-27, March 1979, pp. 270–273. [43] Teshirogi, T., et al. ‘‘Wideband Circularly Polarized Array With Sequential Rotations and Phase Shift of Elements,’’ Intl. Symp. Antennas Propagat. (ISAP1985 ), Tokyo, August 1985, pp. 117–120. [44] Ohmori, S., et al., ‘‘A Phased Array Tracking Antenna for Vehicles,’’ Proc. 2nd Int. Mobile Satellite Conf., Ottawa, Ontario, June 1990, pp. 519–522. [45] Maruyama, S., K. Kadowaki, and Y. Hase, ‘‘Experiments on Message Communications with Hand-Held Terminal,’’ IEICE Trans. B-II, VoI. J72-B-II, No. 7, July 1989, pp. 269–275. [46] Konishi, Y., et al., ‘‘S-Band Mobile Satellite Telephone for Aeronautical Use,’’ Technical Report of IEICE, SAT2002-8, May 2002, pp. 43–48 (in Japanese). [47] Lunsford, J., et al., ‘‘The AMSC/TMI Mobile Satellite Services (MSS) System Ground Segment Architecture,’’ Proc. 13th AIAA Intl. Commun. Satellite Systems Conf. and Exhibit. (ICSSC1992 ), AIAA-92-1855-CP, February 1992, pp. 405–426. [48] Konishi, Y., et al., ‘‘Mobile/Transportable Terminal Antennas for L-band Mobile Satellite Communication Systems,’’ Proc. 17th AIAA Intl. Commun. Satellite Systems Conf. and Exhibit. (ICSSC1996 ), 1996. [49] Konishi, Y., et al., ‘‘L-band Vehicle Antennas for Mobile Satellite Communication Systems,’’ Intl. Workshop on Multi-dimensional Mobile Communications (MDMC’94 ), No. 8-5, November 1994, pp. 163–166. [50] Ohtsuka, M., et al., ‘‘A Planar Array Antenna for Transportable Terminal in Mobile Satellite Communication Systems,’’ 1997 IEICE General Conf., B-1-68, March 1997, p. 68 (in Japanese). [51] Laase, E., ‘‘Aeronautical Internet Services Via Satellite—Present and Planned Systems,’’ Proc. 21st AIAA Intl. Commun. Satellite Systems Conf. and Exhibit. (ICSSC2003 ) Colloquium, April 2003. [52] Mizuno, T., ‘‘Aeronautical Internet Services Via Satellite—Overview of Key Technologies,’’ Proc. 21st AIAA Intl. Commun. Satellite Systems Conf. and Exhibit. (ICSSC2003 ) Colloquium, April 2003. [53] Monk, A., et al., ‘‘Ultra-low Profile Airborne Reflector Antenna Subsystem for Broadband Satellite Communications,’’ Proc. 21st AIAA Intl. Commun. Satellite Systems Conf. and Exhibit. (ICSSC2003 ), April 2003. [54] Nuimura, S., et al., ‘‘Airborne Ku-Band Antenna Subsystem for Satellite Communications,’’ IEICE Technical Report, SAT2004-130, October 2004, pp. 63–68. [55] Tranquilla, J. M., and S. R. Best, ‘‘A Study of the Quadrifilar Helix Antenna for Global Positioning System (GPS) Applications,’’ IEEE Trans. Antennas Propagat., Vol. AP-38, October 1990, pp. 1545–1550. [56] Carter, D., ‘‘Phase Center of Microwave Antennas,’’ IRE Trans. Antennas Propagat., Vol. AP-4, 1956, pp. 597-602. [57] Gerst, C., and R. A. Worden, ‘‘Helix Antenna Take Turn for Better,’’ Electronics, August 1966, pp. 100–110. [58] Kilgus, C. C., ‘‘Shaped Conical Radiation Pattern Performance of the Backfire Quadrifilar Helix,’’ IEEE Trans. Antennas Propagat., Vol. AP-23, May 1975, pp. 392–397. [59] James, J. R., and P. S. Hall, (eds.), Handbook of Microstrip Antennas, IEE Electromagnetic Wave Series 28, London, U.K.: Peter Peregrinus, 1989. [60] Kishk, A. A., and L. Shafai, ‘‘The Effect of Various Parameters of Circular Microstrip Antennas on Their Radiation Efficiency and the Mode Excitation,’’ IEEE Trans. Antennas Propagat., Vol. AP-34, No. 8, August 1986, pp. 969–976.

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[61] Huang, J., ‘‘Finite Ground Plane Effect on the Microstrip Antenna Radiation Patterns,’’ IEEE Trans. Antennas Propagat., Vol. AP-31, No. 4, July 1983, pp. 649–653. [62] Yasuda, A., ‘‘Status and Perspective of GPS,’’ IEICE Trans. B, Vol. J82-B, 1999, pp. 1207–1215. [63] Tehrani, H., and K. Chang, ‘‘Multifrequency Operation of Microstrip-fed Slot-ring Antennas on Thin Lowdielectric Permittivity Substrates,’’ IEEE Trans Antennas Propag., Vol. AP-20, 2002, pp.1299–1308. [64] Dahele, J. S., ‘‘Dual-Frequency Stacked Annular Ring Micro-Strip Antenna,’’ IEEE Trans. Antennas Propagat., Vol. AP-35, 1987, pp. 1281–1285. [65] Jan, J. Y., and K. I. Wrong, ‘‘Single-Feed Dual-Frequency Circular Micro-Strip Antenna with an OpenRing Slot,’’ Microwave Optical Tech. Lett., Vol. 22, 1999, pp. 157–160. [66] Nioisaka A., ‘‘A Study of ITS Vehicle Mounted Multi-Band Antenna,’’ Technical Report of IEICE, AP2001-167, 2002 (in Japanese). [67] Boccia, L., G. Amendola, and G. Di Massa, ‘‘A Dual Frequency Micro-Strip Patch Antenna for High Precision GPS Applications,’’ IEEE Antenna Wireless Propagat. Lett., Vol. 3, 2004, pp. 57–60. [68] Yang, F., and Y. Rahmat-Samii, ‘‘A Low Profile Circularly Polarized Curl Antenna over an Electromagnetic Band-gap (EMG) Surface,’’ Microwave Optical Tech. Lett., Vol. 31, 2001, pp. 165–168. [69] Kazama, Y., S. Kumagi, and T. Shiokawa, ‘‘Vertical Polarization Single-Feed Dual-Frequency Microstrip Antenna with an Arc-Shaped Slot,’’ IEICE Electron Express, Vol. 2, 2005, pp. 60–63. [70] Kumagi, S., Y. Kazama, and T. Shiokawa, ‘‘A Study on Multi-Band Antennas for Vehicles,’’ Proc. 11th World Congress on ITS Nagoya, No. 3244, 2004, p. 216. [71] Kazama, Y., S. Kumagai, and T. Shiokawa, ‘‘A Novel Multi-Band Antennas for ITS Applications,’’ Proc. JINA2004, 2004, pp. 444–S445. [72] Ogata, D., et al., ‘‘Elliptical Slot-Ring Antenna for Multi-Frequency Operation,’’ Proc. Intl. Symp. Propagat. (ISAP2004), Vol. 2, 2004, pp. 689–692. [73] Ogata, D., et al., ‘‘A Study of an Elliptical Slot Ring Antenna,’’ 2004 IEICE National Convention, B-1-162, 2004 (in Japanese). [74] Miura, Y., et al., ‘‘Studies on Three Frequency GPS Antenna,’’ Technical Report of IEICE, AP2004-135, 2004 (in Japanese). [75] Garg, R., and P. Bhartia, Micro-Strip Antenna Design Handbook, Norwood, MA: Artech House, 2001, pp. 441–463. [76] Kumagai, S., Y. Kazama, and T.Shiokawa, ‘‘Single-Layer Single-Feed Multi-Band Antennas,’’ IEICE Trans. B, Vol. J89-B, No. 9, September 2006, pp. 1603–1612. [77] Chen, W. C., ‘‘A Broad-Band Annular-Ring Microstrip Antenna,’’ IEEE Trans. Antennas Propagat., Vol. AP-30, September 1982, pp. 918–922. [78] Ogawa, M., et al., ‘‘Radial Line Microstrip Array Antenna Using Film Substrate for Mobile BS Antenna System,’’ 1998 IEICE General Conference, B-1-153, March 1998 (in Japanese). [79] Yang, F., and Y. Rahamt-Samii, ‘‘A Single Layer Dual Band Circularly Polarized Microstrip Antenna for GPS Applications,’’ IEEE AP-S Intl. Symp., Vol. 4, June 2002, pp. 720–723. [80] Yang, F., and Y. Rahamat-Samii, ‘‘Switchable Dual-Band Circularly Polarized Patch Antenna with Single Feed,’’ Electronics Lett., Vol. 37, August 2001, pp. 1002–1003. [81] Boccia, L., G. Amendola, and G. D. Massa, ‘‘A Dual Frequency Microstrip Patch Antennas for GPS Applications,’’ IEEE Antennas Wireless Propagat. Lett., Vol. 3, 2004, pp. 157–160. [82] Evans, J. V., ‘‘Satellite Systems for Personal Communication,’’ Proc. IEEE, Vol. 86, No. 7, July 1998, pp. 1325–1341. [83] Williamson, M., ‘‘Satellite Constellations in the Ascendant,’’ IEE Review, Vol. 44, No. 5, September 1998, pp. 209–213. [84] James, J. R., ‘‘Realizing Personal Satcom Antennas,’’ IEE Electronics and Communications Eng. J., April 1998, pp. 73–82. [85] Lisi, M., ‘‘Satellite Communications Systems in the 90s and for the Next Millennium,’’ Microwave Engineering Europe, May 1999, pp. 15, 16, 18, 20, and 22. [86] Pattan, B., Satellite-Based Global Cellular Communications, New York: McGraw-Hill, 1997.

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[87] Siwiak, K., Radiowave, Propagation and Antennas for Personal Communications, 2nd ed., Norwood, MA: Artech House, 1998. [88] Schuss, J. J., et al., ‘‘The IRIDIUM Main Mission Antenna Concept,’’ IEEE Trans. Antennas Propagat., Vol. 47, No. 3, March 1999, pp. 416–424. [89] Fujimoto, K., et al., Small Antennas, Research Studies Press, New York: Wiley, 1987. [90] Agius, A. A., ‘‘Antennas for Handheld Satellite Personal Communications,’’ Ph.D. Dissertation, University of Surrey, 1999. [91] Leisten, O. P., Y. Vardaxoglou, and E. Agboraw, ‘‘Simulating the Dielectric-Loaded Quadrifilar Helix Antenna Using a Brute-Force TLM Approach,’’ Proc. 15th ACES Conf., Vol. 1, March 15–20, 1999, p. 479. [92] Nicolardis, G., O. Leisten, and Y. Vardaxaglou, ‘‘TLM Investigation of Dielectric-Loaded Bifilar Personal Telephone Antennas,’’ NCAP99, University of York, March 31–April 1, 1999, pp. 16–19. [93] United Kingdom Patent Application, No. 9803273.3, February 17, 1998. [94] Kumar, A., Fixed and Mobile Terminal Antennas, Norwood, MA: Artech House, 1991, Chapter 5.

Chapter 12 UWB Antennas Tadahiko Maeda

12.1 UWB SYSTEMS: INTRODUCTION Beginning with the importance of antennas in ultra-wideband (UWB) telecommunications for future wireless communications applications, this chapter provides a cohesive vision of the analysis and evaluation of antenna systems over the UWB frequency spectrum. This chapter also covers the basic principles and theoretical and practical technologies required to enable spectral-efficient antenna systems to coexist with other wireless systems, as well as an assessment of the potential structural characteristics of UWB wireless telecommunications systems. In 2002, the U.S. Federal Communications Commission (FCC) allocated the unlicensed use of UWB frequency spectrums [1]. Three types of UWB system applications that require UWB frequency spectrum include: (1) short-range communication and measurement of peer-to-peer contact both indoors and outdoors at 3.1 to 10.6 GHz; (2) radars that prevent vehicle collisions and detect an object’s position and movement with center frequency larger than 24.075 GHz in the 22- to 29-GHz band; and (3) inner-wall surface inspection in medical and underground exploration by radar imaging systems. In particular, UWB systems are gaining attention for short-range communication and measurement applications [in (1) above] because of the possibility of generating extremely short pulses with the simple switching circuit prevalent inside microprocessors; therefore, ultra-broadband antennas have come to play an important role as a key component in the realization of the system. In short, the FCC’s deregulation of the UWB frequency spectrum triggered the accelerated development of antennas capable of covering the entire UWB frequency band. 543

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Such antennas may be divided into several groups based on categories or criteria: (1) antenna type, (2) shape of the radiating element, (3) feed line structure, and (4) an antenna’s working principle. 12.2 REQUIREMENTS FOR UWB ANTENNAS As the origin of the UWB concept, an Impulse Radio (IR) system [2] that falls under the category of baseband transmission was principally studied during the initial phase. This could be done to constitute a low-cost, ultra-high-speed wireless network based on RF-CMOS technology. The greatest obstacle in the practical use of IR is that it occupied bandwidth far exceeding the bandwidth needed for conventional technologies. What is the major difference in requirements between broadband antennas and UWB antennas? Before the advent of the UWB system, a modulated signal’s bandwidth in wireless communication was relatively narrow compared to its carrier frequency. For example, a transmitted television signal has a relative bandwidth to the center frequency of about 5%, whereas the total assigned frequency band extends more than one octave. On the other hand, in a UWB system, maintaining the ultra-wideband waveform of the signal is very important. In short, phase characteristics (in other words, group delay characteristics) of a UWB antenna are the most important parameters for UWB communication links. This point was not focused on when evaluating ordinary frequency-independent antennas. This requirement becomes more serious for IR UWB systems than for multiband UWB systems in which the entire bandwidth is divided into multiple subbands. Another more practical requirement for UWB antennas relates to the cost of the antenna or the cost of the related technologies. The original motive for developing a UWB system was for cost-effective (or low-cost), short-range, high-speed wireless communication. The upper frequency of a UWB system is 10.6 GHz, a relatively high frequency for which engineers need different technologies compared to conventional mobile communication technologies that mainly use the UHF frequency band. This creates several difficulties with both simulation and experiments during the practical stages of development. 12.2.1 Basic Principle of UWB Antennas As is common knowledge, the two types of extremely broadband antennas are: (1) selfsimilar antennas, and (2) self-complementary antenna. The self-similar antenna is similar to itself against frequency variations, as represented by the biconical antenna. The principle of the self-complementary antenna [3, 4], invented by Professor Mushiake at Tohoku University in the late 1940s, is that its self-complementary structure shows constant impedance. If the self-complementary antenna has infinite size, the input impedance of the most fundamental self-complementary antenna becomes 188⍀ without any fluctuation, for an infinite range of frequencies.

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The geometries of self-similar antennas are specified by angles. The finite-size biconical structure shown in Figure 12.1 is the most common example of an extremely broadband antenna. A bow tie antenna is a planar modification of the biconical antenna. Since the actual antenna is limited in size, the lower limit of the operating frequency is determined by the largest size of the structure and the upper limit of the operating frequency is determined by the physical structure in the vicinity of the feeding point. 12.2.2 Modeling and Structure of Feeding Points Given its flexibility, the finite-difference time-domain (FDTD) method has been widely used in recent years for the numerical simulation of antenna systems as well as UWB antennas. When calculating an antenna’s input impedance, modeling an antenna’s feeding structure when the physical dimension of the feeding point is insufficiently small compared to the wavelength is important. One of the most common RF connectors in antenna experiments for input impedance measurement is the SMA receptacle (or connector), where the most common distance between the inner conductor and the outer conductor is about 1 to 2 mm. When comparing calculated results with measurement results for higher frequency bands, we often encounter the issues of fabricating the actual feeding point for the actual measurement and of modeling the feeding point during simulations. The SMA feed port mainly affects the third and fourth resonance by shifting their resonant frequencies [5]. Figure 12.2 shows a circular disc monopole antenna placed on a finite size ground plane.

Figure 12.1 Biconical antenna.

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Figure 12.2 A circular disc monopole antenna placed on a finite size ground plane and a measurement setup with magnifying factor of 1.6.

The calculated and measured results of a disc monopole antenna with a radius of 20 mm placed on a finite size ground plane where all measurements performed using the measurement setup with magnifying factor of 1.6 are shown in Figure 12.2. The size of the gap between the lower edge of the metallic disc and the upper surface of the dielectric material of the SMA (Sub Miniature Version A) connector varied from 1.0 to 0.5 mm and measured input impedances for both gaps are shown in Figure 12.3(a, b). As shown in Figure 12.3(c, d), the VSWR is improved when the size of the gap is 0.5 mm rather than 1.0 mm. Comparing Figure 12.3(a) with Figure 12.3(b), it is important to narrow the gap to lower the reactance of the input impedance closer to ‘‘zero’’ over the 5-GHz range. Several feeding models with calculated accuracy including the convergence characteristics for simulations are summarized in [6]. Compared with the hard-source feed model, transmission line feed results show excellent agreement with measured results. Additionally, simulation results for a horn antenna with a parallel line feeding section [7], as an example of ground penetrating radar, and the FDTD analysis of a dielectricloaded, horn-fed, bow tie antenna with precise modeling for the feeding structure, dielectric, and resistor, are reported [8]. Figure 12.4 demonstrates the difference in the simulated results of a circular disc dipole with a radius of 12.5 mm and a feeding gap of 1.5 mm for two feeding structures: (1) the ‘‘one-cell gap model,’’ and (2) the improved feeding

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Figure 12.3 (a) Input impedance: Gap = 1.0 mm. Note: actual measurement frequency range was from 1/1.6 GHz to 11/1.6 GHz with Gap = 1.6 mm. (b) Input impedance: Gap = 0.5 mm. Note: actual measurement frequency range was from 1/1.6 GHz to 11/1.6 GHz with Gap = 0.8 mm. (c) VSWR: Gap = 1.0 mm. Note: actual measurement frequency range was from 1/1.6 GHz to 11/1.6 GHz with Gap = 1.6 mm. (d) Input impedance: Gap = 0.5 mm. Note: actual measurement frequency range was from 1/1.6 GHz to 11/1.6 GHz with Gap = 0.8 mm.

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Figure 12.3 (continued).

gap model (which assumes an infinitesimally narrow feeding gap instead of a one-cell gap [9] incorporating the effects of the radius of the feeding point). The difference is a result of the difference in geometry in the drive-point region of the two feeding models. In other words, conducting numerical simulation including the feed line or RF connector, as well as the antenna itself, is important.

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Figure 12.4 Measurement results for disc monopole antenna with FDTD simulations where Model A corresponds to the improved feeding gap model and Model B to the one-cell gap model.

12.2.3 Current Distributions of Circular Disc Monopole Antenna The effects of the ground plane size of a circular disc monopole antenna were investigated [10]. The results demonstrate that the antenna has omnidirectional radiation patterns over the entire bandwidth and reducing the width of the ground plane without any resulting degradation in performance can minimize the antenna’s size. Current distributions of a circular disc planar radiating element are shown for various frequencies to demonstrate the basic radiation mechanism of a disc monopole antenna. Figure 12.5 shows the current distributions of a circular disc monopole antenna placed on a finite size ground plane [11]. As shown in the figures, the amplitude of the currents along the edge is higher than that of the current around the center area of the disc over a wide frequency range, from lower band to higher band. The observance of horizontal stripes in Figure 12.5(a) at 3.1 GHz shows that the center portion of the disc also works as a radiator, along with the edges. However, as frequency increases, the amplitude of the current around the center area becomes lower, from the upper portion to the lower portion of the disc as shown in Figure 12.5(b, c). Figure 12.6 shows the calculation results for the case where both the length and width of ground plane were 1.5 times and the radius of the disc was twice. Because the dimensions of the antenna were twice, the current distribution at the center furthermore decreased and the number of standing waves increased along the edge.

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Figure 12.5 Current distributions for circular disc monopoles. (a) 3.1 GHz: Radius = 12.5 mm; Ground plane = 100 × 100 mm; Gap = 1.0 mm. (b) 6.85 GHz: Radius = 12.5 mm; Ground plane = 100 × 100 mm; Gap = 1.0 mm. (c) 10.5 GHz: Radius = 12.5 mm; Ground plane = 100 × 100 mm; Gap = 1.0 mm. (From: [11].  2005 IEICE.)

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Figure 12.6 Current distributions for circular disc monopoles. 10.0 GHz: Radius = 25 mm; Ground plane = 150 × 150 mm; Gap = 0.7 mm. (From: [11].  2005 IEICE. Reprinted with permission.)

The current distributions shown in Figures 12.5 and 12.6 imply the importance of the feeding structure of UWB antennas especially for the higher frequency band.

12.3 CHARACTERISTICS OF POPULAR UWB ANTENNAS This section discusses both type and shape of UWB antenna radiators. Among the several parameters used to assess UWB antennas, the flat impedance characteristics of a UWB antenna to cover the entire UWB frequency band is one of the most important requirements. To determine the basic and first order contributions of each portion of the radiator to the impedance characteristics of a UWB antenna, we make the fair assumption that the lowest (closest to the feeding point) portion of the radiating element primarily affects input impedance characteristics in a higher frequency range, whereas the entire portion (such as the total length) of the radiating element primarily affects input impedance characteristics at a lower frequency range. In experiments, we often encounter difficulties in measuring the input reactance at high frequency ranges. Therefore, scale models of an antenna are commonly used since the physical dimensions of the originally designed antenna may be scaled down or up by changing the frequency used during actual measurements. This creates a trade-off between the accuracy of the fabrication and the total size of the experiment’s setup. Based on the principal structures of an antenna, two major types of UWB antennas exist: (1) three-dimensional antennas (or flat radiating elements placed on a ground plate), and (2) planar antennas. A planar antenna is usually fed through a microstrip line or coplanar waveguide and the planar structure enables the radiating element to be fabricated on the same dielectric material on which the transmitter and receiver are fabricated.

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12.3.1 Three-Dimensional UWB Antennas Two types of opened-out [12] transmission lines (unbalanced transmission lines and balanced transmission lines) are used to realize the smooth transition from the nonradiated electric fields along transmission lines to the radiated electric fields through antennas. The biconical antenna, an example of a balanced opened-out modification, has inherent broadband properties and possesses omnidirectional pattern and stable impedance characteristics over a wide range of frequencies. The properties (electrical characteristics) of a biconical antenna are a function of its apex angle. The angle resulting in an input impedance of 100 ohms is about 90°. In contrast to the biconical antenna, the conical antenna is an example of unbalance opened-out modification. To obtain a very low VSWR over the entire UWB bandwidth, a conical radiating element and its ground plate have been modified to form a smooth transitional shape known as a volcano-smoke shape, as shown in [13]. Also, a circular ball and its modified structures (family) are used for wideband radiating elements placed on a ground plate. One of the simplest circular ball modifications is a circular disc monopole that exhibits a relatively good match to 50-ohm coaxial cables. While a circular disc-type radiating element does not have an axially symmetric structure in the azimuth plane, a circular disc plate radiating element provides little effect on the radiation pattern. Also, a horizontally top-loaded small disc attached at the center of the horizontal edge of a half-circular disc radiating element was proposed to reduce the total size of the circular disc monopole [14] as shown in Figure 12.7. A knight’s helm shape, double-sided printed circuit board antenna (3 × 3 cm) in Figure 12.8 was also proposed for a low-cost UWB antenna [15] where W1 = 15 mm, L1 = 12.5 mm, L2 = 11.5 mm, L3 = 26 mm, and L4 = 7.5 mm. To achieve good impedance characteristics, the antenna uses the following three techniques: (1) dual slots on the rectangular patch radiating element, (2) tapered connection

Figure 12.7 Vertical half disc-loaded monopole antenna with a horizontally top-loaded small disc. (From: [14].  2005 IEEE.)

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Figure 12.8 The knight’s helm shape antenna. (From: [15].  2005 IEEE.)

between the rectangular patch and the feed line, and (3) a partial ground plane flushed with the feed line [15]. The characteristics of annular planar monopole antennas [shown in Figure 12.9(a)] with different circular holes and feed gaps have been studied experimentally by Chen [16] and the results demonstrated that the proposed antenna is capable of providing broad impedance characteristics even with over half the circular element removed. In contrast to axially symmetric radiating elements, planar radiating elements could suffer pattern degradation because of their geometry, especially at higher frequency. As shown in Figure 12.9(b), the concept of using holes in the radiating element was applied to the planar inverted cone antenna to enhance both pattern performance and the VSWR [17].

Figure 12.9 (a) Geometry of annular planar monopole antenna. (From: [16].  2002 IEEE.) (b) Twocircular-hole planar inverted cone antenna. (From: [17].  2004 IEEE.)

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When this antenna is modified into a dipole form, the geometry is similar to modified tapered-slot antennas and planar elliptical dipole antennas [17], especially with respect to the lowest edge of the radiating element. The characteristics of four planar dipoles (shown in Figure 12.10), including their current distributions were studied [18]. As can be seen with the circular disc monopole shown in the previous section, the current densities at the edges of four planar dipoles are much greater than the densities at the center area of the radiators. The current distribution for a center-fed square dipole with a shorting pin [18] (Ant. D in Figure 12.10) shows that, without a short pin at one side of the edge, current flows on both sides from the center—where a feed pin is connected—are in the opposite direction, whereas the current flow directions are the same with a short pin. The short pin also affects the current flow on the half side of the radiator in which a short pin is connected.

Figure 12.10 Antenna geometries. Ant. A: a center-fed square dipole; Ant. B: a beveled center-fed square dipole; Ant. C: an offset square dipole; Ant. D: a center-fed square dipole with a shorting pin. (From: [18].  2005 IEEE. Reprinted with permission.)

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Valderas [19] analyzed UWB folded-plate monopole antennas employing transmission line modeling between upper and lower rectangular radiating elements in which both plates are fed with a short, thin wire segment at the source connection point. Neglecting the edge effects at the lowest edge of the radiating element, the radiation pattern of this type of radiating element can be determined primarily by the vertical current flow on the radiator. This is especially the case with a symmetrically shaped antenna fed at the center of the radiating element since current flows in the opposite direction from the feeding point along the lowest edge [19]. This is a simplified discussion but it illustrates several basic aspects of the radiation mechanism of planar UWB antennas. 12.3.2 Planar UWB Antennas Several types of UWB antenna radiating elements have been reported in the literature, including square, triangle, circular, and hexagonal elements. Some of these UWB antennas have a monopole structure with their ground planes perpendicular to the radiators. This can be seen as a drawback since integrating the ground plane and radiator with the printed circuit boards is difficult. A thin wire dipole or monopole suffers from narrow impedance characteristics. Using a planar configuration instead of a thin wire configuration as the radiating element for wideband antennas is useful. Several basic types of planar radiating elements exist for UWB antennas. The general principle of these radiating elements is to use a wide-shaped radiator (usually where the width of the radiator tapers along the axis). As shown in Figure 12.11, the taper may occur at several points along the dimensions of an antenna: (1) taper in diameter or width of the radiating element, (2) taper in the distance between the upper and lower radiating element, (3) taper in the gap at the feeding point,

Figure 12.11 Types of taper in UWB antennas.

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and (4) taper in the width of a slot antenna or slotline. In short, tapers in the physical structure are critical for the wideband operation of UWB antennas. A Vivaldi antenna [20] may be regarded as a flared-out planar transmission line. The tapered radiating element produces a uniform phase front through the transmission line to a larger aperture. Several types of endfire tapered slot antennas, including exponentially tapered (Vivaldi), linearly tapered, and constant-width antennas, on dielectric substrate were studied and several consistent empirical design rules have been demonstrated [21]. Preliminary studies on design parameters of a Vivaldi notch-antenna array have been conducted [22] by the moment method with triangular basis function and impedance matrix interpolation [23]. The antenna element parameters can be subdivided into: (1) a stripline-to-slotline transition, (2) a tapered slot, and (3) a stripline stub and slotline cavity. By introducing an equivalent circuit for a Vivaldi notch-antenna array element, the number of structural parameters of the array has been reduced since the equivalent circuit enables the elimination of stub-related structural parameters. An analytical model for the current distribution on a Vivaldi antenna has been derived using an improved model of an exponentially tapered nonuniform transmission line [24] and a practical example of Vivaldi antenna with a Double-Y balun is shown in Figure 12.12. A conformal double exponentially tapered slot antenna was fabricated on a 200-␮ m thick liquid crystal polymer (LCP) substrate with an 18-␮ m thick copper layer and tested [25] for UWB applications. The total length of the antenna was 130.70 mm and the maximum aperture slot size was 42.92 mm. Because of the substrate’s mechanical flexibility, the antenna could be flexed in a conformal shape, such as that for an automobile hood or bumper [25]. A major effect of the conformation is an increase in cross-polarization radiation, while the main lobe radiation pattern was maintained stable even with the conformation.

Figure 12.12 Practical example of Vivaldi antenna with a Double-Y balun. (From: [24].  2006 IEEE. Reprinted with permission.)

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12.3.3 CPW Feed The coplanar waveguide (CPW) configuration facilitates the use of an active device placed near the antenna’s radiating element and is suitable for monolithic integrated circuits. The performance and characteristics of a CPW-fed circular disc monopole antenna are investigated in terms of several design parameters, including the effects of the feed gap, the width of the ground plane, and the dimension of the CPW-fed circular disc monopole [5]. The geometry of the CPW-fed circular disc monopole is shown in Figure 12.13. While the dimension of the CPW primarily affects the first resonant frequency, the width of the feed gap significantly changes the impedance characteristics above 8 GHz [5]. In short, the first resonant frequency is determined by the dimension of the circular disk and the width of the ground plane changes the higher resonant frequencies substantially. An antenna configuration comprising a 50-ohm CPW and a pair of 100-ohm slotlines con-

Figure 12.13 Geometry of the CPW-fed circular disk monopole. (From: [5].  2005 IEEE. Reprinted with permission.)

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nected to tapered ring slotlines shown in Figure 12.14 [26] demonstrates an impedance bandwidth with VSWR < 2 from 3.1 GHz to over 12 GHz. To suppress the cable leakage current with printed dipole UWB antennas, a CPWfed printed dipole structure with thin leakage-blocking slots (shown in Figure 12.15) placed along the feeding CPW was presented. Additionally, two types of leakage currents (shown in Figure 12.16) on the outer surface of the feeding coaxial cable’s outer conductor were examined in detail [27]. Total leakage currents occurred as follows: (1) leakage (first type) occurs at the interface between the coaxial cable and the CPW, and (2) leakage (second type) comes from the current along the CPW trace, but only after it has traveled over the surface of the bottom radiator as shown in Figure 12.16. The second type of leakage current can be suppressed by introducing two thin slots as shown in Figure 12.15 (note: these slots are not effective to prevent the first type of leakage current). Also, circular and elliptical

Figure 12.14 Geometry of coplanar waveguide-fed tapered ring slot antenna. (From: [26].  2006 IEEE. Reprinted with permission.)

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Figure 12.15 CPW-fed printed dipole UWB antenna with two leakage-locking slots. (From: [27].  2006 IEEE. Reprinted with permission.)

Figure 12.16 Two types of leakage currents on the outer surface of the outer conductor of the feeding coaxial cable. (From: [27].  2006 IEEE. Reprinted with permission.)

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CPW-fed (shown in Figure 12.17) and microstrip-fed (shown in Figure 12.18) antennas were studied [28]. Four fabricated prototypes are comprised of elliptical or circular stubs that excite similar-shaped slot apertures shown in Figure 12.17. In short, the first resonance is controlled by the diameter (or the ‘‘equivalent’’ diameter when referring to elliptical stubs) of the tuning stubs. Based on the overall radiation characteristics, it has been concluded that two prototypes having different feeding structures—(1) CPW-fed and (2) microstrip-fed—appear to radiate with the same way.

Figure 12.17 Geometry of CPW-fed slot antenna. (From: [28].  2006 IEEE. Reprinted with permission.)

Figure 12.18 Geometry of microstrip-fed elliptical slot antenna. (From: [28].  2006 IEEE. Reprinted with permission.)

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12.3.4 Multilayer Technologies Another feed line suitable for a multilayer configuration is a shielded strip-line. A circularshaped monopole (radius of 7.5 mm) antenna is fed through a shielded strip-line to allow for integration in multilayer system-on-package technologies. As shown in Figure 12.19, the antenna was fabricated with a multilayer structure consisting of a dual-layer substrate, with each substrate layer having thickness of 1.575 mm and with metal etched in the two sides [29] where Ws = 20 mm, r = 7.5 mm, Ls = 20 mm, Ld = 1.5 mm, and Lg = 37 mm. While a traditional ring antenna (shown in Figure 12.20) may be categorized into a narrowband antenna [30], a proximity-coupled configuration was applied to a UWB annual ring antenna fed by a microstrip line.

Figure 12.19 Geometry of the dual-layer UWB antenna. (From: [29].  2006 IEEE. Reprinted with permission.)

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Figure 12.20 Geometry of the UWB annual ring antenna: (a) annual ring layer; (b) microstrip feed-line layer; (c) bottom ground plane layer; and (d) cross-section view. (From: [30].  2006 IEEE. Reprinted with permission.)

The antenna, consisting of multiple layers, included: (1) an annual ring antenna layer, (2) a microstrip feed-line layer, and (3) a bottom ground plate layer, and has a return loss better than 10 dB from 2.8 to 12.3 GHz and a maximum gain of 5 dBi at 7 GHz [30]. 12.3.5 Band-Rejection for Coexistence with Other Wireless Systems UWB antennas are also necessary to suppress interference signals from existing wireless systems, like IEEE 802.11a standard, allowing electromagnetic compatibility of a UWB system with existing wireless systems. Therefore, several physical configurations are studied to obtain the notched characteristics in such wireless frequencies. Typical bandrejection antennas consist of two components: (1) a radiating element with broadband radiation characteristics, and (2) an attached element (or removed part) that provides the band-notched characteristics. Several combinations [31–33] of these two components are

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shown in Figure 12.21, Figure 12.22, and Figure 12.23 and can be categorized in resonant structure filtering [34]. Qiu concluded that both a C-shape attachment and a ring-shape attachment show broader band-rejection characteristics compared to the other two types of attachments [35] and proposed an antenna structure design with two parts: (1) an annual ring plane monopole, and (2) a C-shape attached element as shown in Figure 12.24. A staircase-shape planar monopole antenna having a half-bow tie radiating element and a modified ground plane structure [36] shown in Figure 12.25 was proposed for reducing interference with existing wireless network standards, such as IEEE 802.11a in the United States (5.15–5.35 GHz, 5.725–5.825 GHz) and HiPERLAN/2 in Europe (5.15–5.35 GHz, 5.47–5.725 GHz) [37]. A U-shaped slot on the topside of the antenna [36], with dimensions 25 × 26 × 1 mm, plays the role of the band-rejected filter. It has been concluded, based on the investigation, that the area near the radiating element feeding point, where current distributions are strong, is a sensitive part of the tuning point. Also, the reactance of the input impedance can be easily controlled by both the extended ground plane and two slits placed near the area [36]. A slit-based concept for band-rejection has also been applied to a three-dimensional radiating element [38] as shown in Figure 12.26. Several stub configurations, including a curbed stub inside a spherical monopole radiating element placed over a substrate and ground plane, were proposed and studied to control the creation of stop bands in the antenna’s matching bandwidth [38]. In addition to the notch filtering structures shown in this section, a stepped impedance UWB slot

Figure 12.21 Geometry of the circular monopole with two slits and CPW feeding line. (From: [31].  2004 IEEE. Reprinted with permission.)

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Figure 12.22 Narrowband resonant structures with triangular notch and elliptical notch. (From: [32].  2003 IEEE. Reprinted with permission.)

Figure 12.23 Tapered slot antenna with quarter wavelength short stubs for band-stop characteristic. (From: [33].  2004 IEEE. Reprinted with permission.)

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Figure 12.24 Structure of band-notched antenna with C-shape attachment. (From: [35].  2006 IEEE. Reprinted with permission.)

antenna with a four-pole low-pass response was proposed [39] as an example of steppedimpedance line filtering to provide more degrees of freedom in designing the filtering characteristics of the radiating element. 12.4 WIRE-STRUCTURED UWB ANTENNAS AND WIRE-GRID MODELING SIMULATION A structure combining an asymmetric and upside-down discone antenna was proposed for UWB system antennas and radio frequency monitoring. To stabilize the VSWR, a set of tapered cylindrical wires tilted at an angle of 20° was used instead of a disc plate or uniform cylindrical wires [40]. While slight discrepancies exist between measured and simulated results caused by mechanical inaccuracies and feeding problems that were not considered in the simulation, the antenna exhibits a 1:100 impedance bandwidth (180 MHz to 18 GHz) with VSWR below 2.5. 12.4.1 High Efficiency Moment Method Compared to narrowband antennas, in designing UWB antennas, calculating and evaluating an antenna’s characteristics over a wide frequency range is necessary. For example, input impedance and radiation characteristics must be calculated repeatedly for different frequencies during the optimization procedure in the design of antennas’ physical dimen-

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Figure 12.25 Geometry of the proposed staircase-bowtie planar antenna with a U-shaped slot: (a) top view; (b) side view; and (c) bottom view. (From: [36].  2006 IEEE. Reprinted with permission.)

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Figure 12.26 Cutting view of a three-dimensional antenna with a coaxial stub. (From: [38].  2005 IEEE. Reprinted with permission.)

sions and structures. To realize higher performance of electromagnetic numerical calculations, an efficient method of moment algorithm [23, 41] has been proposed. Because the impedance matrix’s rate of change in the frequency domain for segments divided by the moment method is relatively moderate, calculation time is reduced by interpolating the impedance matrix in a frequency range [23, 41, 42]. In actual implementation by a parallel processor system, the impedance matrices at the three frequency points are communicated through data exchange between processors, and the impedance matrix of the middle frequency is interpolated. 12.5 UWB ANTENNAS IN SPECIFIC WIRELESS ENVIRONMENTS 12.5.1 UWB Antennas Used in Unlicensed and Autonomous Wireless Environments Not requiring licenses for UWB systems gives the market an enormous advantage in expanding the system. However, not requiring licenses creates an inherent limitation to legislation in handling possible interferences in establishing the efficient use of a frequency spectrum. Therefore, ‘‘autonomous countermeasures against technological interference,’’ so to speak, at each wireless station including mobile terminals becomes important. In situations with many UWB devices being used simultaneously, such as in offices, the antenna’s role in solving interference problems is valuable. Therefore, it is important for an antenna to be ‘‘radio wave environmentally friendly’’ [11]. Fundamentally and ideally,

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an antenna emits radio waves only in a necessary direction. In theory, the use of an adaptive directional antenna could be required by regulation even for mobile terminals, for example. In practice, the use of an adaptive directional antenna is inseparable from economic efficiency since the cost of the technology strongly affects system penetration and penetration inversely affects the cost of the technology (backward) because of scale economies. Regarding this, the role of antennas for ‘‘radio wave resource sharing’’ will be considered in the last part of this chapter. The UWB system is one of the promising technologies for extremely high-speed wireless handheld terminals for WPAN. As is well known, the available transmission speed of a UWB system depends on the distance between the transmitting and receiving stations. The shorter the distance, the higher the available transmission speed achievable. Let us assume that UWB mobile terminals in use are being held by users (operators). The varied radiation characteristics of UWB antennas, caused by the presence of a human body, could influence the UWB system’s total transmission performance, especially with extreme high-speed transmission available when the communicating terminals are placed within a short range of each other. In this sense, evaluating the radiation characteristics of UWB antennas near the human body is important. Figures 12.27 and 12.28 illustrate two typical setups (or environments) for a WPAN system. The UWB antenna is located near a human body, in this case a human hand. The human body has two major influences on the electrical characteristics of the UWB antenna: (1) effects on input impedance and (2) effects on radiation pattern. 12.5.2 Measurements of Multipath Propagation Environments for UWB Antennas Assessing the performance of UWB antennas, including the propagation environments in which these antennas are used in specific wireless systems, is important. For example, UWB SIMO channel measurements were carried out with the time-domain channel sounder both in an anechoic chamber and in a real indoor environment, with a maximum Tx-Rx antenna separation of approximately 10m [43]. Several attempts have been made to evaluate UWB antennas in an actual environment. Using FDTD simulations to solve large-scale UWB electromagnetic propagation problems is neither easy nor practical; therefore, various stochastic UWB channel models have been proposed. Some include the Modified-SV model [44], the Stochastic tapped-delay-line (STDL) model [45], the ⌬-K model [46], and the Hybrid model [47]. Frequency domain field measurements were performed with a vector network analyzer, a 30-dB gain wideband power amplifier, a 32-dB gain low-noise amplifier, and a pair of vertically polarized wideband planar dipole antennas in various high-rise apartments [48]. Measurement results in the 3- to 10-GHz frequency band were processed to extract small-scale statistics that characterize the multipath behavior of the UWB channel. A generic statistical-based UWB channel model that incorporates multipath component clustering was also proposed based on the modification of the conventional S-V channel

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Figure 12.27 Image of UWB WPAN with shadowing setting. (From: [11].  2005 IEICE. Reprinted with permission.)

model. Additionally, cluster properties in home environments were investigated using a vector network analyzer combined with a uniform linear array antenna and a uniform rectangular array antenna at the transmitting and receiving sides, respectively [49]. Three types of propagation path scattering losses were derived and modeled from the double directional propagation measurement campaign. 12.5.3 Transmission Characteristics of UWB Antennas and Effects of the Human Body Time domain response of a pair of log spiral antennas was shown [50] as an example of a dispersive antenna, which covers the operating frequency range from 1 to 11 GHz. The cause of the dispersed received signal was the shift of the phase center from the bottom to the tip of the antenna as frequency increases. The shift of the phase center along the radiating element is one of the causes of the variation of the transmission characteristics as a function of look angle. Transmission characteristics between two monopole antennas placed on a rectangular ground plane (464 × 321 mm) with the distance between antennas of 236 mm were carried out using three types of monopoles: (1) a rectangular monopole

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Figure 12.28 Image of UWB WPAN with reflection setting. (From: [11].  2005 IEICE. Reprinted with permission.)

(13 × 16 mm), (2) a strip monopole (3 × 16 mm), and (3) a bi-arm rolled monopole (3 × 16 mm) [51]. The transfer responses for several combinations of the antennas were obtained using these monopoles. Measurements were taken, with antennas positioned face-to-face and side-by-side, and waveforms of received signals were compared for both single-band and multiband schemes [51]. The human hand’s influences on transmission characteristics were measured for both the shadowing setting and reflection setting. Measurements were taken with two elliptical disc monopole antennas (28 × 24 mm) placed on a square metallic plate (1m × 1m) with a 100-mm distance between the two antennas. The image setup pictures for both shadowing and reflection settings are shown in Figure 12.29. Measurement results are shown in Figure 12.30, where ‘‘free,’’ ‘‘shad’’ (or ‘‘ref’’), and ‘‘shadcomp’’ (or ‘‘refcomp’’) correspond to without hand, with hand placed 5 mm above the ground plate, and with hand firmly attached on the ground plate, respectively. Because of the size of the antenna used in this experiment, the frequency range with low transmission loss shifted to 20% lower than the original UWB frequency range, and the shadowing loss varied widely depending on the conditions with or without an air gap: the gap between the human hand and the ground plate). In the case of no air gap, the attenuation [shown in Figure 12.30 (a)] increased by about 15 dB. In the reflection

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Figure 12.29 Setups for S21 measurements: (a) shadowing setting; and (b) reflection setting. (From: [11].  2005 IEICE. Reprinted with permission.)

environment shown in Figure 12.29(b), the results for d = 20 mm are also shown. As shown in Figure 12.30(b), fewer level fluctuations were caused by the presence or absence of the air gap. Similar measurements were also performed with different measurement conditions to confirm the aforementioned trends in terms of attenuation and fluctuation. Figure 12.30(c, d) show the results for two elliptical disc monopole antennas (16 × 13 mm) placed on a rectangular metallic plate (1m × 0.5m) with a 300-mm distance between two antennas. The human body effects characterized by the near-field deterministic components, such as a human hand or a human torso, are also important in the ultra-high-speed shortrange UWB. The propagation characteristics of UWB signals specifically near a human head were studied for wireless body area networks (WBANs). The human head was used because the most important human communication organs, for example the eyes and the ears, are located there [52]. Both theoretical and experimental investigations were done to deduce the appropriate propagation model for the human head. Due to the head’s strong attenuation, it has been concluded that direct transmission through the head is negligible and diffraction is the major propagation mechanism. The human body’s effects on UWB signal propagation were measured with a diamond-shape dipole antenna under two multipath environments: (1) low multipath in an auditorium, and (2) dense multipath in an office [53]. Background noise measurements were also taken in advance to assess the level of possible existing interference signals.

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Figure 12.30 (a) Shadowing setting. (From: [11].  2005 IEICE. Reprinted with permission.) (b) Reflection setting. (From: [56].  2005 IEICE. Reprinted with permission.) (c) Shadowing setting with different measurement conditions. (d) Reflection setting with different measurement conditions.

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Figure 12.30 (continued).

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An antenna integrated into a model of a cell phone was attached to two locations: (1) the user’s belt at the hip, and (2) the user’s head. Horizontal patterns were measured while the user rotated on a turntable through 360° in 15° increments. It has also been concluded that a deep null of 23.6 dB on the radiation pattern in a light multipath environment is reduced to 6.8 dB in a dense multipath environment.

12.5.4 UWB Antennas Near the Human Body Frequency-domain on-body propagation channel measurements were performed for the 3- to 9-GHz range with a vector network analyzer and two different antennas: (1) printed horn-shaped, self-complementary antennas (HSCA), and (2) planar inverted cone antennas (PICA) [54]. Mean delay and root-mean-square delay spreads were extracted from the measured data for six receiving locations on real human candidates, with a fixed transmitting antenna attached to each candidate’s left front waist. The path loss exponent was 3.9 for the HSCA case and 2.6 for the PICA case. The mean root-mean-square delay spread was highest in the case where the receiving antenna was located at the back of the candidates and the major propagation mode was guided wave by the human body (surface wave). Two small planar UWB antennas near the human head were simulated with the FDTD method and compared with the measured results in terms of return losses and radiation patterns [55]. It has been concluded that while the human head slightly affected the impedance characteristics of the antennas under test, it significantly influenced radiation patterns as well as average and maximum antenna gains. Figure 12.31(a) shows an image of a wireless mobile terminal under practical use. To assess the effects of the human hand on the radiation characteristics of the wireless terminal, an elliptical slot antenna was selected and calculated with the FDTD method, including the simplified model of human hands shown in Figure 12.31(b, c) [56]. An elliptical slot dipole antenna (each ellipse measured 14 mm in length and 12 mm in width) was placed on a ground plate (100 × 200 mm) simulating a wireless terminal’s PC board. Calculated results of the electrical field without hand for t = 0.6227 ns and t = 0.9101 ns are shown in Figure 12.32(a, b), respectively. The calculations were performed on the X-Y plane (x = −150–+150 mm, y = −30–+100 mm) at z = 0 for the major electric field component Ex. The results with hand are also shown in Figure 12.35(c, d), respectively. The edge-diffracted waves can be seen in Figure 12.32(a) and the waveform distortion caused by the so-called ‘‘ringing’’ was also observed after t = 0.9 ns even without hand as shown in Figure 12.32(b). As can be seen in Figure 12.32(c, d), the human hand placed in the vicinity of the antenna blocks the propagation path from the edge and the major contribution of the radiation is from the aperture between two human hands.

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Figure 12.31 Image picture of wireless mobile terminal and analytical model with human hands: (a) mobile terminal; (b) analytical model (3D view); and (c) analytical model (top view). (From: [56].  2005 IEICE. Reprinted with permission.)

Figure 12.32 Electric fields for the analytical model shown in Figure 12.31. (a) Without hand: (t = 0.6227 ns). (b) Without hand: (t = 0.9101 ns). (c) With hand: (t = 0.6227 ns). (d) With hand: (t = 0.9101 ns). (From: [56].  2005 IEICE. Reprinted with permission.)

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12.6 UWB ANTENNA EVALUATION INDEXES The fractional bandwidth is one of the key parameters of the wireless systems and is determined by the ratio of the bandwidth to the center frequency. While, the arithmetic average of the upper and lower frequencies, f c = ( f H − f l )/2, is often used in defining the center frequency, the geometric average, f c = √ f H × f l , is an alternate definition of the center frequency when frequency is considered on a logarithmic scale [57]. Some wellknown broadband antennas do not show appropriate phase characteristics. For example, the time domain response of a log periodic antenna exhibits a nonlinear phase shift that causes ringing on the received or transmitted waveform even if the antenna shows wideband operation in amplitude. Given the specific amplitude and phase responses in the frequency domain for two UWB antennas, comparing two antennas using certain specific indexes or parameters that are not functions of frequency is convenient. An example of an evaluation index (12.3) was proposed [58] with average amplitude and phase characteristics over the entire UWB frequency band ( f 1 : 3.1 GHz– f 2 : 10.6 GHz). S 21 (␻ ) =

␭ e j (␾ t (␻ ) + ␾ r (␻ )) × 4␲ R

√冠1 − | S11 (␻ ) | 2 冡冠1 − | S22 (␻ ) | 2 冡 Gt (␻ ) Gr (␻ ) (12.1)

GD (␻ ) =

d (␾ (␻ ) + ␾ r (␻ )) d␻ t

(12.2)

f2

1 EvaluationIndex = f2 − f1

冕冠

g ( f ) − g ( f ) 冡 df 2

(12.3)

f1

where f2

1 g( f ) = f2 − f1

冕

g ( f ) df

(12.4)

f1

| S 21 (␻ ) | corresponds to the frequency characteristics of amplitude response and GD (␻ ) to group delay, which is the derivative of the phase response. To obtain the evaluation index for amplitude, | S 21 (␻ ) | is substituted in g ( f ), whereas CD (␻ ) in g ( f ) for the phase index. Flat amplitude response and flat group delay, or constant gradient of phase, over the entire frequency spectrum of transmitted signals must be satisfied to realize so-

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called nondistortion transmission. In short, low VSWR for an antenna is a necessary condition for a UWB system, but not a sufficient condition for a good UWB antenna. The fidelity between two waveforms, x (t ) and y (t ), is defined as follows and corresponds to an index that compares only the shapes of both waveforms, not their amplitudes [59]. The fidelity factor may be used as an evaluation index for total system pulse distortions, in which transfer characteristics of transmitting and receiving antennas are included, since a correlation calculation with the template waveform is commonly used at the receiver side in a UWB system. Variation in the magnitude or phase of the received signal decreases the fidelity index. ∞

冕

F = max

x (t ) ⭈ y (t − ␶ ) dt

−∞

√冕 | ∞

(12.5)

∞

x (t )2 dt | ⭈

−∞

冕|

y (t )2 dt |

−∞

12.7 UWB ANTENNA MEASUREMENTS 12.7.1 Radiation Pattern Measurements The radiation pattern measurements of a prototype portable antenna are known to suffer from the influence of the attached feeding cable for the RF source. A countermeasure to reduce the cable leakage current along the feeding cable is to use balun choke, which generally has a narrow bandwidth. A dual-band balun choke [shown in Figure 12.33(a)] placed on the feeding cable for frequency bands around 920 and 1,795 MHz was experimentally investigated with a three-dimensional radiation pattern measurement [60]. Although the applicable frequency range is limited, using ferrite cores is a possible option for reducing the undesired current effects during antenna measurements. The effectiveness of this method depends on the complex permeability of the ferrite core material as well as the arrangement of the ferrite cores. To apply this method to scaledup model measurements, several ferrite core loading configurations along a coaxial cable were examined for errors both on radiation pattern and on input impedance in the frequency range between 200 MHz and 2 GHz [61]. As a result, localized suppression of the cable leakage current near the antenna is important for impedance measurements, while broad suppression of the cable leakage current along the entire cable is necessary when measuring the radiation pattern. Additionally, appropriate ferrite core structures and arrangements have been proposed [61] to improve the accuracy of wideband antenna measurements. Antenna pattern measurement with a photonic sensor is another option, combined with a compact spherical near-field measurement system for UWB antennas [62]. The

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Figure 12.33 (a) Cross-section of the dual-band balun where wall thickness is 1.5 mm and the diameter of the RF cable is 2.2 mm. (From: [60].  2006 IEEE. Reprinted with permission.) (b) Microstrip-type balun. (From: [65].  2006 IEEE. Reprinted with permission.)

antenna size fabricated on the photonic sensor was 2.4 × 2 mm and could be considered an infinitesimal electric dipole antenna with no metallic feeding cable, which minimally influences the antenna under test even at the highest frequency range of a UWB system. 12.7.2 Impedance Measurements As is well known, balun is a combined abbreviation of the words [63] ‘‘balance’’ and ‘‘unbalance.’’ The role of balun in antenna measurements is extremely important for both impedance and radiation pattern measurements, especially for balanced-type antennas since standard RF measurement instruments have unbalanced RF coaxial connectors. The performance of a balun is dependent on frequency and is, in some cases, very sensitive to frequency. For example, the bazooka balun, one of the most common baluns used in practical applications, must be designed with the length of the bazooka equal to a quarter wavelength at the design frequency. This is because the bazooka length is the key parameter for a large impedance factor in suppressing the undesired current flow on the surface of the coaxial cable’s outer conductor. While a dual-band balun has been proposed [60] and

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is useful to measure dual-band mobile terminals, such as 920 and 1,795 MHz, such baluns are inherently of narrow bandwidth and will not be applicable for measuring UWB antennas. A balun’s basic function is the transition from balanced mode to unbalanced mode and vice versa. Modifying the structural configuration from a coaxial (unbalanced) line to a parallel (balanced) line is used to realize a smooth modification of the electromagnetic mode from unbalanced to balanced. This is called a taper or cutaway balun. Gradually modifying the structure reduces internal reflections along the taper section, which is about a half wavelength of the lowest frequency [64]. The same principle was also applied to a microstrip balun configuration [shown in Figure 12.33(b)] for a UWB TEM Horn antenna [65]. To reduce the balun’s length, a tapered coaxial balun with an alumina dielectric (⑀ r = 9.5) between the center conductor and the tapered conductor was proposed [66] and measured in the UHF band. Measured results demonstrated that the efficiency of the proposed balun increased by as much as 30% in comparison to the conventional balun. A compact on-chip 24-GHz balun with a standard IC fabrication process was proposed and measured to evaluate an on-chip antenna on the same Si substrate [67]. Additionally, a practical method of characterizing a balun with an ordinary single-ended, two-port vector network analyzer is provided [68]. 12.7.3 Scale Model Measurements A scaled-down model is primarily used to reduce the antenna system’s total size during measurements, whereas a scaled-up model is often used when magnifying the physical dimensions of the antenna under test becomes necessary. Several reasons exist for using a scaled-up model: (1) restriction of the upper frequency limit of the measurement instruments, such that a vector network analyzer with an upper frequency limit of 8 GHz may be used for a UWB antenna’s simulated measurements with a scaling factor of 2; and (2) restriction on manufacturing accuracies, such as the accuracies of feeding point structures. While scale model measurements are convenient and easy to apply to antenna systems consisting of only metallic material, the method will not be applicable in measuring the effects of a human body on an antenna system without appropriate scale model phantoms. To apply the scale model to measurements for antennas used near a human body, developing scale model phantoms for various conditions, including frequency bands and scale factors, is important. Table 12.1 summarizes the required relationships between the electrical parameters for a full-scale model and those for a scale model with scale factor (or magnifying factor: k) 1/k. A basic study on the TX-151 scaled-down model phantom was performed based on a series of measurements for various combinations of composition amounts. The extracted results from a multiple regression analysis are reported [69] for the scale factors from 1/2 to 1/15.

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Table 12.1 Geometrical Scale Model

Parameter Length Wavelength Frequency Permeability Permittivity Conductivity

Original Model (Full-Scale) l ␭ f ␮ ⑀ ␴

Scaled Model l ′ = l /k ␭ ′ = ␭ /k f ′ = kf ␮′ = ␮ ⑀′ = ⑀ ␴ ′ = k␴

12.7.4 Impedance Measurements with Two Coaxial Cables An alternative to using a physical balun for input impedance measurements is the S-Parameter Method [70]. Also, it is possible and convenient to use a two-port vector network analyzer with built-in functionality based on the method. The basic principle of this measurement method is to view a balanced antenna as a two-port network that can be determined with a standard vector network analyzer. In this method, two unbalanced coaxial lines from a two-port vector network analyzer are connected to the antenna. Figure 12.34 shows a measurement setup using two coaxial cables. As seen in this figure, two semi-rigid coaxial test cables (2.19 mm in diameter) with SMA Plug connecters are used for measurements with a two-port network analyzer with the built-in functionality. Before measurements, full port calibration was done using an automatic calibration unit. Step 1: Two semi-rigid extension cables [shown in Figure 12.35(a)] with type and length equal to those of the two test cables [shown in Figure 12.35(b)] are connected to the measurement cables. The other ends of the measurement cables are connected to port #1 and port #2 of the vector network analyzer. Step 2: The open ends of the two semi-rigid extension cables are connected to the automatic calibrator to perform full port calibration. Step 3: Two extension cables are removed and replaced by two semi-rigid coaxial test cables and the other ends of the two center conductors are open circuits. Step 4: Electrical length compensation has been achieved with an open or short condition depending on the practical condition at the two center conductors. Step 5: A pair of radiating elements of an antenna under test is connected to two center conductors of the two semi-rigid coaxial test cables to perform the antenna’s impedance measurement. Measurements for a scale model [shown in Figure 12.35(c) with magnifying factor of 8] of a circular disc-triangular plate antenna were obtained over the 250- to 1,500-MHz frequency range. Figure 12.35 illustrates the antenna’s impedance versus frequency where

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Figure 12.34 Measurement setup with two unbalanced coaxial cables. (a) A pair of semi-ridged coaxial extension cables. (b) A pair of semi-rigid coaxial test cables. (c) A circular disc-triangular plate radiating elements with magnifying factor of 8 for scale-model measurement where two coaxial cables (or cable for unbalanced measurement) were arranged to be perpendicular to the plane of the radiating elements. (d) Close-up of the feeding point with two coaxial cables as an example of the measurement setup for the S-Parameter Method.

‘‘Model-S’’ corresponds to the measurements with S-Parameter Method and ‘‘Calc.’’ to the calculated results with FDTD, respectively. ‘‘Model-1’’ is the measured impedance of the antenna with unbalanced measurement in a setting where the coaxial cable’s center conductor is connected to the lower edge of a circular disc element, and the outer conductor to the apex of a triangle element. ‘‘Model-2’’ evaluates the opposite setting, where the center conductor is connected to the apex of the triangle element, and the outer conductor to the circular disc element. Because of the large magnifying factor of 8 relative to the full-scale model, the measured results using two coaxial cables are in reasonable agreement with the calculated results, even for the reactance measurement. From the results shown in Figure 12.35, one might conclude that the better agreement is in resistance than in reactance because the physical structure and the reactive energy near the feeding point influence the reactive part of the antenna’s input impedance. This highlights the importance of the scale model measurement for an intensive experiment on an antenna’s feeding structure. While the applicable frequency of the measurement with two coaxial cables is currently limited, expanding the applicable frequency to an upper frequency range is another expected issue.

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Figure 12.35 Measurement results for the antenna shown in Figure 12.34(c) with calculated results: (a) resistance; and (b) reactance.

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12.8 INTEGRATED ANTENNA DESIGN APPROACH BASED ON LSI TECHNOLOGY Two important conditions are to be satisfied regarding the output frequency spectrum of the UWB transmitter: (1) the mandatory FCC spectrum mask, and (2) optimized SNR at the receiver side. An integrated design approach [71] is proposed for planar UWB antennas under the FCC spectrum mask constraint to obtain near-maximal SNR. Integrated compact wireless LSIs, in which on-chip antennas are embedded, have always been the prime goal for antenna system engineers and wireless system designers. Thanks to the recent rapid progress of RF-CMOS technologies, and due to better linearity and spurious response, differential configuration (or architecture) is coming into use even for RF circuits. However, as is well known, a coaxial cable and a microstrip line are used as the most common RF connections and are categorized as an unbalanced transmission line. Therefore, differential configuration often requires interfacing with unbalanced circuits, and a balun is typically used as an interface between a differential line and an unbalanced line. Antenna engineers are motivated to design a mobile terminal antenna’s input impedance to be ‘‘unbalanced 50 ohms’’ for practical reasons. Actually, the choice of ‘‘unbalanced 50 ohms’’ enables a terminal’s convenient connection to measurement instruments for both production and industrial standard tests required by regulations. However, an integrated antenna approach offers an antenna engineer additional degrees of freedom in design goals. Then, an unbalanced 50-ohm transmission line need not be mandatory for antenna specifications. For example, the complete IR-UWB impulse generator with Bi-CMOS process was designed and tested to be combined with differentially fed antennas having input impedance of 100 ohms [72]. 12.9 RADIO WAVE RESOURCE SHARING WITH TECHNOLOGY LEADERSHIP AND THE ROLE OF THE ANTENNA [11] Because telecommunication regulations have been formulated based on available technologies, with economic efficiency taken into consideration at any given time, communications regulations are forecasted to be deregulated because of a great controlling power being assumed by the development of the computer, which enables enormous processing power at the mobile terminal side. This trend accelerates so-called end-to-end oriented protocols in wireless communication that challenge the validity of the natural monopoly enjoyed by wireless services. This creates the possibility that the balance for coexistence based on regulations and laws, including exclusive and conventional frequency allocation, is migrating towards a balance for coexistence by higher technology and multistandards, with the purpose of adaptively sharing radio wave resources. Interference countermeasures for UWB greatly depend on technology that will be developed in the future. Given the competitive nature of the market, products with the

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technology to eliminate or alleviate the problems unique to UWB are expected to become advantageous to surviving in the market. Seen in this light, antennas with a superior interference adaptability are desired, leaving many opportunities for antenna engineers to address new historic tasks related to practical applications of technologies to portable devices. In particular, once licensing regulations for conventional wireless communication were removed, it became necessary for multiple devices and independent systems to coexist within economic principles. Given this background, it is fundamentally important that systems do not radiate radio waves in a direction other than originally intended; also important is the study of the original role of antennas as the only device that can control the shape of radiation. Directivity, including adaptability, is a function unique to the antenna and key to fully utilizing the scarce resource in wireless communication— frequency spectrum—for future society. REFERENCES [1] ‘‘First Report and Order, Revision of Part 15 of the Commission’s Rule Regarding Ultra Wide Band Transmission Systems,’’ Federal Communications Commission, FCC 02-48. April 22, 2002. [2] Win, M. Z., and R. A. Scholtz, ‘‘Impulse Radio: How It Works,’’ IEEE Communications Letters, Vol. 2, No. 2, February 1998, pp. 36–38. [3] Mushiake, Y., ‘‘The Input Impedance of a Slit Antenna,’’ Joint Convention Record of Tohoku Sections of IEE and IECE of Japan, June 1948, pp. 25–26. [4] Mushiake, Y., Self-Complementary Antennas: Principle of Self-Complementary for Constant Impedance, New York: Springer, 1996. [5] Liang, J., et al., ‘‘Study of CPW-Fed Circular Disc Monopole Antenna for Ultra Wideband Applications,’’ IEE Proc. Microwaves Ant. Propag., Vol. 152, No. 6, December 2005, pp. 520–526. [6] Hertel, T. W., and G. G. Smith, ‘‘On the Convergence of Common FDTD Feed Models for Antennas,’’ IEEE Trans. Ant. Propag., Vol. 51, No. 8, August 2003, pp. 1771–1779. [7] Venkatarayalu, N. V., et al., ‘‘Numerical Modeling of Ultrawide-band Dielectric Horn Antennas Using FDTD,’’ IEEE Trans. Ant. Propag., Vol. 52, No. 5, May 2004, pp. 1318–1323. [8] Lee, M.-H., et al., ‘‘Modeling and Investigation of a Geometrically Complex UWB GPR Antenna Using FDTD,’’ IEEE Trans. Ant. Propag., Vol. 52, No. 8, August 2004, pp. 1983–1991. [9] Watanabe, S., and M. Taki, ‘‘An Improved FDTD Model for the Feeding Gap of a Thin-Wire Antenna,’’ IEEE Microwave and Guided Wave Letters, Vol. 8, No. 4, April 1998, pp. 152–154. [10] Liang, J., et al., ‘‘Analysis and Design of UWB Disc Monopole Antennas,’’ IEE Ultra Wideband Communications Technologies and System Design, July 2004, pp. 103–106. [11] Maeda, T., ‘‘UWB Antennas: Antenna and Propagation Technologies for Ubiquitous Ultra High Speed Wireless Communications and Perspectives,’’ IEICE Trans., Vol. J88-B No. 9, September 2005, pp. 1586–1600. [12] Kraus, J. D., and R. J. Marhefka, Antennas: For All Applications, 3rd ed., New York: McGraw-Hill, 2002, p. 60. [13] Kraus, J. D., and R. J. Marhefka, Antennas: For All Applications, 3rd ed., New York: McGraw-Hill, 2002, p. 43. [14] Lee, J. W., C. S. Cho, and J. Kim, ‘‘A New Vertical Half Disc-Loaded Ultra-Wideband Monopole Antenna (VHDMA) with a Horizontally Top-Loaded Small Disk,’’ IEEE Ant. Wireless Propag. Letters, Vol. 4, 2005, pp. 198–201.

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[15] Low, Z. N., J. H. Cheong, and C. L. Law, ‘‘Low-Cost PCB Antenna for UWB Applications,’’ IEEE Trans. Ant. Propag. Letters, Vol. 4, 2005, pp. 237–239. [16] Chen, Z. N., et al., ‘‘Annular Planar Monopole Antennas,’’ IEE Proc. Microwaves Ant. Propag., Vol. 149, No. 4, August 2002, pp. 200–203. [17] Suh, S.-Y., W. L. Stutzman, and W. A. Davis, ‘‘A New Ultrawideband Printed Monopole Antenna: The Planar Inverted Cone Antenna (PICA),’’ IEEE Trans. Ant. Propag., Vol. 52, No. 5, May 2004, pp. 1361–1364. [18] Wu, X., and Z. N. Chen, ‘‘Comparison of Planar Dipoles in UWB Applications,’’ IEEE Trans. Ant. Propag., Vol. 53, No. 6, June 2005, pp. 1973–1983. [19] Valderas, D., et al., ‘‘Design of UWB Folded-Plate Monopole Antennas Based on TML,’’ IEEE Trans. Ant. Propag., Vol. 54, No. 6, June 2006, pp. 1676–1687. [20] Gibson, P. J., ‘‘The Vivaldi Aerial,’’ Proc. 9 the European Microwave Conf., 1979, pp. 101–105. [21] Yngvesson, K. S., et al., ‘‘Endfire Tapered Slot Antennas on Dielectric Substrates,’’ IEEE Trans. Ant. Propag., Vol. AP-33, No. 12, December 1985, pp. 1392–1400. [22] Shin, J., and D. H. Schaubert, ‘‘A Parameter Study of Stripline-Fed Vivaldi Notch-Antenna Arrays,’’ IEEE Trans. Ant. Propag., Vol. 47, No. 5, May 1999, pp. 879–886. [23] Newman, E. H., ‘‘Generation of Wide-Band Data from the Method of Moments by Interpolating the Impedance Matrix,’’ IEEE Trans. Ant. Propag., Vol. 36, No. 12, December 1988, pp. 1820–1824. [24] Chiappe, M., and G. L. Gragnani, ‘‘Vivaldi Antennas for Microwave Imaging: Theoretical Analysis and Design Considerations,’’ IEEE Trans. Instrument. Measurement, Vol. 55, No. 6, December 2006, pp. 1885–1891. [25] Nikalaou, S., et al., ‘‘Conformal Double Exponentially Tapered Slot Antenna (DESTA) on LCP for UWB Applications,’’ IEEE Trans. Ant. Propag., Vol. 54, No. 6, June 2006, pp. 1663–1669. [26] Ma, T.-G., and C.-H. Tseng, ‘‘An Ultrawideband Coplanar Waveguide-Fed Tapered Ring Slot Antenna,’’ IEEE Trans. Ant. Propag., Vol. 54, No. 4, April 2006, pp. 1105–1110. [27] Kwon, D., and Y. Kim, ‘‘Suppression of Cable Leakage Current for Edge-Fed Printed Dipole UWB Antennas Using Leakage-Blocking Slots,’’ IEEE Trans. Ant. Wireless Propag. Letters, Vol. 5, No. 1, December 2006, pp. 183–186. [28] Angelopoulos, E. S., et al., ‘‘Circular and Elliptical CPW-Fed Slot and Microstrip-Fed Antennas for Ultrawideband Applications,’’ IEEE Trans. Ant. Wireless Propag. Letters, Vol. 5, No. 1, December 2006, pp. 294–262. [29] Sanz-Izquierdo, B., et al., ‘‘Compact UWB Monopole for System-on-Package Applications,’’ IEEE Antenna Technology Small Antennas and Novel Metamaterials, March 2006, pp. 68–71. [30] Ren, Y.-J., and K. Chang, ‘‘An Annual Ring Antenna for UWB Communications,’’ IEEE Trans. Ant. Wireless Propag. Letters, Vol. 5, No. 1, December 2006, pp. 274–276. [31] Yoon, H., et al., ‘‘A Study on the UWB Antenna with Band-Rejection Characteristics,’’ IEEE AP-S Int. Symp. Digest, Vol. 2, June 2004, pp. 1784–1787. [32] Schantz H. G., G. Wolence, and E. M. Myszka, III, ‘‘Frequency Notched UWB Antennas,’’ IEEE Ultra Wideband Systems and Technologies, November 2003, pp. 214–218. [33] Yoon, I.-J., et al., ‘‘Ultra Wideband Tapered Slot Antenna with Band-Stop Characteristic, ‘‘ IEEE AP-S Int. Symp. Digest, Vol. 2, June 2004, pp. 1780–1783. [34] Schantz, H., The Art and Science of Ultrawideand Antennas, Norwood, MA: Artech House, 2005, pp. 277–282. [35] Qiu, J., et al., ‘‘A Planar Monopole Antenna Design with Band-Notched Characteristics,’’ IEEE Trans. Ant. Propag., Vol. 54, No. 1, January 2006, pp. 288–291. [36] Cho, Y. J., et al., ‘‘A Miniature UWB Planar Monopole Antenna with 5-GHz Band-Rejection Filter and the Time-Domain Characteristics,’’ IEEE Trans. Ant. Propag., Vol. 54, No. 5, May 2006, pp. 1453–1460. [37] Kerkhoff, A., and H. Ling, ‘‘Design of a Planar Monopole Antenna for Use with Ultra-Wideband (UWB) Having a Band-Notched Characteristic,’’ IEEE AP-S Int. Symp., June 2003, pp. 830–833.

586

[38] Pele´, I., A. Chousseaud, and S. Toutain, ‘‘Multi-Band Ultra Wide Band Antennas,’’ Proc. The European Conference on Wireless Technology, October 2005, pp. 281–284. [39] Schantz, H., The Art and Science of Ultrawideband Antennas, Norwood, MA: Artech House, 2005, pp. 283–285. [40] Kim, K.-H., J.-U. Kim, and S. Park, ‘‘An Ultrawide-Band Double Discone Antenna with the Tapered Cylindrical Wires,’’ IEEE Trans. Ant. Propag., Vol. 53, No. 10, October 2005, pp. 3403–3406. [41] Virga, K. L., and Y. Rahamat-Samii, ‘‘Efficient Wide-Band Evaluation of Mobile Communications Antennas Using [Z] or [Y] Matrix Interpolation with the Method of Moments,’’ IEEE Trans. Ant. Propag., Vol. 47, No. 1, January 1999, pp. 65–76. [42] Mittra, J. M., and N. Huang, ‘‘Improving the Convergence of the Iterative Solution of Matrix Equations in the Method of Moments Formulation Using Extrapolation Techniques,’’ IEE Proc. Microwaves Ant. Propag., Vol. 150, No. 4, August 2003, pp. 253–257. [43] Keignart, J., et al., ‘‘UWB SIMO Channel Measurements and Simulations,’’ IEEE Trans. Microwave Theory and Techniques, Vol. 54, No. 4, April 2006, pp. 1812–1819. [44] Foerster, J., (ed.), ‘‘Channel Modeling Sub-Committee Report Final,’’ IEEE802.15-02/490, November 2002. [45] Cassioli, D., M. Z. Win, and A. F. Molisch, ‘‘ The Ultra-Wide Bandwidth Indoor Chanel—From Statistical Model to Simulation,’’ IEEE Journal on Selected Areas in Communications, Vol. 20, No. 6, 2002, pp. 1247–1257. [46] Pendergrass, M., ‘‘Empirically Based Statistical Ultra-Wideband Channel Model,’’ IEEE P802.15-02/240-SG3a, July 2002. [47] Kunisch, J., and J. Pamp, ‘‘Radio Channel Model for Indoor UWB WPAN Environments,’’ IEEE P802.15-02/281-SG3a, June 2002. [48] Chong, C.-C., and S. K. Yong, ‘‘A Generic Statistical-Based UWB Channel Model for High-Rise Apartments,’’ IEEE Trans. Ant. and Propag., Vol. 53, No. 8, August 2005, pp. 2389–2399. [49] Haneda, K., J. Tadada, and T. Kobayashi, ‘‘Cluster Properties Investigated from a Series of Ultrawideband Double Directional Propagation Measurements in Home Environments,’’ IEEE Trans. Ant. Propag., Vol. 54, No. 12, December 2006, pp. 3778–3788. [50] Schantz, H., The Art and Science of Ultrawideand Antennas, Norwood, MA: Artech House, 2005, pp. 42–43. [51] Chen, Z. N., ‘‘Novel Bi-Arm Rolled Monopole for UWB Applications,’’ IEE Trans. Ant. Propag., Vol. 53, No. 2, February 2005, pp. 672–677. [52] Zasowski, T., et al., ‘‘UWB Signal Propagation at the Human Head,’’ IEEE Trans. Microwave Theory and Techniques, Vol. 54, No. 4, April 2006, pp. 1836–1845. [53] Welch, T. B., et al., ‘‘The Effects of the Human Body on UWB Signal Propagation in an Indoor Environment,’’ IEEE Journal on Selected Areas in Communications, Vol. 20, No. 9, December 2002, pp. 1778–1782. [54] Alomainy, A., et al., ‘‘Comparison Between Two Different Antennas for UWB On-Body Propagation Measurements,’’ IEEE Trans. Ant. Wireless Propag. Letters, Vol. 4, 2005, pp. 31–34. [55] Chen, Z. N., et al., ‘‘Small Planar UWB Antennas in Proximity of the Human Head,’’ IEEE Trans. Microwave Theory and Techniques, Vol. 54, No. 4, April 2006, pp. 1846–1857. [56] Asanuma, K., and T. Maeda, ‘‘A Basic Study on the Short Range Transmission Characteristics of High Speed Wireless Terminals Including the Effects of the Metallic Case and Human Hands,’’ IEICE Trans., Vol. J88-B, No. 9, September 2005, pp. 1650–1661. [57] Schantz, H., The Art and Science of Ultrawideand Antennas, Norwood, MA: Artech House, 2005, pp. 38–40. [58] Wu, X. H., and Z. N. Chen, ‘‘Design and Optimization of UWB Antennas by a Powerful CAD Tool: PULSE KIT,’’ IEEE AP-S Int. Symp. Digest, Vol. 2, June 2004, pp. 1756–1759. [59] Klemm, M., et al., ‘‘Novel Small-Sized Directional Antenna for UWB WBAN/WPAN Applications,’’ IEEE Trans. Ant. Propag., Vol. 53, No. 12, December 2005, pp. 3884–3896.

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[60] Icheln, C., J. Krogerus, and P. Vainikainen, ‘‘Use of Balun Chokes in Small-Antenna Radiation Measurements,’’ IEEE Trans. Instrument. Measurement, Vol. 53, No. 2, April 2004, pp. 498–506. [61] Kozutsumi, N., and T. Maeda, ‘‘The Influence of the Leakage Current Along a Feed Cable and Improvement Effects on the Evaluation of Antenna Characteristics Using Ferrite Cores,’’ IEICE Trans., Vol. J90-B, No. 9, September 2007, pp. 896–905. [62] Hirose, M., S. Kurokawa, and K. Komiyama, ‘‘Compact Spherical Near-Field Measurement System for UWB Antennas,’’ IEEE AP-S Int. Symp. Digest, Vol. 2B, July 2005, pp. 692–695. [63] Kraus, J. D., and R. J. Marhefka, Antennas: For All Applications, 3rd ed., New York: McGraw-Hill, 2002, p. 803. [64] Duncan, J. W., and V. P. Minerva, ‘‘100:1 Balun Transformer,’’ Proc. IRE, February 1960, pp. 156–164. [65] Chung, K., S. Pyun, and J. Choi, ‘‘Design of an Ultrawide-Band TEM Horn Antenna with a MicrostripType Balun,’’ IEEE Trans. Ant. Propag., Vol. 53, No. 10, October 2006, pp. 3410–3413. [66] Choi, J.-G., S.-H. Yi, and K.-H. Kim, ‘‘Development of a Novel Tapered Balun for the UWB UHF Coupler,’’ IEEE Power Modulator Symp., May 2004, pp. 493–496. [67] Shamim, A., et al., ‘‘On-Chip Antenna Measurements: Calibration and De-embedding Considerations,’’ IEEE Instrumentation and Measurement Technology Conference, May 2005, pp. 463–466. [68] Belkin, S., ‘‘Differential Circuit Characterization with Two-Port S-Parameters,’’ IEEE Microwave Magazine, Vol. 7, No. 6, December 2006, pp. 86–99. [69] Yamamoto, S., D. Ushirogochi, and T. Maeda, ‘‘A Basic Study of Scale Model Phantom,’’ IEICE Trans., Vol. J89-B, No. 9, September 2006, pp. 1837–1841. [70] Meys, R., and F. Janssens, ‘‘Measuring the Impedance of Balanced Antennas by an S-Parameter Method,’’ IEEE Ant. Propag. Magazine, Vol. 40, No. 6, December 1998, pp. 62–65. [71] Lu, G., P. Spasojevic, and L. Greenstein, ‘‘Antenna and Pulse Designs for Meeting UWB Spectrum Density Requirements,’’ IEEE Ultra Wideband Systems and Technologies, November 2003, pp. 162–166. [72] Bagga, S., et al., ‘‘Codesign of an Impulse Generator and Miniaturized Antennas for IR-UWB,’’ IEEE Trans. Microwave Theory and Techniques, Vol. 54, No. 4, April 2006, pp. 1656–1666.

Chapter 13 Antennas for RFID Young Joong Yoon

Radio frequency identification (RFID) is currently one of the growing technologies for wireless identification and it brings considerable influence on areas of antenna development. The aim of this chapter is to provide antenna designs with the most comprehensive knowledge. This chapter is designed to introduce the RFID system briefly, and to provide the design and characteristics of many types of reader and tag antennas so that users can acquire an overall understanding of antennas in RFID areas. Moreover, information about reader antennas related to mobile communication for mobile RFID systems, characteristics of tag antennas on metallic surface, and measurement for tag antennas are included. 13.1 THE CHARACTERISTICS OF AN RFID SYSTEM 13.1.1 What Is RFID? An RFID system is a kind of wireless communication system that identifies an object without contact. RFID technology is next generation technology that will replace the BarCode system, as well as improve the slow recognition speed and increase the small storage capacity of the current system. Connected to a network, it will be applicable to various areas such as industry, distribution, public facilities, military services, and telecommunication. The RFID system was developed in the 1970s to trace missile trajectories. In the 1980s, as the size and the cost of RFID tags were reduced, RFID started seeing use for animal management, distribution of products, and other industries. In the 1990s, with the development of radio frequency technologies, cheap, high technology tags were developed 589

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and various tag types such as cards, labels, and coins appeared. In the 2000s, as wireless recognition technology becomes increasingly important, RFID has been developed as a core technology in various systems such as electronic cash, distribution, transportation, and security systems. An RFID system is shown in Figure 13.1. It is composed of a RFID tag that is attached to an object with its information, a reader that reads the information on the tag by RF communication, and a server that processes the information from the reader. Once the reader transmits an electromagnetic wave to the tag, the tag is activated by the energy of the continuous wave signal. The activated tag transmits its information back to the reader. The reader receives the transmitted energy from the tag and thereby obtains the information of the object to which the tag is attached. Furthermore, by feeding the tag information onto a network, the tag can act as a sensor. In this manner, the RFID system is expected to be the core technology of Ubiquitous Sensor Networks (USNs), which can communicate whenever and wherever. The reader is composed of RF circuits, a modulator/demodulator, a time signal processing module, and a protocol processor. The accuracy of the reader is affected by antenna performance and the surrounding environment. The tag is composed of a chip, which stores the unique identification code of the tag, and an antenna. The tag can be divided into active and passive tag, according to the existence of a power supply device. Generally, active tags are read/write type and offer advantages such as longer read range and the ability to reduce the required operating power of the reader. However, they have drawbacks of limited operating time, high cost, and larger size since a battery is housed inside the tag. Passive tags are read-only type and have strengths including light weight, low cost, and the ability to be used for a very long time. However, they have drawbacks such as short read range and high power consumption in the reader. Figure 13.2 shows an example of the passive tag attached to a piece of luggage, and it is used at the airport to check the departure and arrival of the luggage. The RFID system can be used for a wide range of applications. For example, it can be used in process management in industry, supply chain management (SCM), access

Figure 13.1 RFID system and Ubiquitous Sensor Network.

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Figure 13.2 An example of RFID tag: it is attached to a carrier used for an automatic luggage system at the airport.

control, storage/stock control in the distribution industry, delivery service control, library management, parking control, automatic road toll charging, finance, electric commercial transactions, electric money, and patient supervision in hospitals. Standardization of RFID technology has been ongoing in ISO/IEC and EPCglobal, international standardization organizations, since the latter half of the 1990s. Also, research on applications of RFID has been proceeding in other international organizations such as ITU and IEFT. Meanwhile, studies on guidelines for RFID have been carried out at the UID center in Japan and ETSI in Europe [1]. 13.1.2 Operating Frequencies The allocation of RFID frequencies is done independently in respective nations globally. RFID is mostly utilized in five frequency ranges: low frequency (125 kHz, 135 kHz), high frequency (13.56 MHz), amateur band (433.92 MHz), ultra high frequency (860 to 960 MHz), and microwave frequency (2.45 GHz). Each frequency range has different wavelengths and different abilities to receive signals across distances and to pass through opaque materials such as metals, sodium, graphite, and liquids. Since the characteristics of the frequencies in different bands vary, the potential and existing applications are different. RFID systems that use 125 kHz and 135 kHz in the low frequency range are the oldest in existence. The communications are governed by ISO specification 18000-2, and they comply with a uniform standard around the world. The communication range is only several tens of centimeters and the tag read rate is relatively slow since it uses an inductively coupling operating mechanism. These systems are mostly used for animal

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tagging, access control, and vehicle immobilizers. RFID systems that use 13.56 MHz in the high frequency range are governed by three publications: ISO/IEC specification 18000-3, ISO/IEC specification 15693, and ISO/IEC specification 14443, Parts A and B. The RFID systems are operated by a magnetic coupling mechanism and are used for smart cards, access control, luggage control, biometric identification, libraries, and apparel management systems. RFID systems that use 433.92 MHz in the amateur band frequency range are governed by ISO/IEC specification 18000-7. The wavelength of this frequency band is about a meter, and can propagate around obstacles such as vehicles and containers. These RFID systems have the longest read range and can be used in almost all countries around the world. RFID systems that use 860 to 960 MHz in the UHF range are governed by ISO/IEC specification 18000-6 and EPCglobal Gen1/2. These RFID tags are the least expensive to produce and are used for supply chains, electric toll collection, and asset management. However, they have the drawback that each nation uses different frequencies in the UHF band; for example, America and Canada use 902 to 928 MHz, Europe uses 860 to 868 MHz, Japan uses 950 to 956 MHz, and Korea uses 908.5 to 914 MHz. RFID systems that use 2.45 GHz in the microwave frequency range are governed by ISO/IEC specification 18000-4. Tags of these RFID systems have the fastest read rates and the best susceptibility to the opaque materials. These RFID systems are used for electric toll collection and real-time location of assets. Table 13.1 summarizes the characteristics of RFID systems according to the various frequency bands [1]. 13.1.3 Operating Principles The RFID system can be classified into the active and passive RFID system by the existence or absence of a power source in the tag. An active tag has a transmitter to send signals to the reader and an internal power source to supply power to the tag’s circuitry. Therefore, it can generate strong signals derived from its own battery and can provide great readable range. A passive tag, on the other hand, does not have a dedicated power supply and depends on RF power received from the reader and temporarily stores a small portion of the power to excite a quick response. Moreover, the rectifier is necessary for a passive tag to convert ac power transmitted from the reader into dc power to activate the circuitry on the tag, and the passive tag can respond to the reader within the range which it can obtain the dc power beyond the threshold level. So the readable range of the passive tag is limited. The choice of active and passive tags is practically dependent on their applications because active tags can provide significantly long range of communication but they are more expensive than passive tags. Therefore, active tags can be used in applications that require relatively small numbers of tags with long readable range. Examples include tracking, security, management, and real-time location system. Unlike active tags, supply chain, identification, distribution, and tracking can be included in applications of passive tags.

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Table 13.1 Frequency Characteristics Frequency

125, 135 kHz

13.5 MHz

433.92 MHz

Standards

ISO/IEC 18000 Part 2

ISO/IEC 18000 Part 7

Characteristics

Relatively expensive performance Hardly degraded by environment

ISO/IEC 18000 Part 3, ISO/IEC 15693, ISO/ IEC 14443 Parts A and B Lower cost than tags operating at low frequency Adequate for applications requiring short read range, multiple tag identification

RFID applications

Animal tagging, access control, vehicle immobilizers

Access control, payment ID, item level tagging, biometrics, library books, laundries, apparel, pharmaceuticals Passive Passive Inductive coupling method

Active tags identifying containers, vehicles, other realtime location system applications

Tag Type Operating principles Read range Opaque materials Read rates

0.6m Not susceptible Slow

0.6m Somewhat susceptible Slow

Long read range Environment sensing such as humidity shocking

860 to 960 MHz ISO/IEC 8000 Part 6, EPCglobal Gen-1 and Gen-2 standards Possible to manufacture at the cheapest cost by the development of IC technology Best performance characteristics in multiple identification and read range Supply chain (case and pallet level), asset management, and access control charging

2.45 GHz ISO/IEC 18000 Part 4

Similar performance characteristics with tag in 900 MHz Performance mostly affected by environment

Security, access control, work tracking for factory automation, real-time location system

Active Passive Active/Passive Electromagnetic backscatter coupling method 30m Somewhat susceptible Fast

4m to 5m Very susceptible Fast

10m Very susceptible Very fast

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The operating principles of RFID systems can be divided into inductive coupling configuration and electromagnetic backscatter coupling configuration. RFID systems use these two mechanisms to supply continuous power signal to the tag and to communicate. RFID systems of low frequency or short range systems (< 1m) use an inductive coupling mechanism. This mechanism is shown in Figure 13.3. It can be treated as a simple magnetic altering field, since the wavelength of the frequency is much longer than the distance between the reader and tag. The operating procedure is as follows: Once the tag antenna detects the field of the reader, the tag switches on and off the loaded resistor by its information to change the impedance of the reader antenna coil. Thus, the reader obtains the tag information by variation of the reader antenna voltage. An electromagnetic backscatter coupling configuration is shown in Figure 13.4, and it is used in UHF frequencies, microwave frequencies, and mid to long distance systems (> 1m). The operating procedure is as follows: Once the reader transmits the RF signal,

Figure 13.3 Inductive coupling.

Figure 13.4 Electromagnetic backscatter coupling.

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the tag antenna receives the signal and turns on and off the loaded resistor according to the tag information to vary the reflection characteristics of the antenna. The backscattered signal radiates in free space, and some portion of the signal is detected by the reader antenna. An active tag can be used for cases where the tag is placed very far from the reader, where the backscattering wave from the tag cannot be recognized by the reader. Since an additional battery is housed inside the tag, even though the reader cannot supply enough energy for the tag to operate, it is possible for the tag to be recognized by the reader. Also, the active tag has a ‘‘sleep’’ or ‘‘stand-by’’ mode. Namely, the tag operates only when it detects the signal of the reader to prolong the operating time by saving the power of the battery. 13.1.4 Read Range The most important tag performance characteristic is read range. Read range means the maximum distance that the RFID reader can detect the backscattered signal from the tag. RFID systems have different required read range according to the application. For example, in tagging application, reader and tag operate with a 0.6m range. If the read range of the reader is longer, the reader cannot distinguish tags between users since it will detect many tags from different users at once. Generally, since sensitivity of the reader is typically high in comparison with that of the tag, the read range is defined by the tag response threshold. Read range is also sensitive to the tag orientation, the material of the object to which the tag is attached, and the propagation environment. The read range r can be calculated using Friis freespace formula as follows: r=

␭ 4␲

√

Pt Gt Gr ␥ P th

(13.1)

where ␭ is the wavelength, P t is the power transmitted by the reader, G t is the gain of the transmitting antenna, G r is the gain of the receiving tag antenna, P th is the minimum threshold power necessary to provide sufficient power to the RFID tag chip, and ␥ is the power transmission coefficient between the chip and the antenna given by [2].

␥=

4R c R a

| Zc + Za | 2

,

0≤␥≤1

(13.2)

where Z c = R c + jX c is the chip impedance and Z a = R a + jX a is the antenna impedance. From (13.1) we see that the read range is determined by the EIRP (P t G t ) of the reader, the tag antenna gain G r , and the transmission coefficient ␥ . Thus, the gain of the transmitting and receiving antenna and the impedance matching between the antenna and the

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chip of the RFID are directly connected with the read range. As such, the antenna is the most important element in the RFID system. 13.2 READER ANTENNAS When obtaining the information of the object to which the tag is attached in the RFID system, it is required for the reader to be able to recognize the tag at a long range. In this case, the reader antenna should be designed as a high gain antenna to have long read range. In some cases, it is designed as a beam-forming antenna in order to lend the ability of focusing. However, it is not always desirable for the reader antenna to have a wide and long read range. According to the circumstances of its use, the reader antenna should match its application, particularly because of issues related to infringement of privacy. The polarization of the antenna must also be considered when designing the antenna. A linearly polarized reader antenna can be used in circumstances where the orientation of the tag antenna is fixed, such as in an automated assembly line in a factory. When the orientation of the tag antenna varies randomly, such as at a check-out counter in a supermarket, an antenna with circular polarization is required for communication regardless of the orientation of the tag antenna. The reader antenna does not have a space restriction compared to the tag antenna. However, for a mobile RFID system, a miniaturized reader antenna is required, and the hand effect should be also considered in designing and fabricating the antenna. Finally, a dual band or multiband antenna is used to cover the various frequency bands of RFID applications. 13.2.1 Fixed Reader A fixed reader is applied in situations where objects on which tags are attached pass through the reader and gather the information of the tags, such as highway tollgates, conveyor systems in plants, or traffic card systems. In these cases, the antenna for the reader usually does not have a limitation in size, and must have very high performance and long read range. As previously stated, the operation principle of the RFID system can be categorized as either an inductively coupling configuration or an electromagnetic backscatter coupling configuration. These two mechanisms have different ways of transferring power between the reader and tag, and thus the structure of the antennas is different. Inductive Coupling An RFID system with inductive coupling is usually used in the low frequency band (e.g., below 125 kHz, 13.95 MHz). In these bands, the wavelength is very long and the nearfield region is very large. Hence, using radiation of electromagnetic waves such as dipole

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antenna is impossible. Therefore, in LF and HF band RFID systems, information is interchanged using magnetic coupling between the reader antenna and tag antenna. Generally an N-turn loop antenna is used as a reader antenna and a tag antenna. The antenna’s performance is not determined by the radiation resistance and gain. Instead, how well the reader antenna and tag antenna interact can be determined from the mutual inductance between the reader antenna and the tag antenna. The communication between the N-turn loop reader antenna and tag antenna is shown in Figure 13.5(a). If current I 1 flows into the reader antenna, magnetic flux density is generated in the loop. Only some of the magnetic flux passes to the inside of the tag antenna. At this time, mutual inductance M 12 is generated by the magnetic flux ⌽12 passing through the tag antenna, and it is given by M 12 =

N 12 ⌽12 ␮ 0 N 1 N 2 = I1 4␲

冕冕

dl 1 ⭈ dl 2 R

(13.3)

C1 C2

where N 1 and N 2 are the turn number of the reader antenna and tag antenna, respectively. C 1 and C 2 are the circumferences of the loops. R is the distance between the differential lengths, dl 1 and dl 2 . In (13.3), we see that the mutual inductance M 12 is proportional to the turn number N 1 and N 2 of the reader antenna and tag antenna. Namely, as the turn number of the antenna is increased, the better integration between the reader antenna and tag antenna is possible. Figure 13.5(b) shows the commercial HF band reader antenna.

Figure 13.5 RFID reader antennas with inductive coupling: (a) coupling of two N-turn antennas via a partial magnetic flux, and (b) HF band reader antenna. (Photo courtesy LS Industrial Systems Co.)

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Electromagnetic Backscatter Coupling An RFID system with an electromagnetic backscatter coupling mechanism is more often used at higher frequency than HF band. A planar type antenna, especially microstrip antenna, is generally used for the fixed reader antenna. A microstrip patch has high gain and it can be easily designed to have circular polarization. The theoretically calculated directivity of the antenna is high, about 8 dBi. With simple structure modification, it can have a circular polarization. However, it has a relatively narrow bandwidth (i.e., 1% ∼ 3% fractional bandwidth). The fundamental structure of the microstrip patch antenna is shown in Figure 13.6(a). The height of the substrate is h. There is a patch with width w and length l on the substrate, and there is a ground plane under the substrate. The length of the patch l is about a halfwavelength of the center frequency; hence, by tuning this parameter, the center frequency can be determined. The center frequency of the microstrip patch antenna is given by f=

c 2L e √⑀ reff

(13.4)

where c is the speed of light, ⑀ reff is the effective dielectric constant considering the fringing fields, and L e is the effective length of the patch antenna, and is slightly longer than the physical length l.

Figure 13.6 RFID reader antennas with electromagnetic backscatter coupling: (a) geometry of the antenna; and (b) UHF band reader antenna. (Photo courtesy LS Industrial Systems Co.)

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A practical width w that leads to good radiation efficiencies is given by w=

c 2f

√

2 ⑀r + 1

(13.5)

There are many ways to feed the microstrip antenna. Figure 13.6(a) illustrates the probe-fed microstrip patch antenna. Other feeding methods include microstrip line feed, proximity coupled feed, and aperture coupled feed. The choice of the feeding method is also important in designing the antenna so as to be appropriate to the application, as each has respective advantages and disadvantages. The patch antenna shown in Figure 13.6(a) has a linear polarization. Figure 13.6(b) shows the commercial UHF band reader antenna using a patch antenna with circular polarization. It has separate antennas for transmitting and receiving to achieve the isolation between them. The polarizations of each antenna are right handed circular polarization (RHCP) and left handed circular polarization (LHCP), respectively. Figure 13.7 shows various methods to obtain circular polarization. The four antennas in Figure 13.7 obtain circular polarization via the method of diagonal slot, truncated patch, dual feeding, and stub, respectively. There are also other methods to realize circular polarization [3, 4]. Antennas with different structures, including dipole antenna, monopole antennas, and planar inverted-F antenna (PIFA), can also be used according to the demands of the application and the environment. 13.2.2 Mobile Reader A mobile reader is a device that adds mobility to a fixed reader. The mobile reader can be used for the tags located in the area that cannot be covered by the fixed reader, or it can be used in the field that the end user needs mobility. Some mobile readers, such as handheld reader for industrial RFID applications, PCMCIA reader of card type which can be used by connecting to a PDA or laptop computer, and internal mRFID reader for a mobile communication terminal have been researched and developed. Handheld Reader It is impossible to apply a fixed RFID to cases such as the administration of physical distribution or cargo since the tag-attached objects are too large to move. The handheld reader solves this problem by providing mobility to the reader. The reader and host functions are integrated in one device that is small enough to carry. So, users can get the information of the object anywhere. An example of handheld reader is presented in Figure 13.8. Similar to the fixed reader antennas, handheld reader antennas are usually designed with a planar type antenna so as to have high gain and long read range. However, for the

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Figure 13.7 Circularly polarized (CP) patch antennas.

handheld reader, it is necessary to miniaturize the reader for the sake of mobility, and thus the size of the reader antenna should also be as small as possible. Many methods to reduce the size of the microstrip patch antenna have been reported. First, a substrate with high permittivity can be used. In (13.4), the operating frequency of the antenna is related to the permittivity of the substrate. In this equation, if the antenna length is the same, the operating frequency decreases as the permittivity of the substrate increases. In other words, the required length of the antenna to obtain the same operating frequency reduces as the permittivity of the substrate increases. Hence, the antenna size can be reduced if a material with high permittivity used. The patch antenna size can also be reduced by inserting a slit in the patch. If the slit is inserted in the patch, the current path increases, and consequently the center frequency of the antenna can be lowered. Figure 13.9 shows various miniaturized patch antennas using a slit. Another way to reduce the patch antenna size is to use a shorting wall or shorting pin(s). Figure 13.10(a) shows a field distribution of the microstrip patch antenna. In this

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Figure 13.8 Handheld readers. (Photo courtesy LS Industrial Systems Co.)

Figure 13.9 Miniaturized microstrip patch antenna with slits.

figure, a symmetric field distribution is observed, and radiation occurs at the two radiating slots due to the fringing effect. By applying a shorting wall or shorting pin(s) to the microstrip patch antenna as shown in Figure 13.10(b), the physical size of the patch antenna can be reduced by one-half [4], and the radiation occurs at only one radiating slot, so the radiation resistance decreases. However, the number of edges that affect the radiation of the antenna is reduced by one, and consequently the gain drops. Therefore, in designing the antenna for the handheld reader, it is important to balance the trade-off between size and performance. In Figure 13.10(c, d), shorting pins or one shorting pin can be used. In these cases, the resonant frequency of the antenna is lowered more than the case of using the short wall, and thus more miniaturization of the antenna is possible, while the bandwidth is decreased. PCMCIA Type Reader Another type of handheld reader is the PCMCIA-type reader. This product provides the opportunity to access RFID tags in portable electronic devices, including a PDA or a

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Figure 13.10 Miniaturized microstrip patch antenna with shorting wall/pin(s): (a) field distribution of the microstrip patch antenna; (b) microstrip patch antenna with shorting wall; (c) microstrip patch antenna with shorting pins; and (d) microstrip patch antenna with one shorting pin.

laptop computer, as shown in Figure 13.11. It contains an RFID reader with an external PCMCIA card, and its antenna has the planar shape with a low profile, such as a meander dipole antenna [5]. It is designed considering the miniaturization and the direction of the main beam. mRFID Reader Since the mobile RFID (mRFID) reader is used with a mobile communication terminal, it has the ability to be connected to an Internet through a backbone network of the mobile

Figure 13.11 PCMCIA reader and antenna.

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communication, and to be used in the way that user wants to use it. So to use the mRFID in the handsets, the reader antenna should have dual band or wideband characteristics. Since an mRFID system forms a closed loop system by backscattering from the tag in concept of radar cross-section (RCS) without the system gain in the tag, the antenna with higher efficiency and higher gain than the mobile communication antenna is needed. Recently, since most of the mobile communication terminals use an internal antenna, the mRFID reader antenna can be also realized as an internal antenna. As an internal antenna, inverted-L antenna (ILA), inverted-F antenna (IFA), PIFA, or a miniaturized PIFA are widely used, and they are shown in Figure 13.12. ILA is a structure which a monopole antenna, designed vertical to a ground plane, is bent parallel to a ground plane. The current flows parallel to a ground plane and has a opposite phase to an image current, so it cannot contribute to a radiation in a far-field region. So the radiation resistance is determined by the length of the vertical part of the antenna, and it is smaller than that of the 1/4 wavelength monopole antenna. Also, the impedance matching of the ILA is difficult, since the reactance which is determined by the length of the horizontal part is quite large and capacitive. IFA is the structure to improve the matching characteristics of ILA by adding a short stub next to the feeding point. Generally, it has narrow bandwidth characteristics less than 2% fractional bandwidth. The bandwidth can be increased by inserting an active element to the connecting part of the vertical element and the horizontal element. The bandwidth can be also increased by making the horizontal element as a planar type, and this is the PIFA structure. Miniaturized PIFA changes the horizontal planar structure of the PIFA and has also been investigated. Figure 13.13 shows an example of an internal antenna with high efficiency for operating in the mRFID band and the cellular band [6, 7]. The structure follows varied PIFA divided by four parts, the current distribution according to the number of split parts is shown in Figure 13.14. As the number of elements, N, increases, the current is more uniformly distributed on the radiation component of the antenna. Since the aperture efficiency increases as the amount of current that contributes to radiation increases and the current is widely distributed in the aperture, the antenna efficiency can be increased and the gain proportional to the effective aperture as following (13.6) can be also enhanced.

Figure 13.12 Various internal antenna structures: (a) ILA; (b) IFA; (c) PIFA; and (d) miniaturized PIFA.

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Figure 13.13 Internal antenna in the cell phone operating in mobile RFID and cellular frequency bands.

Figure 13.14 Current distribution by element number: (a) N = 1; (b) N = 2; and (c) N = 4.

G=

4␲

␭

2

Ae =

4␲

␭2

⑀ ap A p

(13.6)

where A e is the effective aperture area, ⑀ ap is the aperture efficiency, A p is the physical size of the antenna, and G is the antenna gain. Since the current flows separately from the feeding structure, which eliminates the leakage currents that lead to degradation of the aperture efficiency and generate the reactance leading to narrow bandwidth, it has an enough bandwidth that can cover the mRFID and cellular band. In the case of integrating the mRFID antenna in the handset terminal, the antenna performance changes as a result of the leakage current of the case and the change of the effective permittivity. Accordingly, designing a method concerning the feeding point of the antenna due to the hand effect, which sensitively affects the antenna performance, has been researched [8].

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13.3 TAG ANTENNAS Tag antennas are different from reader antennas in some characteristics, although both antennas are based on the same system. In this section, the requirements and configurations of the tag antenna will be presented. For the commercial use of an RFID system, it is necessary that the tag antenna can be fabricated at low cost. In fabricating the RFID tag antenna, printing method rather than etching method is usually used because of its simplicity and low cost. However, printing ink for the tag antenna leads to more loss than copper. Low loss conducting ink was recently developed. The impedance of the tag antenna should be conjugate matched to that of the chip in order to transfer maximum power to the chip from the reader. In addition, there is a trade-off between miniaturization and efficiency of the antenna for applications. The antenna must also be designed upon consideration of performance change by the effect of the environment around the tag. In short, when designing a tag antenna, several aspects such as low cost fabrication, impedance matching, miniaturization, and materials around the tag should be considered. 13.3.1 Structure of a Tag Antenna In the previous section, the operating mechanism of the RFID system was classified according to use, either the inductive coupling method or the electromagnetic backscatter coupling method. The tag should operate differently according to these two operating principles, and the structures are also different. The structure of the tag antenna and the operating principles are discussed below. Tag Antennas for Inductive Coupling RFID An RFID system with inductive coupling is usually used in the lower frequency ranges of 125 kHz, 13.95 MHz. Since this system uses low frequencies, a several meters long antenna is needed for efficient operation. Loop antenna is generally used to reduce the antenna size, and N-turn loop antenna is used to increase the radiation resistance [9]. Since this N-turn loop antenna of the tag has the same operating principle as that of the reader, the analyzing method is the same as that of the reader antenna. Tag Antennas for Electromagnetic Backscatter Coupling RFID RFID systems whose operating frequency is higher than HF band operate with electromagnetic backscatter coupling. In this system, a half-wavelength dipole antenna, which is easily fabricated and has a simple structure, is widely used. Since the length of the halfwavelength dipole of the UHF band is about 17 cm, a miniaturized structure is used in

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various applications. One of the widely used structures for miniaturization is the meander structure [10–12]. The meander structure shortens the physical length of the antenna by meandering the straight half-wavelength dipole. Figure 13.15 illustrates the operating principle of the meander antenna by the current distribution. As shown in Figure 13.15, since the currents in the adjacent vertical parts flow in opposite direction, the radiation effects of these vertical currents are cancelled out in the far-field region, and only the horizontal components affect the radiation of the meander antenna. Namely, the radiation resistance decreases and thus the radiation efficiency and the gain also decrease. Because the reactance of the antenna also increases, the antenna bandwidth decreases. A tag antenna with a meander structure has a 5m ∼ 10m read range. A spiral structure has also been employed for tag antennas [13]. As shown in Figure 13.16, the spiral antenna coils a half-wavelength dipole. It has a rectangular shape, and each length of the rectangle is as small as 1/20 wavelength. The current in each plane operates in a similar manner as the small loop antenna, forming one magnetic source. Thus, it has an omni-directional radiation pattern and horizontal polarization. Due to its small radiation resistance, it has more than 2m in read range.

Figure 13.15 Current distribution on a meander antenna.

Figure 13.16 Geometry of a spiral dipole RFID tag antenna.

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A method to miniature the loop antenna for the RFID tag is proposed [14]. In this method, the shunt stub is inserted to the loop antenna to miniature the loop antenna. Another method to miniature the slot ring antenna for the RFID tag by using a fractal structure is proposed [15]. Other studies to miniature the tag antenna have also been reported [16, 17]. These methods offer ease of miniaturization and reduced manufacturing expense by fabricating the antenna on the chip. As presented above, there are various methods to reduce the size of the tag antenna. Additionally, in designing the optimal tag antenna, it is important to consider the wideband characteristics for the variation of the antenna parameters due to the surroundings, and target object, or for worldwide tag. 13.3.2 Impedance Matching One of the most important requirements of a tag antenna is to maximally transmit the received power from the antenna to a tag chip. To satisfy this requirement, the tag antenna should be conjugate matched with the tag chip, which generally has resistance of several to several tens ⍀ and the capacitive reactance of around −100j according to the manufacturing company. Therefore, designing the tag antenna with a good impedance matching is very important in realizing an RFID tag. T-Match T-match is a widely used method for the impedance matching between the tag antenna and tag chip [18–20], and its geometry is shown in Figure 13.17(a). In this figure, a dipole antenna with length l and radius a is connected to a transmission line by another small dipole with length l ′ (l ′ < l ) and radius a ′. The smaller dipole is located at the center of the larger dipole and separated by the distance s [3]. Figure 13.17(b) shows the equivalent circuit of the T-match structure. Z a is the impedance of the large dipole antenna without the T-match. ␣ is the current division factor and is calculated by

Figure 13.17 (a) T-match and (b) its equivalent circuit.

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cosh−1

␣= cosh−1

冉 冉

s 2 − a 2 + a ′2 2s 2 2

2

s + a − a′ 2as

2

冊 冊

≅

ln (s /a ′ ) ln (s /2)

(13.7)

Z t is the impedance at the input terminal for the transmission line mode of the small dipole, and is given as

冉 冊

Z t = jZ 0 tan k

l′ 2

(13.8)

where Z 0 is the characteristic impedance of the two-wire transmission line as shown in Figure 13.17(a), and can be calculated as Z 0 = 60 cosh−1

冉

s 2 − a 2 − a ′2 2aa ′

冊

≅ 276 log10

冉√ 冊 s aa ′

(13.9)

The total input admittance can be written as Yin =

1 Ya 1 = + Z in (1 + ␣ )2 2Z t

(13.10)

The most important characteristic of T-match is that it allows only a portion of the current on the larger dipole flow into the smaller dipole. The radiation power of the antenna is proportional to the multiple of the radiation resistance and the square of the current. Assuming that the input power of the antenna radiates without loss, the input impedance of the antenna becomes larger since a small amount of current flows into the antenna. Also, the small dipole operates as a short-circuited stub, parallel reactance is added to the input impedance. Therefore, the antenna can be easily matched to a tag chip. Figure 13.18 shows the commercial T-match antenna.

Figure 13.18 Antenna with the T-match. (Photo courtesy LS Industrial Systems Co.)

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Inductively Coupled Feeding Inductively coupled feeding is another impedance matching method [21–24]. This method is analogous to the T-match in terms of its operating mechanism. Its geometry and equivalent circuit are presented in Figure 13.19. R body , L body , and C body are the values of the equivalent circuit of the radiating body. M is the mutual inductance between the radiating body and the feed loop, and L loop is the self-inductance of the feed loop. When a sinusoidal current I body flows on the radiating body, the induced current I loop is generated on the feed loop because of the mutual inductance M. Therefore, the impedance of the radiating body is transformed to the input impedance at the balanced input port as shown in the equivalent circuit. In this case, the amount of coupling is determined by the distance d and the size of the feed loop. Moreover, the self-inductance L loop is added to the input impedance by series connection. This additional reactance varies according to the size of the small loop. As such, it is possible to readily realize good matching between an antenna with high impedance and a tag chip with small impedance since the additional reactance can be changed by adjusting the size of the small loop [24]. 13.3.3 Tags on Metallic Surface Tag antennas are generally attached to objects such as commercial goods or boxes. When the tag antenna is attached to metal or dielectric materials, performance variation of the antenna occurs, such as the variation of the radiation pattern, input impedance, and resonant frequency. This could lead to performance degradation of the RFID system. Especially when the tag is attached to the metal, the image current happened by the metal plate causes the destructive interference. In some cases the tag could not be identified. So it is necessary to consider the environment around the tag when designing a tag antenna. Researches on a tag antenna, their characteristics of which are not affected by the attached material, or that can be attached to metallic surfaces, are actively in progress. A patch antenna consists of a proximity-coupled radiating patch and a feeding part [25]. One edge of the radiating patch is shorted to the ground plate to reduce the antenna size. Since the feeding part has a shape of meander shunt stub, it can add parallel inductance at the input port. Generally tag chips have very large capacitance; thus, the parallel inductance makes the antenna easily matched with the tag chip. However, the antenna has a problem that the structure is so complicated because the feeding part is located inside the radiating patch. To solve this problem, a balanced tag antenna is researched, which consists of feed loop and two radiating elements [26]. Because the feed loop is placed at the same layer with radiating elements, it is much easier to mount the tag chip on the antenna. An active tag antenna for the 433-MHz RFID that can be attached to a container is designed [27]. The antenna is miniaturized using slot and slit. Because those antennas require a ground plane, their characteristics do not change even when they are placed on a metallic surface or on a material having different permittivity. Unlike the

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Figure 13.19 Inductive coupled feeding: (a) geometry; and (b) equivalent circuit.

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antennas for a mobile phone, IFA or PIFA for RFID tag must be designed with conjugate matching at chip impedance that is not 50⍀. These IFA or PIFA also have narrow bandwidth and low gain. In the case of IFA or PIFA, a ground plane exists and fabrications of a feeding and shorting pin(s) must be used in manufacturing. This process is more expensive and complex than for general tag antennas of printing type. This also results in higher tag cost. Recently, methods to design the tag antenna as a single layer antenna without a ground plane have been researched [28]. A tag antenna having two radiators which are inner spiral dipole and outer bent dipole in one layer is proposed. These two dipoles yield resonances at adjacent frequencies, so the tag antenna has a broadband characteristic. When the tag antenna is placed on a metallic surface, impedance matching can to be maintained by the inductance at the inner spiral dipole and the capacitance between the tag antenna and the ground. Recently, a tag antenna using a meta-material was proposed [29, 30]. Figure 13.20 shows the operating principle of the tag antenna on an artificial magnetic conductor (AMC) [31]. If an antenna is placed over the AMC, the image current has the same phase with the current of the antenna, so the antenna can work even if it is placed on the conducting plate. 13.3.4 Bandwidth-Enhanced Tag Antennas Miniaturization of the tag antenna is a very important area of study since the tag is expected to be attached to various materials. Generally, an antenna has a narrow bandwidth as it becomes smaller. For example, a meander antenna sometimes has only 1% ∼ 2% fractional bandwidth and this frequency bandwidth may not be wide enough for the desirable RFID system. Since the operating frequency of the tag antenna changes due to the variation of the peripheral effective permittivity according to the attached materials, the tag antenna should cover a wider bandwidth than the required bandwidth of the RFID system.

Figure 13.20 Operating principle of the tag antenna over AMC.

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The bandwidth of the tag antenna can be increased by using double resonances [32]. The antenna consists of two connected meandered dipole, and it is shown in Figure 13.21. Each meandered dipole has resonance at the adjacent frequencies. By using these two resonances, the bandwidth of the tag antenna can be increased. Especially, since this antenna can adjust the double resonance locations on the Smith chart, the tag antenna impedance can be easily matched to the tag chip impedance. A tag antenna structure that has maximum bandwidth using a genetic algorithm is proposed [33]. A wideband tag antenna using a magneto-dielectric substrate instead of a general dielectric substrate is proposed [34]. However, a magneto-dielectric substrate that can be used for a tag antenna is not realized yet. Other than broadband tag antennas, studies on dual band tag antennas are underway [35, 36]. In addition, a reconfigurable antenna using a varactor diode has been used in a tag antenna to overcome the antenna’s performance limitation [37]. 13.3.5 SAW Tags A SAW ID tag system is a wireless identification system in which the radio link between the reader and the tag is furnished by modulated back-scattered waves, as one of many passive UHF band RFID systems. Since SAW devices have a quick response time, it can be used in various fields such as automobile nonstop charging, road sign identification, and the positioning of railway trains. It has many advantages over other identification systems, such as a longer identification distance, the absence of a material link, a lower cost, improvements in terms of rejecting interference signals, and a fast response capability [38]. The radiation efficiency can be decreased by the piezoelectric coupling loss and the dielectric loss of SAW devices. These losses become bigger as SAW devices move closer to the antenna or to the place where the strong current flows. Therefore the location of the SAW device and the antenna structure should be designed considering the antenna efficiency. Since a piezoelectric material generally has high permittivity, it should be designed considering the mismatching due to the variation of the effective permittivity. 13.4 MEASUREMENT OF TAG ANTENNAS The characteristics of reader antenna such as impedance, radiation pattern, and gain can be measured with the typical antenna measurement procedure. However, the measurement

Figure 13.21 Geometry of bandwidth-enhanced tag antenna.

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method of the tag antenna characteristics is different from that of the reader antenna, since it is connected to the tag chip. 13.4.1 Measurement of the Tag Antenna Impedance To establish conjugate matching of the impedance between the tag antenna and tag chip, the exact impedance of the tag antenna should be measured. Generally, the feeding part of the tag antenna that connects to the tag chip is a balanced type structure. In order to measure the impedance using a vector network analyzer that has an unbalanced type probe, a balun is needed for transition. There are many kinds of baluns; among them, the Bazooka balun and Marchand balun are the most widely used. Figure 13.22 shows an example of measuring the tag antenna impedance for a T-match structure using a balun. The exact tag antenna impedance (Z a ) can be calculated from the measured input impedance of the balun (Z in ) through the calibration considering the electric length of the balun. The reflection coefficient between the chip and antenna can then be calculated on the basis of the tag antenna impedance (Z a ) measured in this way and the tag chip impedance (Z C ) as ⌫=

Z C − Z A* ZC + ZA

(13.11)

There is another simple method to measure the tag antenna impedance using an image method [39–41]. However, only antennas with symmetric structure can be measured in this manner. As shown in Figure 13.23, this method treats the symmetric half of the antenna as a ground plane. The antenna impedance can be measured by a network analyzer since the tag antenna with a balanced port is changed to a half antenna with an unbalanced port. At this point, it is necessary to emphasize that the input impedance of the unbalanced port (Z image ) measured using the image method is different from that of the antenna

Figure 13.22 Measurement of the tag antenna impedance using a balun.

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Figure 13.23 Measurement of the antenna impedance by image method.

impedance with a balanced structure (Z A ). Since the radiated power of a half antenna made by the image method is reduced by half, the impedance becomes half that of the original antenna. Hence, the input impedance of the antenna (Z A ) can be calculated by Z image =

1 + ⌫image × Z0 1 − ⌫image

(13.12)

Z A = 2 × Z image where ⌫image is the measured reflection coefficient, Z image is the unbalanced impedance, Z A is the input impedance, and Z 0 is the impedance of a 50⍀ transmission line. Although the image method can only be applied when the two radiating arms of the tag antenna are symmetric and the antenna has a wide ground plane, it is widely used for measuring antenna impedance because it can precisely measure the impedance of the balanced structure without a balun. Figure 13.24 shows an example of measuring the antenna impedance using the image method. 13.4.2 Read Range Measurement The measurement of the read range is the most important aspect of the tag antenna characteristics. The read range measurement of the tag is generally conducted in an anechoic chamber [2]. The measurement method is presented in Figure 13.25. At first, the tag is located in a far-field region, which is the quiet zone of the anechoic chamber that multipaths are minimal. Next, with controlling the attenuator to vary the power of

615

Figure 13.24 Example of measuring the antenna impedance using the image method.

Figure 13.25 Measurement method of the read range in an anechoic chamber.

the reader, the minimum power of the reader that can be detected by the tag is measured. For any transmitter EIRP of interest, the read range is given by r=d

√

EIRP Pmin LG t

(13.13)

where r is the read range, d is the distance between the tag and the reader antenna, P min is the measured minimum output power, L is the cable loss, and G t is the reader antenna gain. 13.4.3 Efficiency Measurement Three ways to measure small antenna efficiency are introduced [42]. Among them, the simplest and best approach is the Wheeler cap method. In this method, the radiation

616

resistance R Rad and the loss resistance R Loss of the antenna are first determined, and then the antenna efficiency ␩ is calculated by

␩=

R Rad R Rad + R Loss

(13.14)

where R Rad + R Loss can be obtained from the antenna impedance when the antenna is in the free space. R Rad is not directly obtained from measurement, but can be obtained by subtracting R Loss from R Rad + R Loss . Since R Loss is the resistance when R Rad is 0, it can be obtained from the measured impedance when the antenna is surrounded by a conducting box or conducting shell so-called Wheeler cap. In this case, the size of the Wheeler cap should be more than ␭ /2␲ in order to reduce the disturbance in the near-field region. It is generally determined to enclose the antenna enough and fabricate it easily. A good contact between the cap and the ground plane is critical for an accurate measurement. REFERENCES [1] Brown, D. E., RFID Implementation, New York: McGraw-Hill, 2006. [2] Rao, K. V. S., P. V. Nikitin, and S. F. Lam, ‘‘Antenna Design for UHF RFID Tags: A Review and a Practical Application,’’ IEEE Transactions on Antennas and Propagation, Vol. 53, No. 8, December 2005, pp. 3870–3876. [3] Balanis, C. A., Antenna Theory, New York: John Wiley & Sons, 2005. [4] Wong, K. L., Compact and Broadband Microstrip Antennas, New York: John Wiley & Sons, 2002. [5] Weigand, S. M., ‘‘Compact Microstrip Antenna with Forward-Directed Radiation Pattern for RFID Reader Card,’’ IEEE Antenna and Propagation Society International Symposium, Vol. 2B, July 2005, pp. 337–340. [6] Lim, Y., et al., ‘‘Antenna Having an Extended Bandwidth of Operation Frequency,’’ Korea, provisional patent application, 10-2006-0047457, filed May 2006. [7] Lim, Y., et al., ‘‘Multi Element Planar Inverted-F Antenna for Enhanced Gain,’’ IEEE Antenna and Propagation Society International Symposium, June 2007, pp. 4689–4692. [8] Lim, Y., et al., ‘‘Feeding Point Determination Regarding Hand Effect of PIFA for Mobile RFID Band,’’ Asia-Pacific Microwave Conference, Vol. 3, December 2006, pp. 1482–1485. [9] Basat, S. S., et al., ‘‘Design and Modeling of Embedded 13.56 MHz RFID Antennas,’’ IEEE Antenna and Propagation Society International Symposium, Vol. 4B, July 2005, pp. 64–67. [10] Michishita, N., and Y. Yamada, ‘‘High Efficiency Achievement by Dielectric Material Loading for a Piled Type Small Meander Line Antenna,’’ IEEE Antenna and Propagation Society International Symposium, Vol. 1B, July 2005, pp. 492–495. [11] Michishita, N., Y. Yamada, and N. Nakakura, ‘‘Miniaturization of a Small Meander Line Antenna by Loading a High ⑀ r Material,’’ 10th Asia-Pacific Conference on Communications and 5th International Symposium on Multi-Dimensional Mobile Communications, Vol. 2, August 2004, pp. 651–654. [12] Nikitin, P. V., S. Lam, and K.V.S. Rao, ‘‘Low Cost Silver Ink RFID Tag Antennas,’’ IEEE Antenna and Propagation Society International Symposium, Vol. 2B, July 2005, pp. 353–356. [13] Chang, K., S. Kwak, and Y. J. Yoon, ‘‘Small-sized Spiral Dipole Antenna for RFID Transponder of UHF Band,’’ Asia-Pacific Microwave Conference, Vol. 4, December 2005, pp. 2687–2690. [14] Ryu, H. K., and J. M. Woo, ‘‘Miniaturisation of Circular Loop Antenna Using Short Stub for RFID System,’’ Electronics Letters, Vol. 42, No. 17, August 2006, pp. 955–956. [15] Padhi, S. K., G. F. Swiegers, and M. E. Bialkowski, ‘‘A Miniaturized Slot Ring Antenna for RFID Applications,’’ Microwaves, Radar and Wireless Communications, Vol. 1, May 2004, pp. 318–321.

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[16] Kim, H., and H. Lee, ‘‘Design of a LTCC Package Antenna for 900MHz RFID Chip Application,’’ IEEE Antenna and Propagation Society International Symposium, July 2006, pp. 4361–4364. [17] Yeoh, W. G., et al., ‘‘A 2.45-GHz RFID Tag with On-Chip Antenna,’’ IEEE Radio Frequency Integrated Circuits Symposium, June 2006, pp. 4–7. [18] Cho, C., H. Choo, and I. Park, ‘‘Broadband RFID Tag Antenna with Quasi-Isotropic Radiation Pattern,’’ Electronics Letters, Vol. 41, No. 20, September 2005, pp. 1091–1092. [19] Schaffrath, M. E., et al., ‘‘Novel Skew Bow-Tie, Skew Patch and Skew Sandglass Tag Antennas for RFID in Paper Reel Logistics,’’ IEEE Antenna and Propagation Society International Symposium, July 2006, pp. 3221–3224. [20] Basat, S. S., et al., ‘‘Design of a Novel High-Efficiency UHF RFID Antenna on Flexible LCP Substrate with High Read-Range Capability,’’ IEEE Antenna and Propagation Society International Symposium, July 2006, pp. 1031–1034. [21] Yang, L., S. S. Basat and M. M. Tentzeris, ‘‘Design and Development of Novel Inductively Coupled RFID Antennas,’’ IEEE Antenna and Propagation Society International Symposium, July 2006, pp. 1035–1038. [22] Choi, W., et al., ‘‘RFID Tag Antenna with a Meandered Dipole and Inductively Coupled Feed,’’ IEEE Antenna and Propagation Society International Symposium, July 2006, pp. 619–622. [23] Cho, C., H. Choo, and I. Park, ‘‘Design of UHF Small Passive Tag Antennas,’’ IEEE Antenna and Propagation Society International Symposium, Vol. 2B, July 2005, pp. 349–352. [24] Son, H. W., and C. S. Pyo, ‘‘Design of RFID Tag Antennas Using an Inductively Coupled Feed,’’ Electronics Letters, Vol. 41, No. 18, September 2005, pp. 994–996. [25] Son, H. W., et al., ‘‘A Low-Cost, Wideband Antenna for Passive RFID Tags Mountable on Metallic Surfaces,’’ IEEE Antenna and Propagation Society International Symposium, July 2006, pp. 1019–1022. [26] Yu, B., et al., ‘‘Balanced RFID Tag Antenna Mountable on Metallic Plates,’’ IEEE Antenna and Propagation Society International Symposium, July 2006, pp. 3237–3240. [27] Cho, W., et al., ‘‘A Planar Inverted-F Antenna (PIFA) to Be Attached to Metal Containers for an Active RFID Tag,’’ IEEE Antenna and Propagation Society International Symposium, Vol. 1B, July 2005, pp. 508–511. [28] Cho, C., H. Choo, and I. Park, ‘‘Design of Novel RFID Tag Antennas for Metallic Objects,’’ IEEE Antenna and Propagation Society International Symposium, July 2006, pp. 3245–3248. [29] Stupf, M., et al., ‘‘Some Novel Design for RFID Antennas and Their Performance Enhancement with Metamaterials,’’ IEEE Antenna and Propagation Society International Symposium, July 2006, pp. 1023– 1026. [30] Feresidis, A. P., et al., ‘‘Artificial Magnetic Conductor Surfaces and Their Application to Low-Profile High-Gain Planar Antennas,’’ IEEE Transactions on Antennas and Propagation, Vol. 53, No. 1, Pt. 1, January 2005, pp. 160–172. [31] Sievenpiper, D., ‘‘High-Impedance Electromagnetic Surfaces,’’ Ph.D. Dissertation, Dept. Elect. Eng., Univ. of California, Los Angeles, CA, 1999. [32] Lee, W., K. Chang, and Y. J. Yoon, ‘‘Small RFID Tag Antenna with Bandwidth-Enhanced Characteristic,’’ IEEE Antenna and Propagation Society International Symposium, July 2006, pp. 1359–1362. [33] Kim, G., and Y. C. Chung, ‘‘Optimization of UHF RFID Tag Antennas Using a Genetic Algorithm,’’ IEEE Antenna and Propagation Society International Symposium, July 2006, pp. 2087–2090. [34] Min, K., T. V. Hong, and D. Kim, ‘‘A Design of a Meander Line Antenna Using Magneto-Dielectric Material for RFID System,’’ Asia-Pacific Microwave Conference, Vol. 4, December 2005, pp. 2701–2704. [35] Diugwu, C. A., and J. C. Batchelor, ‘‘Analysis of the Surface Current Distributions in a Dual Band Planar Antenna for Passive RFID Tag,’’ IEEE Antenna and Propagation Society International Symposium, Vol. 3A, July 2005, pp. 459–462. [36] Jeon, S., et al., ‘‘Dual-band Dipole Antenna for ISO 18000-6/ISO 18000-4 Passive RFID Tag Applications,’’ IEEE Antenna and Propagation Society International Symposium, July 2006, pp. 4285–4288. [37] Hong, W., N. Behdad, and K. Sarabandi, ‘‘Tri-Band Reconfigurable Antenna for RFID Applications,’’ IEEE Antenna and Propagation Society International Symposium, July 2006, pp. 2669–2669.

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[38] Shujian, L., M. Lin, and W. Danzhi, ‘‘A Remote Wireless Identification System Based on Passive Surface Acoustic Wave (SAW) Devices,’’ International Conference on Communications, Circuits and Systems, Vol. 2, May 2005, pp. 27–30. [39] Tikhov, Y., Y. Kim, and Y. Min, ‘‘Compact Low Cost Antenna for Passive RFID Transponder,’’ IEEE Antenna and Propagation Society International Symposium, July 2006, pp. 1015–1018. [40] Tikhov, Y., Y. Kim, and Y. Min, ‘‘A Novel Small Antenna for Passive RFID Transponder,’’ European Microwave Conference, Vol. 1, October 2005. [41] Chang, K., et al., ‘‘A Wireless Identification System Using an Efficient Antenna Based on Passive Surface Acoustic Wave (SAW) Devices,’’ Journal of Korea Electromagnetic Engineering Society, Vol. 7, No. 1, 2007, pp. 12–16. [42] Johnston, R. H., and J. G. McRory, ‘‘An Improved Small Antenna Radiation-Efficiency Measurement Method,’’ IEEE Antennas and Propagation Magazine, Vol. 40, No. 5, October 1999, pp. 40–48.

Chapter 14 Multiple-Input Multiple-Output (MIMO) Systems Santana Burintramart, Nuri Yilmazer, Arijit De, Tapan K. Sarkar, Magdalena Salazar-Palma, Kwok Wa Leung, and Edward Yung

In this chapter, the multiple-input multiple-output (MIMO) system is discussed. MIMO utilizes its space diversity to enhance system performance through increasing the received signal power and/or transmission rate. At the beginning of the chapter, the idea of diversities is introduced and applied to multiple antenna communication systems. Then, the chapter discusses some well-known techniques that theoretically make MIMO systems work along with a performance metric called channel capacity. In the last part of the chapter, MIMO systems are investigated under an electromagnetic viewpoint, as when antennas are deployed in real systems, many assumptions of MIMO systems are not valid; therefore, some performance degradation would be expected. 14.1 INTRODUCTION There has been an explosive growth in the wireless communication area in recent years. This growth has forced the system designers to increase the quality of service, coverage, and bandwidth. MIMO wireless communication has become an active research area for some time since Foschini [1] developed the Bell Laboratories layered space-time system (BLAST). Since then, the potential of using multiple antennas to improve the performance of communication systems, especially the capacity, has become more promising. The MIMO systems offer another domain (i.e., spatial domain) in a system design consideration. With this additional domain, it is possible to increase the system capacity or data transfer rate without increasing the bandwidth of the systems. This improvement in bandwidth 619

620

efficiency of wireless systems is one of the ultimate goals in the Beyond 3G and the 4G era. The MIMO systems also provide a number of advantages over single-antenna systems. Such an advantage includes the reduction of sensitivity for multipath fading through diversity. Research areas in MIMO systems range from information theory, communication system, and signal processing to antenna design. While signal processing and coding are key elements in the development of a MIMO system, antennas and propagations also have a great impact to MIMO system performance. Recently, many researchers have investigated MIMO systems from an electromagnetic perspective [2–4]. It will be shown in this chapter that the electromagnetic effect can deteriorate the performance of MIMO systems. Without considering this effect in the system design, one might not reach the best performance or may even get worse. In this section, we will start from the concept of diversity in general, and then this concept will be applied to the MIMO wireless systems. Some algorithms will be provided to give the reader insight on how the MIMO system works. Then, we will discuss the performance metric, known as channel capacity, for the MIMO systems along with assumptions on the channel knowledge and their impacts on the performance. A tradeoff between data rate gained and an increase in the signal-to-noise ratio (SNR) will be briefly reviewed. Then the discussion will move toward electromagnetic perspective on the MIMO systems based on numerical simulations, followed by the conclusion. 14.2 DIVERSITY IN WIRELESS COMMUNICATIONS As wireless channels experience multipath fading, a signal transmitted over these channels is usually distorted, resulting in the reduction of overall transmission rate since the signal needs to be retransmitted. However, due to the fact that independent signal paths are unlikely to have deep fades simultaneously, it is possible to transmit the same signal over these paths to mitigate the effect of multipath fading. This is the idea behind diversity. In general, diversity can be obtained over time, frequency, and space. 14.2.1 Time Diversity When the wireless channel is time-varying, time diversity can be utilized. This is generally the case for wireless communications since the transmitter and/or the receiver is moving. The information symbols, bearing the same information, are repeatedly transmitted over a period of time so that the symbols experience independent fading. When combining the received symbols together, deep fades will be averaged out over time. This can be achieved if two repeated symbols are transmitted farther apart in time than a coherent time, Tcoh , which is the amount of time over which the propagation remains correlated. This technique is called interleaving. Assuming a signal symbol, s (t ), is transmitted repeatedly L times.

621

The low-pass received signal corresponding to the l th transmission can be represented as follows: r l (t ) =

√E S ␣ l e

−j␾ l

s l (t ) + n l (t ),

l = 1, . . . , L

(14.1)

where ␣ l e −j␾ l represents the random complex gain for the l th transmission, and n l (t ) denotes the additive white Gaussian noise associated with the l th transmission. E s is the transmitted signal power. At the receiver, a combiner combines these L signals together so that the received signal-to-noise power ratio (SNR) per bit, ␥ b , increases compared to when only one symbol is transmitted. This improvement in the received SNR is referred to as a diversity gain. The optimum combiner, maximal ratio combiner (MRC), can be achieved when the complex gain ␣ l e −j␾ l of each transmission is perfectly known [5]. The SNR per bit for this combiner is given by

␥b =

Es N0

L

∑ ␣ l2

(14.2)

l =1

where 冠E s ␣ l /N 0 冡 is the received SNR for the l th transmission. The effects of errors in the estimates of ␣ i and ␾ i for this technique are illustrated in [6, Append. C]. Even though the output SNR can be improved by the repetition of L symbols in time, this obviously reduces the overall transmission rate of the system. Similar diversity can be employed in other dimensions (frequency and space) to improve the rate of transmission as well. 2

14.2.2 Frequency Diversity For a frequency-selective fading channel, where its response is not constant over the bandwidth of interests, the constant complex gain for each channel as in (14.1) cannot be applied. This happens when the transmission bandwidth is much larger than the channel coherent bandwidth, B coh , measuring the frequency bandwidth over which the propagation channel remains correlated. For single-carrier systems, using equalization techniques can mitigate this effect. It can be done in either time or in frequency domain. For multicarrier systems, we subdivide the symbol into small frequency bands such that the frequencyselective fading channel is converted to narrowband flat fading channels. Then, the coded symbols are transmitted through multiple subcarriers, each of which experiences narrowband flat fading response. When the subcarriers are orthogonal to each other, this technique is called orthogonal frequency division multiplexing (OFDM). In the OFDM, the information symbol is first encoded to the coefficients of the discrete Fourier transform (DFT). We consider a burst of encoded N symbols to be transmitted at time t, X t = [X t (1) X t (2) X t (N )]. The transmitter performs the inverse discrete Fourier transform to get the transmitted signal x˜ t as follows:

622

x˜ (t ) =

1 √N

N

∑

k =1

X k e j2␲ kt/N,

0≤t≤T

(14.3)

where T is the transmitted signal interval. To avoid the interburst interference, the cyclic prefix can be appended to the burst of N symbols with a cost of reduction in transmission rate [7]. On the receive channel, the receiver performs the DFT of the received signal to extract the original transmitted symbols. Since all of the information contains in the orthogonal frequencies, the OFDM is a good candidate for frequency-selective fading channels. 14.2.3 Space Diversity Similar to time and frequency diversities, the performance of a wireless system can be enhanced over the spatial dimension as long as the separation between the two transmitted signals is farther apart to ensure that the transmitted signals experience different fading. The distance that the two signals have uncorrelated fading statistic is called coherent distance, S coh [8]. By using space diversity, we can improve received signal quality or even increase transmission rate without changing the bandwidth of the systems. However, the price we pay is at the increased costs of the multiple transmitters and receivers deployed in the systems. The space diversity can be categorized into diversity at the receiver and at the transmitter. At the receiver, many combining techniques can be used to obtain the diversity gain. One of the simplest ways to achieve space diversity is that the receiver with multiple antennas selects the received signal from the antenna that has the best reception according to some criteria such as received signal power, SNR, and so forth. This technique is called selection combining. Another combining technique that is similar to the beam-forming concept is called gain combining, where the combiner weighted sums the received signal from each antenna to improve the signal quality. Examples of techniques based on the gain combiner are equal gain combining (EGC) and maximal ratio combining (MRC). Some of these techniques will be discussed in the following sections. Diversity techniques can be also implemented at the transmitters to prevent an increase in cost at the receivers due to multiple receiving antennas. This is more practicable for mobile phone systems when the mobile terminals are designed to keep their prices as low as possible. However, the diversity at the transmitter usually requires knowledge of the channels. In a time division duplex (TDD) system, channel information can be fed back to the transmitter so that it can generate a weighted average to enhance the transmitted signal quality. For other systems, some kinds of feedback from the receiver to the transmitter need to be implemented; otherwise, the diversity cannot be applied at the transmitter side. As seen so far, when talking about the diversity, in fact, we are actually trying to send signals through multiple independent paths: time, frequency, or space. The effect of

623

multipath fading can be mitigated due to the independent nature of the paths. These paths can be combined to further enhance signal quality or transmission rate. MIMO-OFDM is one such technique where both space and frequency are used to increase the diversity. It is important to note that those paths need to be statistically independent or uncorrelated in order to utilize this diversity. Many coding schemes known as space-time coding have been proposed in the literature to enhance the independency of various channels. Some of these coding schemes will be discussed in the next sections.

14.3 MULTIANTENNA SYSTEMS We have seen previously that the combining techniques can be utilized in both time and spatial domains. For multiantenna systems, we are mainly concerned with the space diversity rather than the others. This diversity can be implemented on either the transmitter or the receiver side, leading to multiple-input single-output (MISO) or single-input multiple-output (SIMO) systems depending on where the diversity is applied. We first start with the SIMO, which in principle has the same advantage as in a MISO case except that for the MISO case the channel knowledge at the transmitter is more difficult to obtain than at the receiver. Later on we will extend this idea to MIMO systems. Recall the gain combining technique mentioned above, the system model at the receiver for the M-antenna systems is shown in Figure 14.1. The weighted and summed output, y, entering the detector is given by

Figure 14.1 Gain combining diversity for M-antenna receiver.

624

M

y=

M

∑ h i r i = ∑ √E S h i ␣ i e −j␾ s + n˜ i i

i =1

(14.4)

i =1

where ␣ i e −j␾ i is the complex channel gain assumed to have its distribution as a zeromean complex Gaussian random variable with unit variance at the i th receiver chain. h i denotes a complex weight for the i th receiver. s is the transmitted signal from the transmitter and n˜ i = h i n i is the zero-mean additive white Gaussian noise with variance N 0 at the receiver. The weighting coefficient h i can be chosen in several ways. In the EGC technique, these weights are chosen such that the received signals are cophased with each other or h i = e j␾ i. The average output SNR of this combiner for Rayleigh fading is as follows [5]:

␥ EGC =

冉

␲ Es 1 + (M − 1) N0 4

冊

(14.5)

The MRC mentioned earlier can also be applied when the weighting coefficients are chosen such that h i = ␣ i e j␾ i. The average output SNR of the MRC is given by

␥ MRC =

Es M N0

(14.6)

However, it is noted here again that the receiver needs to have knowledge about the channel gains to obtain this optimum output SNR. These combining techniques can be implemented at the transmitter in a similar fashion as long as the channel gains are known at the transmitter; otherwise, other techniques need to be implemented. Delaydiversity and phase-sweeping schemes [7] are examples of these possible techniques that the transmitter tries to simulate multipath fading environment to increase its diversity. 14.4 MIMO SYSTEMS Wireless systems with multiple antennas at the transmitter and receiver are considered as MIMO systems. The multiple antennas at the both ends of the systems provide independent paths in the multipath fading environment. As seen with multiple antennas, the signals can be enhanced through diversity gain. Thus, it is possible to utilize this diversity gain in MIMO systems as well. In MIMO systems, another achievable gain that improves a system performance is called the multiplexing gain—an increase in the transmission rate. The multiplexing gain comes from the fact that a MIMO channel can be decomposed into a number of independent channels. By having independent channels, multiple symbols can be transmitted simultaneously, which results in increasing the data transfer rate in comparison to a single antenna system.

625

A simple input-output relationship for a narrowband flat fading MIMO system with N T transmit and N R receive antennas is given by y = Hx + n wherein

y = 冋 y 1 , y 2 , . . . , y N R 册T

is

the

vector

(14.7) of

received

signals

and

x = 冋x 1 , x 2 , . . . , x N T 册 is the vector of transmitted signals, n = 冋n 1 , n 2 , . . . , n N R 册T is the noise vector where its components are assumed to be additive white Gaussian noise, and H denotes the N R × N T channel matrix, whose h ij component represents the channel gain from the j th transmit antenna to the i th receive antenna. Figure 14.2 shows the MIMO system model. With the knowledge of H, the MIMO channel can be decomposed into K parallel channels, where K ≤ min (N R , N T ). A common approach in decomposing the MIMO channel is via the singular value decomposition (SVD) [9]. The SVD of the channel matrix H is given by T

H = U⌺V H

(14.8)

where U = 冋 u 1 , u 2 , . . . , u N R 册 ∈ C N R × N R, V = 冋 v 1 , v 2 , . . . , v N T 册 ∈ C N T × N T, and ⌺ is a diagonal matrix of singular values, ␴ i , of H. { ⭈ }H denotes a complex conjugate transpose. By using V and U in encoding the input signal x and decoding the output signal y, it is equivalent to transforming a transmission through MIMO channels into multiple transmissions using parallel single-input single-output (SISO) channels. This is shown mathematically as follows

Figure 14.2 MIMO system model.

626

y˜ = U H y = U H (U⌺V H )x + U H n y˜ = ⌺V H x + U H n y˜ = ⌺V H x + n˜ where y˜ is the output of the decoded systems and n˜ = U H n is the transformed noise. Note that since U is a unitary matrix, U H U = I, an identity matrix, and U H n does not change the distribution of the noise. Thus, n˜ and n have the same distribution. However, V H x represents a linear transformation of the input signal x, which is not designed in practice. We, therefore, further encode the input signal as x˜ = Vx, so that the input signal x will be transmitted through the uncoupled channels. This is sometimes referred to as MIMO beam-forming in the literature, which should not be confused with the context of beamforming in antenna perspective, in which beam patterns are concerned. The overall channel decomposition is shown in the following formulation and is intuitively explained in Figure 14.3. y˜ = U H y = (U H U) ⌺V H x˜ + U H n y˜ = U H y = (U H U) ⌺ (V H V) x + U H n

(14.9)

y˜ = ⌺x + n˜ Based on this decomposition, multiple signals can be transmitted simultaneously, which increases the data rate of the system without changing the bandwidth of the systems. However, the performance of the transmission strictly depends on the singular values of the channel matrix since they represent gains of the decomposed channels. It is also possible that the matrix H is not full rank. In that case, the decomposed channel will be useful for transmission of data only by the mode with singular values larger than a given

Figure 14.3 Parallel decomposition of MIMO channel.

627

threshold. This results in a multiplexing gain of K, where K ≤ min (N R , N T ). When H is full rank, K = min (N R , N T ), the channel is called a rich scattering environment. This is where we gain benefits from multiple antenna systems. Different assumptions are made about the knowledge of the channel matrix. These assumptions lead to different schemes for MIMO systems. In the next section, we will first discuss channel capacity followed by an assumption about knowledge of the channel and its impact to the MIMO systems. 14.5 CHANNEL CAPACITY OF THE MIMO SYSTEMS In this section, fundamental limit on the spectral efficiency in MIMO systems is discussed. The quantity that represents the maximum error-free transmission rate is called the channel capacity. The channel capacity for additive white Gaussian noise channels was first introduced by Claude Shannon [10, 11]. This quantity represents the transmission rate per hertz of bandwidth (bits/s/Hz) of information. To understand how multiple antenna systems can increase the capacity, we first consider the capacity of a SISO system. We will assume the channel to be a narrowband flat fading model unless otherwise stated. From (14.1) we repeat here that the received signal for a single transmission can be given by r (t ) =

√E S ␣ e

−j␾

s (t ) + n (t )

(14.10)

If E S is the transmitted signal power and N 0 is the noise power, the channel capacity of this SISO system can be written as

冉

C 1 = log2 1 +

冊

ES ⭈ ␣2 , N0

bits/s/Hz

(14.11)

For MIMO systems with N R receive and N T transmit antennas in (14.7), the capacity is defined as follows [12]: C=

max Tr (R ss ) = N T

|

log2 I N R +

ES HR xx H H NT N0

|

(14.12)

wherein I N R is an identity matrix of dimension N R × N R , R ss = E {xx H } is the covariance matrix of the transmitted signal x. E {⭈} is the expected value operator. |⭈| and Tr (⭈) denote the determinant and the trace of a matrix, respectively. Tr (R ss ) = N T is the total average power constraint at the transmitter. This capacity represents the information rate per unit bandwidth that can be transmitted with negligible error over the MIMO channel. Clearly, to achieve the maximum MIMO channel capacity, we need to optimize R xx that is the correlation of the transmitted signals.

628

Next we consider the situation that the channel capacity will be maximized. We first assume that the number of transmit and receive antennas are equal, N T = N R = M. When the transmitted signal is uncorrelated, R xx = I M , the channel capacity reduces to

|

C = log2 I M +

ES HH H MN 0

|

(14.13)

Assuming the channel matrix is orthogonal and ␣ 11 = ␣ 22 = . . . = ␣ MM , this results in the capacity of

冉

2

C = M log2 1 +

E S ␣ 11 N0 M

冊

(14.14)

where ␣ ii denotes the channel gain from the i th transmit antenna to the i th receive antenna. Since the multiplication by the factor of M outside the logarithm has more offset than the division by M inside the logarithm, this is equivalent to increasing the channel capacity by a factor of M when compared with a single channel transmission in (14.11). In other words, when the channel matrix is orthogonal and the transmitted signals are uncorrelated, transmitting information through M × M MIMO channels is equivalent to transmitting signals via completely uncoupled M parallel transmissions. Note that all other factors need to be the same for all the parallel transmissions in this case. Later it will be shown that by using the parallel decomposition of MIMO channels, an increase in channel capacity can be obtained. 14.6 CHANNEL KNOWN AT THE TRANSMITTER In practice, the channel information at the receiver can be easily obtained when compared to the transmitter since to have channel information at the transmitter one requires some kind of feedback to send the information back to the transmitter. Thus, we will consider only the assumptions related to the knowledge of the channel at the transmitter side throughout this chapter. At the receiver side, the knowledge of the channel is assumed. When the transmitter knows the channel information, H, (referred to as a closed-loop system) either by feedback from the receiver or through estimation using reciprocity in the TDD systems [12, 13], the MIMO channel decomposition can be utilized to increase the capacity. Assuming that the transmitted signals are uncorrelated (i.e., R xx = I N T ) and given the eigen decomposition HH H = Q⌳Q H, the MIMO channel capacity in (14.12) can be written as

|

C = log2 I N R +

ES Q⌳Q H N0 NT

|

(14.15)

629

K

C=

∑ log2

i =1

冉

1+

ES ␭ N0 NT i

冊

(14.16)

where ␭ i is the eigenvalue of HH H and K is the rank of the channel matrix H. This channel capacity can be interpreted as the sum of capacities of K channels, each having the gain of ␭ i . Since the value of ␭ i is not the same for all the channels as in the ideal orthogonal channels, we need to allocate the power for each channel properly to maximize the capacity. 14.6.1 Water-Filling Algorithm Since the power gain for each channel is not equal, the best way to transmit data efficiently is to spread the transmitted power such that each channel contains similar energy. This leads to a well-known solution for power allocation as water-filling solution. The algorithm of this power allocation is given as follows [12, 14]: 1. Select the number of parallel channels: m = K. m 1 P 1 , where ␭ i is the eigenvalue in the 2. Determine the constant: ␮ = T + m m ∑ ␭ =1 i i

descending order (␭ i ≥ ␭ j , for i < j ) and P T denotes total transmitted power available. 1 3. Determine the power allocated to the i th subchannel: p i = ␮ − , for ␭i i = 1, . . . , m. 4. If p m > 0, then we set E S i = p i for i = 1, . . . , m ; otherwise, we set E Sm = 0 and m = m − 1. Then we rerun the algorithm in step 2.

The water-filling algorithm will allocate the transmitted power according to the subchannel gain shown by the eigenvalue ␭ i . Whenever the subchannel gain is low (high attenuation), the algorithm will discard the use of that subchannel, which in turn reduces the number of parallel channels available in the MIMO system. 14.7 CHANNEL UNKNOWN AT THE TRANSMITTER So far, we have discussed the use of MIMO systems and their benefits, specifically, an increase in the channel capacity over the SISO systems when the transmitters have knowledge about the channels. In the absence of this information, it is not possible to allocate transmitted power efficiently over the transmit antennas. The best strategy is to transmit equal power for each transmit antenna. If the transmitted signals are chosen such that they are uncorrelated, the channel capacity is given in (14.16). As the capacity is a

630

function of the eigenvalues, which are not known to the transmitter in this case, the multiplexing gain may be reduced due to the fact that some eigenvalues may be too small to convey the transmitted information. However, one could utilize the diversity gain for the MIMO systems as well through space-time coding [14, Chap. 5]. Next we will show an example of a coding scheme that provides a diversity gain even when there is no information about the channels at the transmitter.

14.7.1 Alamouti Scheme With an appropriate coding scheme, the transmit diversity can be obtained even in the absence of channel information. Alamouti [15] has proposed a simple coding scheme that can provide a diversity order of 2N R for the system of two transmit antennas and N R receive antennas. Let us consider a MIMO system with 2 × 2 antennas and assume that the transmitter has no channel information. The channels are assumed to be frequency flat fading and assumed to remain constant over the transmissions of two symbols. The channel matrix, H, is given by

H=

冋

h 11

h 12

h 21

h 22

册

(14.17)

In the first symbol period, two different symbols, s 1 and s 2 , are transmitted simultaneously from antenna 1 and antenna 2, respectively, with the energy per bit per transmission is E s /2. At the second symbol period, however, the transmitter sends −s *2 from antenna 1 and s *1 from antenna 2 instead, where s* denotes complex conjugate of s. The received signals at the receive antennas over two consecutive symbol periods, y 1 and y 2 , can be expressed as

√ 冋 √ 冋

y1 =

y2 =

ES 2

ES 2

册冋 册 冋 册 册冋 册 冋 册

h 11

h 12

s1

h 21

h 22

s2

h 11

h 12

h 21

h 22

−s *2 s *1

+

+

n1

n2

n3

n4

(14.18)

(14.19)

where n 1 , n 2 , n 3 , and n 4 are the AWGN samples with zero-mean and power N 0 . At the T receiver, the received signals are put together to form a signal vector y = 冋 y 1 y *2 册 . The signal vector y can be expressed as

631

y=

y=

冤 冥 冤冥 h 11

h 12

h 21 h *12

h 22 −h *11

h *22

−h *21

√

ES 2

√

ES H s+n 2 A

冋册 s1

s2

n1

+

n2 n *3

(14.20)

n *4

T T where s = 冋s 1 s 2册 and n = 冋n 1 n 2 n *3 n *4册 . According to the structure of H A , it H 2 2 2 2 implies that H A is orthogonal or H A H A = 冠 | h 11 | + | h 12 | + | h 21 | + | h 22 | 冡 I 2 × 2 is a H diagonal matrix. Thus, if the receiver performs z = H A y, we will get

z=

√

ES 2

冠 | h 11 | 2 + | h 12 | 2 + | h 21 | 2 + | h 22 | 2 冡 I 2 × 2 s + n˜

(14.21)

H

where n˜ = H A n is a complex Gaussian noise vector with zero mean and covariance 2 2 2 2 E {n˜n˜ H } = 冠 | h 11 | + | h 12 | + | h 21 | + | h 22 | 冡 N 0 I 2 × 2 . We see that the receiver decouples the two transmitted symbols, each of which is contained in the components of z, zi =

√

ES 2

冠 | h 11 | 2 + | h 12 | 2 + | h 21 | 2 + | h 22 | 2 冡 s i + n˜ i ,

i = 1, 2

(14.22)

This gives the SNR for each received symbol as follows:

␩=

冠 | h 11 | 2 + | h 12 | 2 + | h 21 | 2 + | h 22 | 2 冡 E s 2N 0

(14.23)

It yields an increase in the received SNR according to the summation of the channel gains. When the channel is considered as a spatially white channel (i.e., E { | h ij | 2 } = 1), the Alamouti scheme yields the maximum diversity gain, which is equal to four for this 2 × 2 MIMO system. Note that the transmission rate does not get improved by using this scheme, as the transmission of two symbols requires two symbol periods. In other word, there is no multiplexing gain, but there is diversity gain for this MIMO system. 14.8 DIVERSITY-MULTIPLEXING TRADE-OFF In the previous section, we see that even though the multiplexing gain cannot be obtained due to the absence of any knowledge of the channel, but with a proper transmission

632

scheme, the diversity gain can be achieved. Zheng and Tse have shown in [16] that it is not possible to achieve the maximum of both diversity and multiplexing gains at the same time. We briefly review this fundamental trade-off in this section. According to [16], a scheme is said to achieve the multiplexing gain r if R (SNR ) =r log (SNR )

(14.24)

log Pe (SNR ) = −d log (SNR )

(14.25)

lim SNR → ∞

and achieve the diversity gain d if lim SNR → ∞

where Pe (SNR ) is the probability of error at a given SNR and R (SNR ) is the rate of a code scheme as a function of SNR. For a given scheme with block length of l ≥ N T + N R − 1, the optimal trade-off between the diversity and the multiplexing gains that any scheme can reach in the case of Rayleigh-fading MIMO channel can be given by [16] d (r ) = (N T − r ) (N R − r ),

r = 0, 1, . . . , min (N T , N R )

(14.26)

where d (r ) represents the trade-off curve which is a piecewise-linear function connecting the points (r, d (r )). The trade-off curves for the case of N T = N R = 2 and N T = N R = 3 are shown in Figure 14.4. It is seen that the Alamouti scheme for the 2 × 2 system considered in the previous section has the maximum diversity while there is no increasing in the transmission rate. However, since this fundamental limit is derived from the channel capacity, constrained for an arbitrarily low bit error rate, when this constraint is relaxed, it is possible to have a full-diversity and full-rate transmission [14, 17]. 14.9 MIMO UNDER AN ELECTROMAGNETIC VIEWPOINT So far we have discussed the idea behind MIMO systems, their algorithms, and their performance from a statistical point of view. However, real world physics dictate these concepts be viewed under a Maxwellian framework which is vector in nature. Secondly, all the statistical analyses have been done using point source antenna theory, which really does not exist in practice. There are additional hidden subtleties which are missed in the point source model. For example, the entire field radiated by a point source is akin to far-field electromagnetic radiation and which is real and the power can be computed from either the electric or the magnetic field. But for a finite length dipole there are both near and far fields. In the near field this simple approximation done in the signal processing literature does not hold. Moreover, there is mutual coupling between the antennas, which

633

Figure 14.4 Diversity-multiplexing trade-off for 2 × 2 and 3 × 3 MIMO systems.

can completely alter the nature of the conclusions if their effects are taken into account. Rigorous work has been done and is well established under a scalar signal processing, communication, and information theory framework. However, research topics in MIMO systems under an electromagnetic point of view have been recently studied [2–4]. As we know, signal processing algorithms—array processing to be more specific-were originally developed for sonar applications. Later on they were adapted for wireless communications where antennas are parts of the systems. Unfortunately, these two arenas are disjoint. While sonar signal is a scalar, the electromagnetic signal is a vector [18, 19]. Thus, great care needs to be taken into account in system and/or algorithm development to obtain the best performance. Also, for an antenna the space and time variables are not independent, rather they are related, as an antenna is both a temporal filter and also a spatial filter, and that is why one uses Maxwell’s equations to study antennas. These equations capture the fundamental real life physics, which is missing in a scalar statistical analysis. In this section, we illustrate the vector nature of the MIMO electromagnetic system through a few simulated numerical examples. The goal of this section is to show that without considering the electromagnetic nature of wireless communications, performance degradation will occur. Due to the existence of complicated multipath environments, the discussion

634

in this section will be based only on numerical simulations, without mathematical proof, through an accurate electromagnetic analysis [20]. 14.9.1 Case Study 1 To get an intuitive idea, we first look at a simple scenario when the SIMO system of N T = 1 and N R = 3 is considered. The receiver is the MRC combiner mentioned earlier in Section 14.3. Note that it is shown in (14.6) that the diversity gain of the MRC is equal to three in this case (three receive antennas), which results in three times higher output SNR compared to a single-antenna system. Let us consider all the antennas as ␭ /2 (halfwavelength long) dipoles operating at f = 1 GHz with a radius of ␭ /300 (1 mm). All the antennas are centrally loaded with the complex impedance Z L = 90.57 − j 42.51 [⍀], so that they are operating in the resonant mode, which implies that they are going to radiate maximum power in free space. For the SISO system, two antennas are 100m apart in free space, with 1W of power transmitted, the received current is I SISO = 0.017 + j 0.023 [mA]. This gives a received power of P SISO = 78.0 [nW]. When three identical antennas are used at the receiver, the receive antennas are 5m apart from each other. Note that all the antennas are vertically polarized along the z-direction. The simulation setup for this SISO and SIMO case is shown in Figure 14.5. The currents induced at the receive antennas are I 1 = −0.007 − j 0.028 [mA], I 2 = 0.020 + j 0.021 [mA], and I 3 = −0.007 − j 0.028 [mA], Note that I 1 = I 3 due to the symmetry of the structure. This yields the received power, or P 1 = 75.6, P 2 = 76.2, and

Figure 14.5 SISO and SIMO setups for case 1 (all antennas are vertically polarized): (a) SISO system; and (b) SIMO system.

635

P 3 = 75.6 [nW]. It means that under this ideal case (no noise and losses) 3

P MRC =

∑ P i = 227.4 [nW] or 2.92P SISO . This loss in the diversity gain is due to the

i =1

fact that there is mutual coupling between antennas and the radiated power is a function of 1/R 2, where R is the distance between a transmit and a receive antenna [21]. Noting that the distances from the transmitter to the antenna 1 and 3 in our case is longer than the one to the antenna 2. 14.9.2 Case Study 2 In the second case, let us make the situation more complicated by introducing two objects in the scenario. This causes multipath interference at the receiver. The antennas used in this simulation are exactly the same as in the previous case. Note that all the antennas are vertically polarized along the z-direction. The simulation setup is as follows: 1. The transmit antenna is at the origin. 2. Three receivers are place at the (x, y, z) coordinate as (5, 100, 0), (0, 100, 0), and (−5, 100, 0), respectively, where the coordinate is in meters. 3. A metallic sphere with diameter of ␭ or 0.3m is at (−5, 60, 0). 4. A metallic cube with the dimension of 0.3m × 0.3m × 0.3m is placed at (10, 75, 0). Figure 14.6 shows the setup for this simulation. The received current for the SISO case, when using only Rx2 as a receiver, is I SISO = 0.020 + j 0.021 [mA] or the received power is P SISO = 79.7 [nW]. The currents induced at the receive antennas and their received power for the SIMO case are related to I 1 = −0.006 − j 0.028 [mA], I 2 = −0.021 + j 0.021 [mA], I 3 = −0.006 − j 0.027 [mA], and therefore, P 1 = 73.6 [nW], P 2 = 78.3 [nW], and P 3 = 73.7 [nW], respectively. This is similar as before, even though there is multipath fading, the received power from the MRC is P MRC = 225.6 [nW], which is 2.83P SISO in this case. 14.9.3 Case Study 3 In this simulation, we investigate the multiplexing gain of a MIMO system along with its tolerance to the changing environment. We, in fact, simulate the multipath fading with some degrees of randomness. We use a MIMO system of N T = N R = 2 with the same antenna configuration as before and we use the method of parallel decomposition as our transmit scheme. The two transmitters are separated by a wavelength (␭ ) and the two receivers are also placed ␭ apart. They are located inside a room of dimension 2m × 2m. The room has a metallic wall of height 0.5m. There is no ceiling or floor for the room

636

Figure 14.6 SISO and SIMO setups for case 2.

considered in this simulation. Inside the room, there is one metallic sphere with diameter of 0.4m and one metallic cube with dimension of 0.4m × 0.4m × 0.4m. If we let the room be centered at the origin, where the wall starts from z = 0m, as shown in Figure 14.7, the sphere is centered at (−0.2, 0.3, 0.25). The cube is centered at (0.4, −0.1, 0.25). Two transmit antennas are centered at (0.15, −0.75, 0.175) and (−0.15, −0.75, 0.175). Similarly, the receive antennas are centered at (0.15, 0.75, 0.175) and (−0.15, 0.75, 0.175), respectively. In order to create random multipath fading, we make variations of the locations for the sphere and the cube. However, to keep the fading statistic stationary, the variations of the locations of the two objects are considered as zero-mean Gaussian random variable with a standard deviation ␴ x = ␴ y = ␴ z = ␴ , where ␴ x denotes the standard deviation of the change in the location along x direction, similarly for y and z directions. Before introducing the randomness, we measure the channel matrix H (when the sphere and cube are centered at the right positions) and calculate its SVD. The transmit symbol is x = [x 1 x 2 ]T where x 1 = 1 and x 2 = j. These transmitted symbols are assumed to be on the QPSK constellation mapping. By using the SVD, our encoded symbols are x˜ = Vx as explained in Section 14.4. On the receive part, the receiver decodes the received signal as follows y˜ = U H y. We expect that by parallel decomposition the transmitted symbols x 1 and x 2 will be attenuated proportional to the singular values of H. We note here that the true V and U will be used throughout the simulation, even though there is a variation in the channel matrix H, due to the change in object positions. This is practically true since after the channel matrix is estimated either at the receiver side or also at the transmitter side, the estimated value will be used for a while as long as the channel is considered stationary, which is not necessary to be constant. From our simulation, the exact channel matrix is given by

637

Figure 14.7 (2 × 2) MIMO simulation for case 3.

H=

冋

0.022 + j 0.132

−0.057 + j 0.042

−0.195 − j 0.061

0.245 + j 0.587

册

⭈ 10−3

The SVD of H is as follows: U= ⌺= V=

冋 冋 冋

册

0.017 − j 0.042

0.131 + j 0.990

−0.948 − j 0.315

−0.045 + j 0.009

0.668

0

0

0.148

册

⭈ 10−3

0.298

0.955

−0.627 + j 0.720

0.196 − j 0.224

册

When there is no variation in the scenario, the decoded symbol 0.668 y˜ = ⭈ 10−3 is exactly what is expected to be. Note that since the phases of the j 0.148 two received symbols are correct, the messages can be conveyed successfully. It means that the multiplexing gain of 2 can really be obtained for this MIMO system.

冋 册

638

Now, let us introduce the location variations for the two objects with ␴ = 0.01m, which means the standard deviation for the object position is 1 cm for all directions. We ran the simulations 20 runs and the normalized decoded signal constellation is shown in Figure 14.8. From the simulations, it is clear that with the deviation in object locations the first symbol has almost no effect, and it can be decoded very accurately. For the second symbol, which is attenuated by a factor of 4.5 compared to the first symbol, there is larger variation in its phase. However, this still can be decoded with high accuracy. Next, let us increase the standard deviation further more to ␴ = 0.05m. The simulation was run as before with 20 runs. The normalized decoded signal constellation is shown in Figure 14.9. As seen, even when the standard deviation of the object positions is only 5 cm, or equivalent to 0.17␭ , this 2 × 2 MIMO system cannot resolve the received signals. There is no multiplexing gain in this case as we cannot transmit two symbols simultaneously. Thus, only one symbol can be transmitted at a time or the multiplexing gain is 0. Note that thermal noise is not included in all simulations. Thus, once the vector nature of the problem is introduced, it is possible that no multiplexing gain can be achieved. It is possible that this gain may not be realized even for the case when there is a rich multipath environment as illustrated by this simple example. The use of inverse channel technique based on the reciprocity theorem can provide the maximum multiplexing gain as one can direct energy to each of the receive antenna separately [13].

Figure 14.8 QPSK constellation of the received symbols for ␴ = 1 cm.

639

Figure 14.9 QPSK constellation of the received symbols for ␴ = 5 cm.

14.9.4 Case Study 4 In this example, we illustrate the diversity-multiplexing trade-off in a MIMO system. We use the experimental setup the same as in case 3; however, the sphere and cube are replaced by a metallic box of dimension 1m × 1m × 0.5m placed at the origin, as shown in Figures 14.10 and 14.11. In this case, there is no direct line-of-sight (LOS) transmission from the transmitter to the receiver. For the 2 × 2 MIMO case, the setup is shown in Figure 14.10. The results will be compared with the SISO system shown in Figure 14.11, where there is only one antenna for the transmitter and one for the receiver. Let us consider the SISO case. When the transmitted power is 1W, the received current is I R = 13.989 + j 9.958 [mA]. The power received at the receiver for the SISO case is P SISO = 0.026 [W]. For the MIMO, its channel matrix can be written as H=

冋

册

−0.685 + j 0.122

0.186 + j 0.977

0.186 + j 0.977

−0.685 + j 0.122

The SVD of this channel matrix is as follows:

⭈ 10−3

640

Figure 14.10 2 × 2 MIMO simulation for case 4.

Figure 14.11 SISO simulation for case 4.

641

U= ⌺= V=

冋 冋 冋

册

−0.504 − j 0.495

−0.292 + j 0.644

0.504 + j 0.495

−0.292 + j 0.644

册 册

0.0012

0

0

0.0012

0.707

0.707

−0.707

0.707

The input signal is selected from the QPSK constellation mapping as x = [x 1 x 2 ]T where x 1 = 1 and x 2 = j. Thus, the input to the MIMO system is 0.707 + j 0.707 . To compare the results with the SISO case, we scale the x˜ = Vx = −0.707 + j 0.707 input x˜ such that the total power of 1W is radiated from the MIMO transmitters. Since this MIMO system transmits two symbols at a time, it is equivalent, as the transmitted power is 0.5W per symbol per transmission. The received currents at the receive antennas are as follows is I R1 = −13.791 − j 9.472 [mA] and is I R2 = −1.604 + j 2.491 [mA]. At the receiver, these currents are decoded by using y˜ = U H y as explained in Section 14.4, where I R1 and I R2 are the first and the second components of vector y, respectively. The 0.012 [A]. The received powers received currents at the output of the decoder are y˜ = j 0.012 for each symbol are P R1 = 0.013 [W] and P R2 = 0.013 [W], respectively. Therefore, the total received power for the MIMO case is P MIMO = 0.026 [W], which is equivalent to 0.013W per symbol. However, as mentioned earlier the transmitted power is 0.5 W/ symbol/transmission; for 1 W/symbol/transmission the received power for this MIMO example is 0.026 W/symbol, which is the same as that of the SISO case. This shows that there is no diversity gain in this MIMO example since there is no improvement in the output SNR of the system. However, there is a multiplexing gain of 2 instead as two symbols can be transmitted at a time. This yields an increase in the overall transmission rate of the system. This example shows the diversity-multiplexing trade-off that one needs to consider in designing a MIMO system.

冋

册

冋 册

14.9.5 Case Study 5 An example of multiantenna systems is considered in this example to show that great care is needed when deploying a multiantenna system in practice. It is shown in the previous section that a SIMO and a MISO system would give an improvement in the SNR at the receiver over a SISO system. This is mathematically true when the antennas are ideal point sources and they are operated in free space. This example shows that this improvement in the SNR is not always true especially when the antennas are above a

642

ground plane, which is the case in practice. To make this example more realistic, a transmitter composed of an array of two dipoles is placed 20m above a perfect electric conductor (PEC) ground plane and a receiving dipole antenna is placed 2m above the ground. This scenario represents a base station tower (as a transmitter) and a mobile unit (as a receiver). All of the antennas are conjugated match so that they give maximum power transfer in free space. The operating frequency is at f = 1 GHz and the radius of the dipoles is ␭ /300 (1 mm). The received power of the 2 × 1 MISO system will be compared with the one that would be obtained from a SISO system when its transmitting antenna is placed at the center position of the MISO transmitter. In this comparison, the spacing between the two transmitting antennas for the MISO system and the distance between the transmitter and the receiver are considered as our variables. The spacing between the two transmitting antennas is varied from 0.4␭ –50␭ and the distance between the transmitter and the receiver is varied from 5␭ –100␭ . The scenario of the simulation is shown in Figure 14.12. The received power ratio between the MISO system and the SISO system when they use the same transmitted power is shown in Figure 14.13. From an array processing theory, the gain of 3 dB would be expected from the 2 × 1 MISO system. However, as shown in Figure 14.13, there are many areas (the dark areas) where the received power from the MISO system is less than the one from the SISO system. At some points the MISO system receives as low as −10 dB when compared to the SISO system. Figure 14.14 shows a power ratio when the transmitting antennas are 30␭ apart. It is obvious that there are many positions from the transmitter as the mobile unit moves away from the transmitter that the SISO system performs better than the MISO system.

Figure 14.12 MISO system setup: transmitter is 20m above ground; receiver is 2m above ground; ⌬ = transmitting antenna spacing; D = distance between the transmitter and the receiver. (For the SISO case, the transmitting antenna is placed at the center of the MISO transmitter.)

643

Figure 14.13 Received power ratio between MISO and SISO system when the transmitter is 20m and the receiver is 2m above ground.

Thus, what is predicted from the array processing point of view is not always correct in the presence of ground plane as the electromagnetic wave is vector in nature. These simple examples demonstrate the shortcomings of a total statistical analysis without paying attention to the electromagnetic aspects. That is, without considering the electromagnetic scenario, the MIMO system performance can be misinterpreted. Further study on the electromagnetic effects to MIMO systems still remains open to the researcher. 14.10 CONCLUSIONS In this chapter, the MIMO technology is discussed starting from the basic statistical idea to the implementations. By using multiple antennas at the transmitter and/or at the receiver, the performance of a wireless communication system can be improved. Multiple antennas allow us to transmit signals spatially through a number of independent paths caused by multipath fading. It has shown that the diversity gain or the improvement in the received SNR can be achieved at both ends of the wireless systems via coding schemes. The transmission rate, or multiplexing gain, can also be increased by using multiple antennas

644

Figure 14.14 Received power ratio when the transmitting antennas are 30␭ spacing.

depending on whether the knowledge of channels is available or not. With the knowledge of the channels, the MIMO systems provide a number of independent paths for the transmission. Diversity and multiplexing gain trade-off is discussed as a criterion for consideration in a MIMO system design. The electromagnetic effects to the MIMO channels are shown through numerical simulations. Even though it has not been shown based on a strictly mathematical point of view, it is obvious that without taking into account the electromagnetic effects the expected MIMO system performance may not be realized to its full potential. Great care needs to be considered when designing a wireless system based on an array theory as the vector nature of the electromagnetic fields can totally change the designed system performance. With the introduction of a new dimension (i.e., space) to the wireless communications, the MIMO technology is a promising tool to bring communication systems toward the 4G and beyond. However, its success can only be guaranteed if the design is carried out using fundamental physical principles based on a Maxwellian framework. REFERENCES [1] Foschini, G. J., ‘‘Layered Space-Time Architecture for Wireless Communication in a Fading Environment When Using Multi-Element Antennas,’’ Bell Labs Tech. J., Vol. 1, No. 2, 1996, pp. 41–59.

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[2] Migliore, M. D., ‘‘An Intuitive Electromagnetic Approach to MIMO Communication Systems,’’ IEEE Antennas and Propagation Magazine, Vol. 48, No. 3, June 2006. [3] Migliore, M. D., ‘‘On the Role of the Number of Degrees of Freedom of the Field in MIMO Channel,’’ IEEE Trans. on Antenna and Propagation, Vol. 54, No. 2, February 2006, pp. 620–628. [4] Jensen, M. A., and J. W. Wallace, ‘‘A Review of Antennas and Propagation for MIMO Wireless Communications,’’ IEEE Trans. on Antenna and Propagation, Vol. 52, No. 11, November 2004, pp. 2810–2824. [5] Brennan, D. G., ‘‘Linear Diversity Combining Techniques,’’ Proc. IRE., Vol. 47, June 1959, pp. 1075–1102. [6] Proakis, J. G., Digital Communications, New York: McGraw-Hill, 2001. [7] Bo¨lcskei H., et al., Space-Time Wireless Systems from Array Processing to MIMO Communications, Cambridge, U.K.: Cambridge University Press, 2006. [8] Calhoun, G. M., Third Generation Wireless Systems, Volume 1: Post-Shannon Signal Architectures, Norwood, MA: Artech House, 2003. [9] Watkins, D. S., Fundamentals of Matrix Computations, 2nd ed., New York: Wiley, 2002. [10] Shannon, C. E., ‘‘A Mathematical Theory of Communication,’’ Bell System Tech. J., Vol. 27, June 1948. [11] Shannon, C. E., ‘‘Communication in the Presence of Noise,’’ Proceeding of the IEEE, Vol. 86, No. 2, February 1998, pp. 447–458. [12] Paulraj, A., R. Nabar, and D. Gore, Introduction to Space-Time Wireless Communications, Cambridge, U.K.: Cambridge University Press, 2003. [13] Hwang, S., A. Medouri, and T. K. Sarkar, ‘‘Signal Enhancement in a Near-Field MIMO Environment Through Adaptivity on Transmit,’’ IEEE Trans. on Antenna and Propagation, Vol. 53, No. 2, February 2005. [14] Barbarossa, S., Multiantenna Wireless Communication Systems, Norwood, MA: Artech House, 2005. [15] Alamouti, S. M., ‘‘A Simple Transmit Diversity Technique for Wireless Communications,’’ IEEE J. Sel. Areas. Comm., Vol. 16, No. 8, October 1998, pp. 1451–1458. [16] Zheng, L., and D. N. C. Tse, ‘‘Diversity and Multiplexing: A Fundamental Tradeoff in Multiple-Antenna Channels,’’ IEEE Trans. on Information Theory, Vol. 49, No. 5, May 2003, pp. 1073–1096. [17] Ma, X., and G. B. Giannakis, ‘‘Full-Diversity Full-Rate Complex-Field Space-Time Coding,’’ IEEE Trans. on Signal Processing, Vol. 49, November 2003, pp. 2917–2930. [18] Sarkar, T. K., et al., Smart Antennas, New York: Wiley, 2003. [19] Sarkar, T. K., et al., ‘‘A Discussion About Some of the Principles/Practices of Wireless Communication Under a Maxwellian Framework,’’ IEEE Trans. on Antenna and Propagation, Vol. 54, No. 12, December 2006. [20] Kolundzija, B. M., J. S. Ognjanovic, and T. K. Sarkar, WIPL-D: Electromagnetic Modeling of Composite Metallic and Dielectric Structures, Software and User’s Manual, Norwood, MA: Artech House, 2000. [21] Skolnik M. I., Radar Handbook, 2nd ed., New York: McGraw-Hill, 1990.

Chapter 15 Smart Antennas Nuri Yilmazer, Santana Burintramart, Tapan K. Sarkar, Magdalena Salazar-Palma, Kwok Wa Leung, and Edward Yung

In this chapter, the smart antenna concept is introduced. The smart antenna is an emerging technology that can be used to tackle the capacity, quality, and coverage problems faced by wireless communication under heavy traffic. It finds its applications in many fields such as military, commercial, and medical. The chapter is organized as follows. First, the problems faced in wireless communication are introduced and the need for the smart antennas systems to solve these bottlenecks is explained. Next, the existing algorithms for smart antennas are explained. A direct data domain approach based on a single snapshot of data, in which there is no need to form a covariance matrix, is explained in detail. Next, the eigenvalue method, forward method, backward method, and forward-backward method are introduced, and simulation results and conclusions are provided. 15.1 DEFINITION There has been an explosive growth in the wireless communication area in recent years. This growth has forced system designers to increase the quality of service, coverage, and bandwidth. The smart antenna concept appeared around 1950s. Even though it has been mostly used and designed for military applications—to direct the energy to desired receivers by using an array of transmitters—the smart antenna concept has gained more popularity since the increasing demand in commercial wireless communication. A good source for the historical development of wireless communication can be found in [1]. 647

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Smart antennas are an emerging technology that can be used to tackle the capacity, quality, and coverage problems faced in wireless communication under heavy traffic. Smart antenna systems use a weighted average of the received signals to automatically adjust the beam towards the signal of interest (SOI) under a dynamically changing environment to radiate or receive desired signals while nulling the interferers. It finds its applications in many fields such as military, commercial, and medical. Antennas, used to distribute data to or collect data from surrounding space, have been the most neglected of all the components in a wireless communication system. By using these components in a smart fashion, one can help the designer to use the spectrum efficiently, and increase the quality of the communication network. Smart antenna technology requires the expertise of multidisciplinary areas like antennas, communication, digital signal processing, and adaptive signal processing, to name a few. The reason this particular system is referred to as ‘‘smart antenna’’ is that the antenna beam is steered electronically under dynamically changing environments towards the SOI and the interferers are nulled by using the signal processing techniques in a smart way through a weighted average of the received signals. Multiple antennas provide array gain, diversity gain, and multiuser diversity which increase the signal to interference plus noise ratio (SINR) and help to mitigate the multipath fading effects. Thus, smart antennas can introduce spatial dimension to the design in addition to current multiple access methods. Early implementations of smart antenna arrays were very costly from a computational standpoint as the implemented algorithms were based on the earlier methods established on analog processing and they were not upgraded to exploit the digital nature of the signals. Since the enormous improvement in the analog-to-digital converter (ADC), as well as in signal processing techniques, smart antenna implementation has become feasible. Smart antennas can be implemented using two different methods: switched-beam arrays and adaptive arrays. In switched-beam array systems, the multiple fixed beams are generated to cover the range of interest. The switching algorithm chooses the beam along the direction of maximum signal strength. The switched-beam systems are easier to implement as compared to adaptive arrays. In the case of strong interferers, the method is not efficient since the beam pattern is limited by the main beam width, so interferers may not be nulled out [2–4]. The most common technique to implement the switchedbeam algorithm is done by using Butler matrix array [5]. One point that is often overlooked in this procedure is the inherent assumption that the receivers are assumed to be in the far field of the transmitter. If this condition is not fulfilled then an antenna beam has no meaning, since a beam cannot be defined in the near field of the transmitter. In an adaptive array system, the main beam is steered electronically towards the SOI, and interferers are nulled at the same time by changing the phase of the voltages at the transmitter or receiver antenna array. This is accomplished by taking a weighted average of the received signals. A basic diagram of the system is shown in Figure 15.1, displaying that the main beam is steered towards an intended user or SOI, and simultaneously nulling the interfering user.

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Figure 15.1 Beam forming.

Each antenna element output is multiplied by a complex weight, w i . Multiplying with these weights, the amplitude and the phase are changed, so that the beam is steered. The antenna array pattern is changed dynamically to mitigate multipath effects and interference while increasing the range and reducing the power consumption of the system. In summary, fixed beam system uses multiple fixed beams with narrow beam width. The phase shifts can be implemented by using the Butler matrix so that multiple beams are generated in a simple fashion. Multipath signals and interferers might be enhanced since the systems cannot distinguish between SOI and interferences, so the cochannel interference reduction will be less than in the adaptive system since there is no null steering [6]. Adaptive systems require digital signal processing technology to steer the beam towards the SOI and null the interfering signal from the other directions. They have a better interference rejection performance as compared to the switched-beam systems. Adaptive systems have better coverage and capacity, since they have a better interference rejection. 15.2 WHY SMART ANTENNAS? •

Smart antennas can be used to mitigate the effect of multipath fading in wireless communication. The power of the received signal in a wireless communication

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•

•

generally fluctuates with time as the receiver may be moving or the environment may be changing. Using multiple antennas will give rise to a diversity phenomenon. By taking advantage of the transmit and receive antenna diversities it is sometimes possible to mitigate multipath fading. It reduces the power consumptions by directing the power only along the direction of SOI in a far field environment. Instead of transmitting the signal in all directions as in an original omnidirectional antenna system, the power can be guided towards certain locations and hence reduce the power consumption. Less power is required as compared to the old fashioned systems deployed using omnidirectional antennas. The range is also increased by guiding the narrower beam towards the SOI with the same amount of power. Smart antennas increase the system capacity. It achieves this improvement by maximizing the SINR through nulling the interferers. Smart antennas do the spatial filtering and hence maximize the power towards the SOI.

15.3 INTRODUCTION One of the fundamental limitations in mobile communication systems is that simultaneously many users want to access the base station and thereby establish the first link in the communication chain. The way resources of the base station are distributed to the mobile users is through multiple user access algorithms. Therefore, multiple accesses is implemented by sharing one or more of the four resources of the base station by the various mobile users randomly located in space and time. This sharing can take place in any of the following ways [2, 7]: frequency division multiple access (FDMA), time division multiple access (TDMA), code division multiple access (CDMA), and space division multiple access (SDMA). If a base station has to cover a large geographical area then the region is split into cells where the same carrier frequency can be reused in some of the cells which may not be adjacent to each other [8]. Therefore, for a large number of cells there is a high level of frequency reuse which increases the capacity. In this primitive form the transmitted power of the base station limits the number of cells that may be associated with a base station since the level of interference at a base station is determined by the spatial separation between the cells as the mobile units are using the same frequency. In a TDMA system, each mobile has the entire frequency resource of the base station for a short duration of the time. Each user accesses the entire spectrum of the base station for a finite duration in an ordered sequence. With the advent of digital technology it is possible to have an intermittent connection for each mobile with the base station for a short period of time and in this way the valuable frequency resource of the base station is shared. In a CDMA system, each user is assigned a unique code. In this way the user is allowed to access all the bandwidth like in TDMA and for the complete duration of the call like in FDMA. Since all the users have access to the entire spectrum for all the time, they are interfering

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with each other simultaneously. However, it was found that the capacity of the base station could be further increased by spatially focusing the transmitted power along the direction of the intended users [8]. In this way, the transmission can be achieved at the same carrier frequency simultaneously with different users. This can be accomplished by using an array of antennas at the base station and employing either a switched-beam array or an adaptive beam array to direct the electromagnetic energy to the intended users. Thus, further enhancement in the capacity of a communication system can be achieved through another implementation of this SDMA [9]. This is generally carried out using an adaptive process where we have a collection of antennas called phased arrays. One now dynamically combines the output from each of the antenna elements using different weights. The weights modify the amplitude and the phase of the voltage received at each of the antenna elements. Through an appropriate combination of the voltages that are induced on the antennas, one forms an antenna beam. This antenna beam can be either electronically steered or switched along a certain prefixed direction by selecting a set of a priori weights. Analog beam forming has been going on for a long time since its conception in the 1950s [10]. The advantage of digital beam forming is that one can form any arbitrary low level of sidelobes with an arbitrary small main lobe width along the look direction. The application of the Butler matrix to combine the outputs of the antenna elements is similar in principle to the application of the fast Fourier transform (FFT) to the output voltages available at the antenna elements to form a beam [11, 12]. This is because the far field is simply the Fourier transform of the current distribution on the radiating structure. There is a fundamental philosophical difference in performing the adaptive processing in an analog domain as opposed to carrying out in the digital domain. In analog domain, the weights can never be greater than unity as this will result in a use of an amplifier. Also, in the digital domain one can achieve super-resolution in the processing by being able to cancel interferers within the main lobe, which is not possible in the analog domain. By processing signals in the analog domain, one is limited by the Rayleigh resolution criteria. It states that in order to resolve two closely spaced signals in space, one needs an antenna whose physical size is inversely proportional to the difference in their spatial angles of arrival at the array. Therefore, the closer two signals are located in space, the greater should be the physical size of the antenna in order to separate these two signals. Therefore, the angular resolution of a phased array performing analog processing is determined by the physical length of the array. On the other hand, digital beam forming allows us to go beyond the Rayleigh limit if there is adequate signal strength and enough dynamic range at each of the antenna elements to carry out beam forming [13–15]. When dealing with narrowband electromagnetic signals, a high quality receiving antenna is often composed of an array of half wavelength dipoles, typically spaced a half wavelength apart from each other. One can implement adaptive processing using realistic antenna elements operating in close proximity and incorporate mutual coupling effects. Moreover, there may be coupling between the antenna elements and the platform on which it is mounted. In addition there may be near field scatterers including other antennas, buildings, trees, and so on, near the array that may again distort the beam. A detailed

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study can be found [13, 16]. Also, classical adaptive processing techniques employ multiple snapshots of data to carry out digital beam forming using methods based on the statistics of the data. These algorithms, based on the least mean square (LMS) and the Wiener filtering approach, were developed when the world was analog in nature. Yet with the advent of digital signal processing these same methods are being applied without any modifications and thereby makes their applications in real life prohibitive as they are computationally too expensive and are unsuitable for a fast dynamically changing environment as they require a latent time to collect the snapshots of data [13]. To mitigate these practical shortcomings, recently a direct data domain least squares (D3LS) approach, which operates on a single snapshot of data, has been shown to be very effective in smart antenna technology [13, 17]. The advantages of using a D3LS method over conventional stochastic methodologies are explained in detail [13]. There are various compounding factors like nonstationarity of the data and real-time signal processing issues that are aided by a deterministic model as it is well suited for applications in a highly dynamic environment where processing the data on a snapshot-by-snapshot basis is appropriate. In addition, there is no need to develop a stochastic model for clutter because in a direct data domain approach it is treated as an undesired signal just like any interferer and thermal noise [17]. For a conventional adaptive system a snapshot is defined as the set of voltages measured at the terminals of the antennas. Both the interference and the clutter in this algorithm are treated as undesired electromagnetic signals impinging on the array. Since no covariance matrix is needed in D3LS method and it operates based on a single snapshot, it can be implemented in real time using a modern digital signal processing device. 15.4 BACKGROUND Adaptive beam forming is a method that is able to separate signals which have the same frequency content but are separated in the spatial domain. This provides a means for separating a desired signal from interfering signals in space. An adaptive beam former is able to automatically optimize the array pattern by adjusting the weights associated with each antenna element until a prescribed objective function is satisfied. Adaptive beam forming has been employed in sonar and radar traditionally. It was initiated with the invention of the intermediate frequency sidelobe canceller (SLC) in 1959 by Howells [18]. Applebaum [19] developed the concept of a fully adaptive array in 1966. This algorithm works on the premise that the SNR at the output of the antenna is maximized. The SLC was included as a special case in Applebaum’s work. Another approach, invented by Widrow and Hoff [20], is the LMS algorithm. The LMS algorithm performs well if the defined conditions are chosen carefully. LMS is a simple algorithm to be implemented. Some constraints are defined to ensure that the desired signals are not filtered out while getting rid of unwanted signals. Applebaum’s maximum SNR algorithm and Widrow’s LMS algorithm are very similar algorithms even though they use different approaches. Both algorithms converge to the optimum Wiener solution in the case of stationary signals.

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Capon proposed a different technique for solving the adaptive beam forming problem in 1969. The technique led to an adaptive beam former with a minimum variance distortionless response (MVDR) [21]. Since this algorithm maximizes the likelihood function of the input signal vector, it is also sometimes called the maximum likelihood method (MLM). The MLM is one of the earliest adaptive beam forming techniques that can resolve signals that are separated by a fraction of an antenna beam width. Reed, in 1974, showed that fast adaptivity is achieved by using the sample matrix inversion (SMI) technique in which the adaptive weights can be computed directly from the covariance matrix [22]. The performance of the SMI scheme is not dependent of the eigenvalue spread of covariance matrix, since the covariance matrix is inverted directly. On the other hand, maximum SNR algorithm and the LMS algorithm suffer from slow convergence in the case of large eigenvalue spread of the sample covariance matrix. It is interesting to note that all these algorithms were developed prior to the digital revolution and they really do not exploit adequately the digital nature of the data. It is for this reason a single snapshot-based D3LS algorithm was developed. It is conceptually simple and computationally very fast through the implementation of the FFT and the conjugate gradient method to solve the appropriate equations for the weights [13, 23]. Most of the earlier adaptive techniques were based on statistical methodologies because, in those years, it was not easy to quantify the analog signals of interest. With the advent of digital technology, these techniques were re-employed, this time dealing with digitally sampled data. However, with the design of faster processors, the Wiener filter theory also became available for the enhancement of signals in a noisy environment. It can be seen that the speed of the adaptive processes were greatly enhanced by replacing the LMS algorithm by a conjugate-gradient method, saving CPU time [13, 24]. Algorithms based on LSM will be explained in the following sections. There are several articles on the adaptive algorithms that depend on the statistical methodologies. A good summary about these methodologies can be found in [15]. 15.5 BEAM FORMING In an array beam forming, maximum reception of the signal strength is achieved for a specified direction along which the SOI is arriving from and nulls are placed along the direction of the interfering signals. Thus, the SOI and interferers are at different spatial locations. Each antenna output is multiplied with a complex weight, which adjusts the amplitude and the phase of the induced voltage in such a way that the beam is steered towards the SOI and rejects all the interferers. The conventional and adaptive beam former structure is shown in Figure 15.2. Consider N element antenna array which are linearly spaced, then the array factor can be written as AF (␪ ) =

N −1

∑

n =0

i

An e

2␲ dn sin (␪ ) ␭

(15.1)

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Figure 15.2 A conventional beam former (top) and an adaptive array (bottom).

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Here, d is the distance between the antenna elements. ␪ is the angle of arrival of the signal coming to the antenna array. There are a couple of different algorithms to construct the fixed beam former. Minimum mean square error, minimum variance distortionless responses (MVDR), and maximum likelihood are a few [25]. In this section, we will provide a brief description about some of these algorithms. 15.5.1 Minimum Mean Square Error [15] In this algorithm, the weights of the array can be found by minimizing the mean square error. From Figure 15.2 (bottom), say the output signal y (t ) is summation of the antenna outputs each multiplied with the complex weights, ␻ i . s i (t ) is the i th incoming signal. e (t ) is the error signal, which is the difference between desired signal d (t ) and the array output y (t ). In order to form the beams, a criterion must be satisfied. A cost function is defined to minimize the error so that the beam can be steered towards the SOI. The weights that minimize the error will give the optimum solution. The array output y (t ) can be written as y = wHx

(15.2)

where w H is the complex conjugate transpose of the weight vector w, and signal x is the summation of desired and the undesired signals which may be clutter, other interferers, and thermal noise, and can be written as x (t ) = x s (t ) + x u (t )

(15.3)

where x s (t ) is the desired signal vector, and x u (t ) is the undesired signal vector 1 e

x s (t ) = s (t )

2␲ j d sin (␪ s ) ␭

= s (t ) a s

⯗

j

e

(15.4)

2␲ d (N − 1) sin (␪ s ) ␭

and 1 Number of interferers

x u (t ) =

∑

i =1

e

u i (t )

2␲ j d sin (␪ s ) ␭

⯗

j

e

2␲ d (N − 1) sin (␪ s ) ␭

(15.5)

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␪ s is the arrival angle of the SOI, and ␪ i ’s are the arrival angles of interferers. The weights are chosen in order to minimize the mean-square error (MSE) between the beam former output and the reference signal. The error signal can be written as e (t ) = [d (t ) − w H x (t )]

(15.6)

H is the complex conjugate transpose. The average power of the error to be minimized can be written as E {e 2 (t )} = E {[d (t ) − w H x (t )]2 }

(15.7)

E {e 2 (t )} = E {d 2 (t )} − 2w H r ds + w H R x w

(15.8)

where r dx = E {d (t ) x (t )} is the cross correlation between the desired signal d, and the received signal x. R s = E {x (t ) x H (t )} is the autocorrelation matrix of the received signal x. The minimum MSE can be computed by taking the gradient of (15.8) with respect to w and equating it to zero. So, we get ⵜ (E {e 2 (t )}) = −2r ds + 2R x w = 0

(15.9)

Therefore, the optimum solution for the complex weights will be −1

w opt = R x r dx

(15.10)

This equation is also known as the Weiner-Hopf equation. It is obvious that computation of the optimum weight requires the knowledge of both the correlation matrix of the input signal and the cross correlation matrix between the input signal and the desired signal. Computing the inverse of the autocorrelation matrix can be costly; using the steepest descent algorithm one can reduce the computation time since the weights are computed in a recursive way as will be seen in the following sections. 15.5.2 Minimum Variance Distortionless Response In a minimum variance distortionless response, also known as the minimum variance solution, the goal is to minimize the output noise variance. Array output is given as before y = wHx

(15.11)

y = w H x = w H [x s + x u ] = w H x s + w H x u

(15.12)

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In order to ensure a distortionless response, a constraint must be defined as w H as = 1

(15.13)

where

as =

冋

j

1 e

2␲ d sin (␪ s ) ␭

j

e

2␲ d 2 sin (␪ s ) ␭

j

... e

2␲ dN sin (␪ s ) ␭

册

T

We can compute the variance of the output signal y (t ) as Var ( y ) = w H Rw = w H (R s + R u ) w

(15.14)

R s and R u are the covariance matrices of the desired and undesired signals. By using the Lagrange multiplier method, one can write, ⵜw

冉

冊

w H Ru w + ␤ (1 − w H a s ) = 0 2

(15.15)

␤ is the Lagrange multiplier. This variance can be minimized by taking the gradient of (15.15) and equating it to zero. This results in Ru w − ␤ as = 0

(15.16)

w MVDR = ␤ R u−1 a s

(15.17)

So, the weights will be

␤ can be found as ␤=

1 a sH R u−1 a s

(15.18)

Finally, the optimum weights will be w MVDR =

R u−1 a s a sH R u−1 a s

(15.19)

There are other methods and criteria to solve the adaptive problems; more detailed coverage is presented in [15, 26]. If the environment is changing dynamically, the complex

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weights need to be adjusted in order to track the changes. In the stationary case the weights are found and the beam is fixed, but in adaptive systems these weights must be updated every time new information comes in. The adaptive algorithm chosen is very important since the convergence speed, stability, and complexity are important issues in an adaptive system design. The algorithm must satisfy some chosen criteria for the optimization process. Most commonly used techniques are LMS, recursive least squares (RLS) [15], direct matrix inversion (DMI) [27], neural networks [28], conjugate gradients [24], and constant modulus algorithm (CMA) [29]. The detailed information about these methods can be found in the cited references. Due to limitation in space, we will briefly introduce the LMS algorithm. Least mean square algorithm is introduced by Widrow and Hoff in 1959 [20]. The LMS algorithm recursively computes and updates the weight vector. It is a simple algorithm as compared to other adaptive algorithms. LMS algorithm does the successive corrections to the weight vector iteratively in the direction of the negative gradient vector which eventually leads to the MSE. The updated weight vector, which uses the steepest descent optimization method, is written as [30] w (t + 1) = w (t ) −

1 ␩ [ⵜ (E {e 2 (t )})] 2

(15.20)

Here, ␩ is the step size which controls the rate of convergence of the LMS algorithm. E {e 2 (t )} is the mean square error as defined before in (15.7). Taking the gradient of mean square error gives ⵜ (E {e 2 (t )}) = −2r dx + 2R x w (t )

(15.21)

In this algorithm, evaluating r dx and R x is the major problem. In order to mitigate this bottleneck, the instantaneous values are used R x (t ) = x (t ) x H (t )

(15.22)

r dx (t ) = d (t ) x H (t )

(15.23)

and

So, the updated weights are computed as w (t + 1) = w (t ) + ␩ x (t ) [d (t ) − x H (t ) w (t )]

(15.24)

w (t + 1) = w (t ) + ␩ x (t ) e * (t )

(15.25)

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* denotes the conjugate. The rate of convergence depends on the eigenvalues of the matrix R x . The optimum rate for the step size is chosen to be 0