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HIGH RELIABILITY MAGNETIC DEVICES Design and Fabrication

COLONEL WM. T. MCLYMAN Kg Magnetics, Inc. Idyllwild, California

M A R C E L

H

MARCEL DEKKER, INC.

D E K K E R

Copyright 2002 by Marcel Dekker, Inc. All Rights Reserved.

NEW YORK • BASEL

Library of Congress Cataloging-in-Publication Data McLyman, Colonel William T. High reliability magnetic devices: design and fabrication/Wm. T. McLyman p. cm.—(Electrical and computer engineering; 115) ISBN 0-8247-0818-0 (alk. paper) 1. Magnetic devices—Design and construction—Quality control. 2. Electric inductors. 3. Electronic transformers. I. Title. II. Electrical engineering and electronics; 115 TK454.4.M3 .M35 2002

621.31'4-Kic21

2002073408

This book is printed on acid-free paper. Headquarters Marcel Dekker, Inc. 270 Madison Avenue, New York, NY 10016 tel: 212-696-9000; fax: 212-685-4540 Eastern Hemisphere Distribution Marcel Dekker AG Hutgasse 4, Postfach 812, CH-4001 Basel, Switzerland tel: 41-61-260-6300; fax: 41-61-260-6333 World Wide Web http://www.dekker.com The publisher offers discounts on this book when ordered in bulk quantities. For more information, write to Special Sales/Professional Marketing at the headquarters address above.

Copyright © 2002 by Marcel Dekker, Inc. All Rights Reserved. Neither this book nor any part may be reproduced or transmitted in any form or by any means, electronic or mechanical, including photocopying, microfilming, and recording, or by any information storage and retrieval system, without permission in writing from the publisher. Current printing (last digit): 10 9 8 7 6 5 4 3 21 PRINTED IN THE UNITED STATES OF AMERICA


ELECTRICAL AND COMPUTER ENGINEERING A Series of Reference Books and Textbooks

FOUNDING EDITOR Marlin O. Thurston Department of Electrical Engineering The Ohio State University Columbus, Ohio

1. 2. 3. 4. 5. 6. 7. 8. 9. 10. 11. 12. 13. 14. 15. 16. 17. 18. 19. 20. 21. 22. 23. 24. 25. 26. 27. 28. 29. 30. 31. 32. 33. 34. 35. 36. 37. 38. 39. 40.

Rational Fault Analysis, edited by Richard Saeks and S. R. Liberty Nonparametric Methods in Communications, edited by P. Papantoni-Kazakos and Dimitri Kazakos Interactive Pattern Recognition, Yi-tzuu Chien Solid-State Electronics, Lawrence E. Murr Electronic, Magnetic, and Thermal Properties of Solid Materials, Klaus Schroder Magnetic-Bubble Memory Technology, Hsu Chang Transformer and Inductor Design Handbook, Colonel Wm. T. McLyman Electromagnetics: Classical and Modern Theory and Applications, Samuel Seely and Alexander D, Poulahkas One-Dimensional Digital Signal Processing, Chi-Tsong Chen Interconnected Dynamical Systems, Raymond A. DeCaho and Richard Saeks Modern Digital Control Systems, Raymond G. Jacquot Hybrid Circuit Design and Manufacture, Roydn D. Jones Magnetic Core Selection for Transformers and Inductors: A User's Guide to Practice and Specification, Colonel Wm. T. McLyman Static and Rotating Electromagnetic Devices, Richard H. Engelmann Energy-Efficient Electric Motors: Selection and Application, John C. Andreas Electromagnetic Compossibility, Heinz M. Schlicke Electronics: Models, Analysis, and Systems, James G. Gottling Digital Filter Design Handbook, FredJ. Taylor Multivariable Control: An Introduction, P. K. Sinha Flexible Circuits: Design and Applications, Steve Gurley, with contributions by Carl A. Edstrom, Jr., Ray D. Greenway, and William P. Kelly Circuit Interruption: Theory and Techniques, Thomas E. Browne, Jr. Switch Mode Power Conversion: Basic Theory and Design, K. Kit Sum Pattern Recognition: Applications to Large Data-Set Problems, Sing-Tze Bow Custom-Specific Integrated Circuits: Design and Fabrication, Stanley L. Hurst Digital Circuits: Logic and Design, Ronald C. Emery Large-Scale Control Systems: Theories and Techniques, Magdi S. Mahmoud, Mohamed F. Hassan, and Mohamed G. Darwish Microprocessor Software Project Management, Eli T. Fathi and Cedric V. W. Armstrong (Sponsored by Ontario Centre for Microelectronics) Low Frequency Electromagnetic Design, Michael P. Perry Multidimensional Systems: Techniques and Applications, edited by Spyros G. Tzafestas AC Motors for High-Performance Applications: Analysis and Control, Sakae Yamamura Ceramic Motors for Electronics: Processing, Properties, and Applications, edited by Relva C. Buchanan Microcomputer Bus Structures and Bus Interface Design, Arthur L Dexter End User's Guide to Innovative Flexible Circuit Packaging, Jay J. Miniet Reliability Engineering for Electronic Design, Norman B. Fuqua Design Fundamentals for Low-Voltage Distribution and Control, Frank W. Kussy and Jack L. Warren Encapsulation of Electronic Devices and Components, Edward R. Salmon Protective Relaying: Principles and Applications, J. Lewis Blackburn Testing Active and Passive Electronic Components, Richard F. Powell Adaptive Control Systems: Techniques and Applications, V. V. Chalam Computer-Aided Analysis of Power Electronic Systems, Venkatachari Rajagopalan


41. 42. 43. 44. 45. 46. 47. 48. 49. 50. 51. 52. 53. 54. 55. 56. 57. 58. 59. 60. 61. 62. 63. 64. 65. 66. 67. 68. 69. 70. 71. 72. 73. 74. 75. 76. 77. 78. 79. 80. 81. 82. 83. 84. 85. 86. 87. 88. 89. 90. 91. 92. 93. 94. 95. 96.

Integrated Circuit Quality and Reliability, Eugene R. Hnatek Systolic Signal Processing Systems, edited by Earl E. Swartzlander, Jr. Adaptive Digital Filters and Signal Analysis, Maurice G. Bellanger Electronic Ceramics: Properties, Configuration, and Applications, edited by Lionel M. Levinson Computer Systems Engineering Management, Robert S. Alford Systems Modeling and Computer Simulation, edited by Nairn A. Kheir Rigid-Flex Printed Wiring Design for Production Readiness, Walters. Rigling Analog Methods for Computer-Aided Circuit Analysis and Diagnosis, edited by Takao Ozawa Transformer and Inductor Design Handbook: Second Edition, Revised and Expanded, Colonel Wm. T. McLyman Power System Grounding and Transients: An Introduction, A. P. Sakis Meliopoulos Signal Processing Handbook, edited by C. H. Chen Electronic Product Design for Automated Manufacturing, H. Richard Stillwell Dynamic Models and Discrete Event Simulation, William Delaney and Erminia Vaccari FET Technology and Application: An Introduction, Edwin S. Oxner Digital Speech Processing, Synthesis, and Recognition, Sadaoki Furui VLSI RISC Architecture and Organization, Stephen B. Furber Surface Mount and Related Technologies, Gerald Ginsberg Uninterruptible Power Supplies: Power Conditioners for Critical Equipment, David C. Griffith Polyphase Induction Motors: Analysis, Design, and Application, Paul L Cochran Battery Technology Handbook, edited by H. A. Kiehne Network Modeling, Simulation, and Analysis, edited by Ricardo F. Garzia and Mario R. Garzia Linear Circuits, Systems, and Signal Processing: Advanced Theory and Applications, edited by Nobuo Nagai High-Voltage Engineering: Theory and Practice, edited by M. Khalifa Large-Scale Systems Control and Decision Making, edited by Hiroyuki Tamura and Tsuneo Yoshikawa Industrial Power Distribution and Illuminating Systems, Kao Chen Distributed Computer Control for Industrial Automation, Dobrivoje Popovic and Vijay P. Bhatkar Computer-Aided Analysis of Active Circuits, Adrian loinovici Designing with Analog Switches, Steve Moore Contamination Effects on Electronic Products, Carl J. Tautscher Computer-Operated Systems Control, Magdi S. Mahmoud Integrated Microwave Circuits, edited by Yoshihiro Konishi Ceramic Materials for Electronics: Processing, Properties, and Applications, Second Edition, Revised and Expanded, edited by Relva C. Buchanan Electromagnetic Compatibility: Principles and Applications, David A. Weston Intelligent Robotic Systems, edited by Spyros G. Tzafestas Switching Phenomena in High-Voltage Circuit Breakers, edited by Kunio Nakanishi Advances in Speech Signal Processing, edited by Sadaoki Furui and M. Mohan Sondhi Pattern Recognition and Image Preprocessing, Sing-Tze Bow Energy-Efficient Electric Motors: Selection and Application, Second Edition, John C. Andreas Stochastic Large-Scale Engineering Systems, edited by Spyros G. Tzafestas and Keigo Watanabe Two-Dimensional Digital Filters, Wu-Sheng Lu and Andreas Antoniou Computer-Aided Analysis and Design of Switch-Mode Power Supplies, Yim-Shu Lee Placement and Routing of Electronic Modules, edited by Michael Pecht Applied Control: Current Trends and Modern Methodologies, edited by Spyros G. Tzafestas Algorithms for Computer-Aided Design of Multivariable Control Systems, Stanoje Bingulac and Hugh F. VanLandingham Symmetrical Components for Power Systems Engineering, J. Lewis Blackburn Advanced Digital Signal Processing: Theory and Applications, Glenn Zelnikerand FredJ. Taylor Neural Networks and Simulation Methods, Jian-Kang Wu Power Distribution Engineering: Fundamentals and Applications, James J. Burke Modern Digital Control Systems: Second Edition, Raymond G. Jacquot Adaptive MR Filtering in Signal Processing and Control, Phillip A. Regalia Integrated Circuit Quality and Reliability: Second Edition, Revised and Expanded, Eugene R. Hnatek Handbook of Electric Motors, edited by Richard H. Engelmann and William H. Middendorf Power-Switching Converters, Simon S. Ang Systems Modeling and Computer Simulation: Second Edition, Nairn A. Kheir EMI Filter Design, Richard Lee Ozenbaugh Power Hybrid Circuit Design and Manufacture, Haim Taraseiskey


97. Robust Control System Design: Advanced State Space Techniques, Chia-Chi Tsui 98. Spatial Electric Load Forecasting, H. Lee Willis 99. Permanent Magnet Motor Technology: Design and Applications, Jacek F. Gieras and Mitchell Wing 100. High Voltage Circuit Breakers: Design and Applications, Ruben D. Garzon 101. Integrating Electrical Heating Elements in Appliance Design, ThorHegbom 102. Magnetic Core Selection for Transformers and Inductors: A User's Guide to Practice and Specification, Second Edition, Colonel Wm. T. McLyman 103. Statistical Methods in Control and Signal Processing, edited by Tohru Katayama and Sueo Sugimoto 104. Radio Receiver Design, Robert C. Dixon 105. Electrical Contacts: Principles and Applications, edited by Paul G. Slade 106. Handbook of Electrical Engineering Calculations, edited byArun G. Phadke 107. Reliability Control for Electronic Systems, Donald J. LaCombe 108. Embedded Systems Design with 8051 Microcontrollers: Hardware and Software, Zdravko Karakehayov, Knud Smed Christensen, and Ole Winther 109. Pilot Protective Relaying, edited by Walter A. Elmore 110. High-Voltage Engineering: Theory and Practice, Second Edition, Revised and Expanded, Mazen Abdel-Salam, Hussein Anis, Ahdab EI-Morshedy, and Roshdy Radwan 111. EMI Filter Design: Second Edition, Revised and Expanded, Richard Lee Ozenbaugh 112. Electromagnetic Compatibility: Principles and Applications, Second Edition, Revised and Expanded, David Weston 113. Permanent Magnet Motor Technology: Design and Applications, Second Edition, Revised and Expanded, Jacek F. Gieras and Mitchell Wing 114. High Voltage Circuit Breakers: Design and Applications, Second Edition, Revised and Expanded, Ruben D. Garzon 115. High Reliability Magnetic Devices: Design and Fabrication, Colonel Wm. T. McLyman

Additional Volumes in Preparation Practical Reliability of Electronic Equipment and Products, Eugene R. Hnatek


1926-1993

Dedicated to C. Harris Adams Graduated from California Institute of Technology

1949


Preface This book is intended to provide guidelines and behind-the-scenes background to system and transformer engineers for the design and manufacturing of transformers and inductors of high reliability. There are many applications in which high reliability is a byword, such as manned space vehicles, spacecraft, satellites, flight control systems, missiles, and surveillance drones. Reliability is mandatory because of the high cost of component failure. Pursuing reliability in the manufacturing of transformers and inductors primarily involves attention to detail, coupled with close control in all phases of manufacturing.

I worked at the Jet Propulsion Laboratory (JPL) for almost 30 years as their magnetics design specialist. I have seen all types of vendors and magnetic components, some good and some bad.

Frequently,

components that were rejected were rejected because specifications were not followed. Shortcuts were taken thinking they would save time. Also, in many cases the design engineer would neglect to open the design manual that was provided. The engineer would design and fabricate the transformer or inductor the way he or she had done it on a previous job.

At JPL the guidelines used to design and fabricate high reliability magnetic components were previously found in the DM 509306, Volumes I, II, and III. These books are informative and I still have my original set. The required data is strung out in three volumes, making it very cumbersome to quickly locate anything in them, if one is not familiar with them. JPL finally updated them into a single volume, called JPL D-8208, which is still being revised.

With this book I have tried to bring together all of the existing pertinent literature into one volume. The information in this book comes from many sources: JPL DM 509306, Volumes I, II, and III, JPL D-8208, Mil-STD-981, Mil-T-27, NAVMAT P4855-1A, selected IEEE publications, and discussions with those with years of experience working with these components. Many of the lessons learned by these people have not been captured before in written form. It is hoped that this book will help in achieving standardization and aid in the reduction of the cost of high reliability and the need for custom magnetics. Hopefully it will also provide assistance in preventing design, manufacturing and/or testing mistakes.

The main goal of this book is to provide a comprehensive guide for every aspect of producing a high reliability magnetic component, from choosing the raw materials and construction techniques to in-process inspection, end item testing, and quality assurance recommendation. Colonel Wm. T. McLyman


Acknowledgements I worked at the Jet Propulsion Laboratory (JPL) for almost 30 years. I am proud to say that I worked on almost every major space endeavor that JPL was part of. I had many opportunities to work with Project Manager Tom Gavin, who would use my expertise as the magnetic specialist. It was here that I saw the need for a book that would explain the design and fabrication of high reliability magnetic components. In gathering the material for this book, I have been fortunate in having the assistance and cooperation of JPL, several other companies, and many colleagues. I wish to express my gratitude to all of them.

Jet Propulsion Laboratory Earl A. Cherniack, James C. Arnett, Paul N. Bowerman, Charles J. Bodie, Deputy Section Manager J. K. "Kirk" Bonner, Ph.D. Roberta Certa, Group Lead Fabrication Services Magnetics, Inc. Lowell M. Bosley Mike W. Horgan Scott D. Schmidt Todd A. Wuchevich Micrometals Corp. Dale Nicol Rodon Products, Inc. Steve Freeman Coast Magnetics Satya Dosaj Jai Dosaj Sherwood Associates Edward Sherwood Linear Magnetics Corp. Richard L. Ozenbaugh Fridenberg Research Inc. Jerry Fridenberg

Allen Adams, President


Contents Preface Acknowledgements A History of High Reliability Custom Magnetic Components, 1950 to the Present C. Harris Adams Introduction

James C. Arnett

Symbols

Chapter 1 Transformer and Inductor Design Philosophy Chapter 2 Magnetic Materials Chapter 3 Magnetic Cores Chapter 4 Window Utilization, Magnet Wire, and Insulation Chapter 5 Coil Winding Layer, Foil, and Toroidal Chapter 6 Soldering and Magnet Wire Terminations Chapter 7 Packaging, Enclosures, Mounts, and Headers Chapter 8 Polymeric Impregnate, Embedment, and Adhesives Chapter 9 High Voltage Design Guidelines Chapter 10 Testing, Evaluation, and Quality Assurance


A History of High Reliability Custom Magnetic Components, 1950 to the Present In the years immediately following World War II, the military services recognized the need for and the wisdom of coordinating hardware specifications. During the War each service was procuring hardware using its own specifications. Many specifications were of very similar content. This plurality of similar, but not identical, specifications complicated all aspects of procurement including stocking and quality assurance.

Joint Army-Navy specifications (JAN specs) were first written to coordinate separate

specifications. Then came the Military specifications (MIL) intended for use by all services. This paper traces the history of MIL-T-27, a military specification covering custom magnetic devices. Although MIL-T-27 was initially intended only as a specification for high-grade military magnetics, it has, over the years, come to be used as the document around which "high reliability" magnetic components are specified.

Such procurements used MIL-T-27 in conjunction with other specifications, which imposed

additional requirements. Specification MIL-T-27 (no revision) was issued in September 1949. It was the first issue of an Army, Navy, Air Force joint document covering custom magnetic devices. MIL-T-27 had two parents. These were the Army document 71-4942 and the Navy document 16T30. These documents were those used prior to the issue of MIL-T-27 to specify custom magnetic devices. Since 1949 MIL-T-27 has been subjected to several revisions leading up to the current Revision E. Table 1 lists the progressive revision sequence. Revisions over the years addressed, among other aspects, materials, construction, testing and quality assurance. Since the focus of this paper is directed primarily towards reliability, it is interesting to note that in the "A" revision (1955) the concept of life expectancy was added to the specification. Life expectancies of 10,000, 2500, and

[cm]

Skin effect is illustrated in Figure 1-13. The required wire size is a number 17, magnet wire. The operating frequency is 100 kHz. The skin depth is 0.0209 centimeters. A number 17 magnet wire, operating at 100 kHz, will yield an unused area of 0.00422 cm2, as shown in Figure 1-13. If you take the skin depth s, and, assume it to be the radius of the wire, then, you can calculate the minimum wire area. Take this area and match it with the closest, AWG. Then, take the area of the AWG, and divide it into the required area, and that will be the number of strands. Frequency = 100 kHz #17 Required, area = 0.010398 cm2 Area not used = 0.00422 cm2 8 #26, area = 0.00128 cm2 x 8 = .01024 cm2

Skin depth

Figure 1-13. Skin Depth Illustration. Eddy Current Losses, Proximity Effect Proximity effect is caused by eddy currents induced in a wire, due to the alternating magnetic field of other conductors in the vicinity. Proximity effect is more serious than skin effect because skin effect can be overcome by going to a smaller diameter wire. In the proximity effect, eddy currents, caused by adjacent layers, increase exponentially in amplitude, as the number of layers increases. Proximity effect, skin effect and high frequency together will cause the transformer, with multiple layers, to have losses that are excessive, due to current crowding from the skin effect. With each additional layer, the I2R losses in that layer, increase by the square of the current of the previous layer. Selecting the correct AWG, as well as the winding geometry, is very important in keeping the losses down. The proximity effect can be reduced, significantly, by interleaving the primary and secondary, as shown in Figures 1-14 and Figure 1-15. Copyright 2002 by Marcel Dekker, Inc. All Rights Reserved.

Spatial Illustration 0, mmf

Core

OOO0OOO0

Secondary-layer 2 Secondary-layer 1 Primary-layer 2 — Primary-layer 1

O0000000

Insulation

Bobbin

Magnet wire Center leg

Figure 1-14. Primary and Secondary are Separated.

Core Secondary-layer 2 - —*• Primaiy-layer 2 — —*• Secondary-layer 1 - —*• Primary-layer 1 —

Spatial Illustration (), mmf

00000000

\^

®00@®0O® 00000000 ©0®<S©®©®

^ \^

-y / / Insulation Bobbin

^

Magnet wire Center leg

Figure 1-15. Primary and Secondary are Interleaved. Temperature Rise and Surface Area The heat, generated by the core loss, copper loss, gap loss, and the losses due to the skin effect and proximity effect, produces a temperature rise, which must be controlled to prevent damage to, or the failure of the windings by the breakdown of the insulation at elevated temperatures. This heat is dissipated from the exposed surfaces of the transformer or inductor by a combination of radiation and convection. Therefore, the dissipation is dependent upon the total, exposed surface area of the core and windings. Temperature rise in a transformer winding cannot be predicted with complete precision, despite the fact that many techniques are described in the literature for its calculation. One, reasonably accurate method for open core and winding construction is a homogeneous method. It is also based upon the assumption that the core and winding losses may be lumped together as: Transformer: /> = pcu + / ? / 5

[watts]

Inductor: p

- = Pcu + Pfc + P*'

[watts]

Also, The assumption is made that the thermal energy is dissipated uniformly throughout the surface area of the core and winding assembly. The effective surface area, A,, required to dissipate heat, (expressed as watts dissipated per unit area), is:


4 = ^ , [cm 2 ] "y" is the power density of the average power, dissipated per unit area from the surface of the transformer and, Pv, is the total power lost or dissipated.

The temperature rise that can be expected for various levels of power loss is shown in Figure 1-16. It is based on data obtained from Blume, (1938), for heat transfer, effected by a combination of 55% radiation and 45% convection, from surfaces having an emissivity of 0.95, in an ambient temperature of 25°C, at sea level. Power loss, (heat dissipation), is expressed in watts per square centimeter of the total surface area. Heat dissipation, by convection from the upper side of a horizontal flat surface, is on the order of 15-20% more than from a vertical surface. Heat dissipation, from the underside of a horizontal flat surface, depends upon surface area and conductivity. Below are two, prominently used power loss factors, expressed in watts per square centimeter of the total surface area.

y =0.03^/cm 2 @25°C rise y/ = 0.07H / /cm 2 @25°C rise

1.0 Ambient Temperature

0.1

0.01

I 00

Emissivity 0.95 45% Convection 55% Radiation

0.001 10° C 100° C AT = Temperature Rise, °C Tr =450

Figure 1-16. Temperature Rise Versus Surface Dissipation.


Chapter 2

Magnetic Materials


Table of Contents

1. Introduction 2.

Saturation

3.

Remanence Flux, Br, and Coercivity Hc

4.

Permeability, ju

5.

Hysteresis Loss, Resistivity, p, (core loss)

6.

Introduction to Silicon Steel

7.

Introduction to Thin Tape Nickel Alloys

8.

Introduction to Metallic Glass

9.

Introduction to Soft Ferrites

10. Manganese-Zinc Ferrites 11. Nickel-Zinc Ferrites 12. Introduction to Molypermalloy Powder Cores 13. Introduction to Iron Powder Cores 14. Core Loss


Introduction The magnetic material is the paramount player in the design of magnetic components. The magnetics design engineer has three standard words when making the normal design trade-off study: cost, size, and performance. The engineer will be happy to stuff any two in the bag. The magnetics design engineer is now designing magnetic components that operate from below the audio range to the megahertz range. He is normally asked to design for maximum performance, with the minimum of his parasitic friends' capacitance and leakage inductance. Today, the magnetic materials, the engineer has to work with, are silicon steel, nickel iron (permalloy), cobalt iron (permendur), amorphous metallic alloys, and ferrites. These also have spin-off material variants, such as moly-permalloy powder, sendust powder, and iron powder cores. From this group of magnetic materials, the engineer will make trade-offs with the magnetic properties for his design. These properties are: saturation Bs, permeability u, resistivity p (core loss), remanence Br, and coercivity Hc.

Saturation A typical hysteresis loop of a soft magnetic material is shown in Figure 2-1. When a high magnetizing force is encountered, a point is reached where further increase in H does not cause, useful increase in B. This point is known as the saturation point of that material. The saturation flux density, Bs, and the required magnetizing force, Hs, to saturate the core is shown with dashed lines.

B

Figure 2-1. Typical B-H or Hysteresis Loop of a Soft Magnetic Material. Copyright 2002 by Marcel Dekker, Inc. All Rights Reserved.

Remanence Flux, Br, and Coercivity Hc The hysteresis loop in Figure 2-1 clearly shows the remanence flux density, B r . The remanence flux is the polarized flux remaining in the core after the excitation has been removed. The magnetizing force, -Hc is called coercivity. It is the amount of magnetizing force required to bring the remanence flux density back to zero.

Permeability, ji The permeability of a magnetic material is a measure of the ease in magnetizing the material. Permeability ji, is the ratio of the flux density, B, to the magnetizing force, H.

u = — , [permeability]

The relationship between B and H is not linear, as shown in the hysteresis loop in Figure 2-1. Then it is evident that the ratio, B/H (permeability) also varies. The variation of permeability with flux density B is shown in Figure 2-2. It also shows the flux density at which the permeability is at a maximum.

Bs

u, Permeability

0

Magnetizing Force Figure 2-2. Variation in Permeability ii with B and H.

Hysteresis Loss, Resistivity, p, (core loss) The enclosed area within the hysteresis, shown in Figure 2-1, is a measure of the energy lost in the core material during that cycle. This loss is made up in two components: (1) the hysteresis loss and (2) eddy current loss. The hysteresis loss is the energy loss when the magnetic material is going through a cycling state. The eddy current loss is caused when the lines of flux pass through the core, inducing electrical currents in it. These currents are called eddy currents and produce heat in the core. If the electrical resistance of the core is high, the current will be low; therefore, a feature of low-loss material is high


electrical resistance. In the norm, when designing magnetics components the core loss is a major design factor. Core loss, can be controlled by selecting the right material and thickness. Selecting the correct material, and operating within its limits, will prevent overheating that could result in damage to the wire insulation and/or the potting compound.

Introduction to Silicon Steel Silicon steel was one of the first alloys to be used in transformers and inductors. It has been greatly improved over the years and is probably, pound for pound, the most, widely used magnetic material. One of the drawbacks in using steel in the early years was, as the material became older, the losses would increase. With the addition of silicon to the steel, the advantages were twofold: it increased the electrical resistivity, therefore, reducing the eddy current losses, and it also improved the material stability with age.

Silicon steel offers high saturation flux density, a relatively good permeability at high flux density, and a moderate loss at audio frequency. One of the important improvements made to the silicon steel was in the process called cold rolled, grain-oriented, AISI type M6. This M6 grain oriented steel has exceptionally low losses and high permeability. It is used in applications requiring high performance and the losses will be at a minimum.

Introduction to Thin Tape Nickel Alloys High permeability metal alloys are based primarily on the nickel-iron system. Although Hopkinson investigated nickel-iron alloys as early as 1889, it was not until the studies by Elmen, starting in about 1913, on properties in weak magnetic fields and effects of heat-treatments, that the importance of the Ni-Fe alloys was realized. Elmen called his Ni-Fe alloys, "Permalloys," and his first patent was filed in 1916. His preferred composition was the 78 Ni-Fe alloy. Shortly after Elmen, Yensen started an independent investigation that resulted in the 50Ni-50Fe alloy, "Hipernik," which has lower permeability and resistivity but higher saturation than the 78-Permalloy, (1.5 tesla compared to 0.75 tesla), making it more useful in power equipment. Improvements in the Ni-Fe alloys were achieved by high temperature anneals in hydrogen atmosphere, as first reported by Yensen. The next improvement was done by using grain-oriented material and annealing it, in a magnetic field, which was also in a hydrogen atmosphere. This work was done by Kelsall and Bozorth. Using these two methods, a new material, called Supermalloy, was achieved. It has a higher permeability, a lower coercive force, and about the same flux density as 78-Permalloy. Perhaps the most important of these factors is the magnetic anneal, which, not only increases permeability, but also provides a "square" magnetization curve, important in high frequency power conversion equipment. Copyright 2002 by Marcel Dekker, Inc. All Rights Reserved.

In order to obtain high resistance, and therefore, lower core losses for high frequency applications, two approaches have been followed: (1) modification of the shape of metallic alloys and (2) development of magnetic oxides. The result was the development of thin tapes and powdered alloys, in the 1920's, and thin films in the 1950's. The development of thin film has been spurred by the requirements of aerospace power conversion electronics from the mid 1960's to the present. The Ni-Fe alloys are available in thicknesses of 2 mil, 1 mil, 0.5 mil, 0.25 and 0.125 mil. The material comes with a round or square B-H loop. This gives the engineer a wide range of sizes and configurations from which to select for his/her design. The iron alloy properties for some of the most popular materials are shown in Table 2-1. Also given in Table 2-1 is the Figure number for the B-H loop of each of the magnetic materials.

Table 2-1 Magnetic Properties for Selected Iron Alloys Materials.

Iron Alloy Material Properties Material Name

Silicon

Composition

3% Si

Initial

Flux Density

Curie

dc, Coercive

Density

Typical

Permeability

Tesla

Temperture

Force, He

grams/cm

B-H Loop

Of~*

Oersteds

5

Figures

Hi

Bs

1.5 K

1.5-1.8

750

0.4-0.6

7.3

(2-3)

0.8 K

1.9-2.2

940

0.15-0.35

8.15

(2-4)

2K

1.42-1.58

500

0.1-0.2

8.24

(2-5)

12 K-100 K

0.66-0.82

460

0.02-0.04

8.73

(2-6)

10K-50K

0.65-0.82

460

0.003-0.008

8.76

(2-7)

97% Fe

Supermendur*

49% Co 49% Fe 2%V

Orthonol

50%Ni 50% Fe

Permalloy

79% Ni 1 7% Fe 4% Mo

Supermalloy

78%Ni 1 7% Fe 5% Mo

* Field Anneal


B, tesla l.O~

1.2-

Magnesil sil ^steresis Loop

: /" /

0.80.4-

| 1 1.6

- 0.4 1.2

0.8

).4

-

0.8

1.2

1

H, oersteds

- 0.4

1

~ 0.8

^y :

~ 1.2 1.6

Figure 2-3. Silicon B-H Loop: 97% Fe 3% Si.

1.6 Supermendur DC Hysteresis Loop

1.2 0.8 0.4 0.4

1.6

1.2

0.8

0.4

0.8

1.2

1.6

H, oersteds

0.4 0.8 1.2 1.6

Figure 2-4. Supermendur B-H Loop: 49% Fe 49% Co 2% V.


B, tesla -r

1.6

1.2

Orthonol DC Hysteresis Loop 0.8 0.4 0.2 0.8

0.6

0.4

0.:

0.4

0.4

0.6

0.!

H, oersteds

0.8 1.2 1.6

Figure 2-5. Orthonol B-H loop: 50% Fe 50% Ni.

B, tesla 0.8 -r

Square Permalloy 80 DC Hysteresis Loop

0.12 0.16

Figure 2-6. Square Permalloy 80 B-H loop: 79% Ni 17% Fe 4% Mo.


Supermalloy DC Hysteresis Loop

0.04 0.08 0.12 0.16

Figure 2-7. Supermalloy B-H Loop: 78% Ni 17% Fe 5% Mo.

Introduction to Metallic Glass The first synthesis of a metallic glass, drawing wide attention among material scientists, occurred in 1960. Klement, Willens and Duwez reported that a liquid, AuSi alloy, when rapidly quenched to liquid nitrogen temperature, would form an amorphous solid. It was twelve years later that Chen and Polk produced ferrous-based metallic glasses in useful shapes with significant ductility. Metallic glasses have since survived the transition from laboratory curiosities to useful products, and currently, are the focus of intensive technological and fundamental studies.

Metallic glasses are generally produced, by liquid quenching, in which a molten metal alloy is rapidly cooled, at rates on the order of 10* degrees/sec.; through the temperature, at which crystallization normally occurs. The basic difference between crystalline, (standard magnetic material), and glassy metals is in their atomic structures. Crystalline metals are composed of regular, three-dimensional arrays of atoms, which exhibit long-range order. Metallic glasses do not have long-range structural order. Despite their structural differences, crystalline and glassy metals of the same compositions exhibit nearly the same densities.

The electrical resistivities of metallic glasses are much larger, (up to three times higher), than those of crystalline metals of similar compositions. The magnitude of the electrical resistivities and their temperature coefficients in the glassy and liquid states are almost identical. Copyright 2002 by Marcel Dekker, Inc. All Rights Reserved.

Metallic glasses are quite soft magnetically. The term, "soft," refers to a large response of the magnetization to a small-applied field. A large magnetic response is desirable in such applications as transformers and inductors. The obvious advantages of these new materials are in high frequency applications with their high induction, high permeability and low core loss. There are four amorphous materials that have been used in high frequency applications: 2605SC, 2714A, 2714AF and Vitroperm 500F. Material 2605SC offers a unique combination of high resistivity, high saturation induction, and low core loss, making it suitable for designing high frequency dc inductors. Material 2714A is a cobalt material that offers a unique combination of high resistivity, high squareness ratio Br/Bs, and very low core loss making it suitable for designing high frequency aerospace transformers and mag-amps. The Vitroperm 500F is an iron based material with a saturation of 1.2 tesla and is wellsuited for high frequency transformers and gapped inductors. The high frequency core loss for the nanocrystal E 2000 is much lower than ferrite, even operating at a high flux density. The amorphous properties for some of the most popular materials are shown in Table 2-2. Also given in Table 2-2 is the Figure number for the B-H loop of each of the magnetic materials.

Table 2-2. Magnetic Properties for Selected Amorphous Materials.

Amorphous Material Properties Material

Major

Initial

Flux Density

Curie

dc, Coercive

Name

Composition

Permeability

Tesla

Temp.

Force, He

Hi

B5

°C

Oersteds

5

Figures

1.5K

1.5-1.6

370

0.4-0.6

7.32

(2-8)

0.8K

0.5-0.65

205

0.15-0.35

7.59

(2-9)

2K

0.5-0.65

205

0.1-0.2

7.59

(2-10)

30K-80K

1.0-1.2

460

0.02-0.04

8.73

(2-11)

2605SC

8 1 % Fe

Density

Typical

grams/cm" B-H Loop

13.5%B 3. 5% Si

2714A

66% Co

E 1000

15% Mo 4% Fe

2714AF

66% Co 15% Mo 4% Fe

Nanocrystal Vitroperm 500F*

* Vitroperm is the trademark of Vacuumschmelze.


B, Tesla 1.6 T Metglas Type 2605SC DC Hysteresis Loop

1.2 0.8 0.4

H, oersted H

1

1

0.6

1

f-

0.4

0.2

0.2 0.4 0.6 0 4 H, oersted

1.2 - 1.6

Figure 2-8. Amorphous 2605SC B-H Loop: 78% Ni 17% Fe 5% Mo.

B, Tesla

0.6 0.5 ( - r Metglas Type 2714A DC Hysteresis Loop

0.4

-

0.3 0.2

-

o.i 1

1

0.05

1

1

0.03

i

1

0.01 I

0.01 . Q i

i

0.03 t

t

0.05 i

i

H, oersted

- 0.2 - 0.3 - 0.4

J- ) 0.5

- n^

Figure 2-9. Amorphous 2714A B-H Loop: 78% Ni 17% Fe 5% Mo.


Tesla

Metglas Type 2714AF DC Hysteresis Loop

Figure 2-10. Amorphous 2714AF B-H Loop: 78% Ni 17% Fe 5% Mo.

Vitroperm 500F 10 Hz

Figure 2-11. Vitroperm 500F B-H loop: 78% Ni 17% Fe 5% Mo.


Introduction to Soft Ferrites In the early days of electrical industry, the need for the indispensable magnetic material was served by iron and its magnetic alloys. However, with the advent of higher frequencies, the standard techniques of reducing eddy current losses, (using laminations or iron powder cores), was no longer efficient or cost effective. This realization stimulated a renewed interest in "magnetic insulators," as first reported by S. Hilpert, in Germany, in 1909. It was readily understood that, if the high electrical resistivity of oxides could be combined with desired magnetic characteristics, a magnetic material that was particularly well-suited for high frequency operation would result. Research to develop such a material was being performed in various laboratories all over the world, such as, by V. Kato, T. Takei, and N. Kawai in the 1930's in Japan, and by J. Snoek of the Philips' Research Laboratories in the period 1935-1945, in the Netherlands. By 1945, Snoek had laid down the basic fundamentals of the physics and technology of practical ferrite materials. In 1948, the Neel Theory of ferromagnetism provided the theoretical understanding of this type of magnetic material. Ferrites are ceramic, homogeneous materials composed of oxides; iron oxide is their main constituent. Soft ferrites can be divided into two major categories; manganese-zinc and nickel-zinc. In each of these categories, changing the chemical composition or manufacturing technology can manufacture many different Mn-Zn and Ni-Zn material grades. The two families of Mn-Zn and Ni-Zn ferrite materials complement each other and allow the use of soft ferrites from audio frequencies to several hundred megahertz. Manufacturers do not like to handle manganese-zinc in the same area, or building with nickelzinc, because one contaminates the other, which leads to poor performance yields. The basic difference between Manganese-Zinc and Nickel-Zinc is shown in Table 2-3. The biggest difference is ManganeseZinc has a higher permeability and Nickel-Zinc has a higher resistibility. Shown in Table 2-4 are some of the most popular ferrite materials. Also, given in Table 2-4, is the Figure number for the B-H loop of each of the materials. Table 2-3. Comparing Manganese-Zinc and Nickel-Zinc Basic Properties.

Basic Ferrite Material Properties Materials Manganese Zinc Nickel Zinc

Tesla

Curie Temperture, °C

dc, Coercive Force, H c Oersteds

0.3-0.5 0.3-0.5

100-300 150-450

0.04-0.25 0.3-0.5

Initial Permeability

Flux Density P D

Hi 750-15 K 15-1500


max

Resistivity Q - cm 10-100

106

Manganese-Zinc Ferrites This type of soft femte is the most common, and is used in many more applications than the nickel-zinc ferrites. Within the Mn-Zn category, a large variety of materials are possible. Manganese-zinc ferrites are primarily used at frequencies less than 2 MHz.

Nickel-Zinc Ferrites This class of soft ferrite is characterized by its high material resistivity, several orders of magnitude higher than Mn-Zn ferrites. Because of its high resistivity, Ni-Zn ferrite is the material of choice for operating from 1-2 MHz to several hundred megahertz. The material permeability, u ni , has little influence on the effective permeability, ue, when the gap dimension is relatively large, as shown in Table 2-5.

Table 2-4. Magnetic Properties for Selected Ferrite Materials.

Ferrites Material Properties Magnetic

Initial

Material

Permeability

Tesla

Name

Hi

K

Flux Density Residua! Flux

Curie

dc, Coercive

Density

Typical

Tesla

Temperture

Force, He

grams/cm

B-H Loop

Bs(o; 1 5 Oe

Br

O/"~*

Oersteds

5

Figures

1500

0.48T

0.08T

>230

0.2

4.7

(2-12)

R

2300

0.50T

0.12T

>230

0.18

4.8

(2-13)

P

2500

0.50T

0.12T

>230

0.18

4.8

(2-13)

F

5000

0.49T

0.1 OT

>250

0.2

4.8

(2-14)

W

10,000

0.43T

0.07T

>125

0.15

4.8

(2-15)

H

15,000

0.43T

0.07T

>125

0.15

4.8

(2-15)

Table 2-5. Permeability, and its Effect on Gapped Inductors.

Com] taring Material Permeabilities Material K R P F *Core , ETD44

Urn

1500 2300 2500 3000


Gap, inch 0.04 0.04 0.04 0.04

Gap, cm 0.101 0.101 0.101 0.101

*MPL, cm 10.4 10.4 10.4 10.4

He

96 98 99 100

Bm, Tesla

K Material

0.5

1.0 25 °C

1.5

2.0

2.5

Bm = 0.460T (a) 15 oersted

100 °C Bm = 0.3 SOT @ 15 oersted

Figure 2-12. Ferrite B-H loop, K Material at 25 and 100°C.

P & R Material

0.2

25 °C Bm = 0.500T (u) 15 oersted 100 °C Bm = 0.375T (g> 15 oersted

Figure 2-13. Ferrite B-H loop, P & R Material at 25 and 100 °C.


Bm, Tesla 0.5 T F Material

Figure 2-14. Fernte B-H loop, F Material at 25 and 100 ° C.

W & H Material

0.2

100 ''C Bm - 0 220T in' 15 oersted

Figure 2-15. Ferrite B-H loop, W & H Material at 25 and 100 ° C.


Table 2-6. Ferrite Materials, Manufacturers' Cross Reference.

Ferrite Material Cross Reference 1500 Power

Permeability Application

3000 5000 2500 Power Filter Power Material Designation

2300 Power

Manufacturer's Magnetics Thomson LCC Philips Comp. Fair-Rite

1

K 3F4

Siemens

N47

TDK Corp. MMG Ceramic Mag Tokin Ferrite Int.

PC50 MN67

2

R

L2 3F3 78 N67 PC44 F44 MN80 HBM

3

P B2 3C85 77 N27 PC44 F5 2500B TSF-05

F Bl 3C81 N41 H7A F5C MN8CX 3100B TSF-10

J A4 3E2A 75 T35 HP5 F-10 MN60 5000B TSF-15

10,000 Filter

15,000 Filter

W A2 3E5 76 T38 H5C2 F-39 MC25 12001H

H

T46 H5D

1. High Frequency power material 250 kHz & up. 2. Lowest loss at 80°-100°C, 25 kHz to 250 kHz. 3. Lowest loss at 60°C-80°C.

Introduction to Molypermalloy Powder Cores The nickel-iron (Ni-Fe) high permeability magnetic alloys (permalloy) were discovered in 1923, and in 1927. Permalloy alloys were successfully used in powder cores, greatly contributing to the carrier wave communications of the time.

In the early 1940's, a new material, trademarked molybdenum permalloy powder, (MPP), was developed into cores by the Bell Telephone Laboratory and the Western Electric Company. This new material was developed for loading coils, and filtering coils, and transformers at audio and carrier frequencies in the telephone facility. The use of such cores has been extended to many industrial and military circuits. The stability of permeability and core losses, with time, temperature, and flux level, are particularly important to engineers designing tuned circuits and timing circuits. This new material has given reliable and superior performance over all past powder core materials.

Molybdenum permalloy powder, [2 Molybdenum (Mo)-82 Nickel (Ni)-16 Iron (Fe)], is made by grinding hot-rolled and embrittled cast ingots; then, the alloy is insulated and screened to a fineness of 120 mesh for use in audio frequency applications, and 400 mesh for use at high frequencies. Copyright 2002 by Marcel Dekker, Inc. All Rights Reserved.

In the power conversion field, the MPP core has made its greatest impact in switching power supplies. The use of MPP cores and power MOSFET transistors has permitted increased frequency, resulting in greater compactness and weight reduction in computer systems. The power supply is the heart of the system. When the power supply is designed correctly, using a moderate temperature rise, the system will last until it becomes obsolete. In these power systems there are switching inductors, smoothing choke coils, common mode filters, input filters, output filters, power transformers, current transformers and pulse transformers. They cannot all be optimally designed, using MPP cores. But, in some cases, MPP cores are the only ones that will perform in the available space with the proper temperature rise.

Introduction to Iron Powder Cores The development of compressed iron powder cores as a magnetic material for inductance coils stemmed from efforts of Bell Telephone Laboratory engineers to find a substitute for fine iron-wire cores. The use of iron powder cores was suggested by Heaviside, in 1887, and again, by Dolezalek in 1900.

The first iron powder cores of commercially valuable properties were described by Buckner Speed, in U.S. Patent No. 1274952, issued in 1918. Buckner Speed and G.W. Elman published a paper in the A.I.E.E. Transactions, "Magnetic Properties of Compressed Powdered Iron," in 1921. This paper describes a magnetic material, which is well-suited to the construction of cores in small inductance coils and transformers, such as those used in a telephone system. These iron powder cores were made from 80 Mesh Electrolytic Iron Powder. The material was annealed, then, insulated by oxidizing the surface of the individual particles. In this way, a very thin and tough insulation of grains of iron was obtained; this did not break down when the cores were compressed. A shellac solution was applied to the insulated powder as a further insulator and binder. This was how toroidal iron powder cores were manufactured by Western Electric Company, until about 1929. Today's iron powder cores are manufactured much the same way, using highly pure iron powder and a more exotic insulator and binder. The prepared powder is compressed under extremely high pressures to produce a solid-looking core. This process creates a magnetic structure with a distributed air-gap. The inherent high saturation flux density of iron combined with the distributed air-gap produces a core material with initial permeability of less than 100, and with high-energy storage capabilities.

The dc current does not generate core loss, but an ac or ripple current does generate core loss. Iron powder material has higher core loss than some other more expensive core materials. Most dc-biased inductors have a relatively small percentage of ripple current and, thus, core loss will be minimal. However, core loss will sometimes become a limiting factor in applications with a relatively high percentage of ripple current at


very high frequency. Iron powder is not recommended for inductors with discontinuous current or transformers with large ac flux swings.

Low cost, iron powder cores are typically used in today's, low and high frequency power switching conversion applications, for differential-mode, input and output power inductors. Because iron powder cores have such low permeability, a relatively large number of turns are required for the proper inductance, thus keeping the ac flux at a minimum. The penalty for using iron powder cores is usually found in the size and efficiency of the magnetic component.

There are four standard powder materials available for power magnetic devices: Molypermalloy (MPP) Powder Cores with a family of curves, as shown in Figure 2-20; High flux (HF) Powder Cores with a family of curves, as shown in Figure 2-21; Sendust Powder Cores, (Kool Mu), with a family of curves, as shown in Figure 2-22; and Iron Powder Cores, with a family of curves, as shown in Figure 2-23. The powder cores come in a variety of permeabilities. This gives the engineer a wide range in which to optimize the design. The powder core properties for the most popular materials are shown in Table 2-7. Also, given in Table 27, is the Figure number for the B-H loop of each of the powder core materials. In Table 2-8 is a listing of the most popular permeabilities for each of the powder core materials.


Table 2-7. Powder Core Material Properties.

Powder Core Material Properties Material Name MPP

High Flux Sendust (Kool Mu) Iron Powder

Initial Flux Density dc, Coercive Curie Composition Permeability Temperture Force, He Tesla Oersteds °C Bs Hi 14-550 0.7 0.3 80% Ni 450 20% Fe 1 14-160 50% Ni 1.5 360 50% Fe 26-125 85% Fe 1 0.5 740 9% Si 6% Al 5.0-9.0 4.0- 100 0.5 - 1.4 770 100%Fe

grams/cm 5 8.5

Typical B-H Loop Figures (2-16)

8

(2-17)

6.15

(2-18)

3.3 - 7 . 2

(2-19)

Density

Table 2-8. Standard Powder Core Permeabilities.

Standard Powder Core Permeabilities Powder Material

MPP

High Flux

Sendust (Kool Mu)

Iron Powder

Initial Permeability, a,

10 14 26 35 55 60 75 90 100 125 147 160 173 200 300 550


X X X

X X

X

X

X

X X X

X X X X X X X

X X X

X

X X X X X

Molypermalloy MPP 125 Perm

50 100 150 200 250 I 250 200 150 100

50

Figure 2-16. Molypermalloy Powder Core, 125 Perm.

Tesla High Flux HF 125 Perm

50 100 150 200 250 i 1 1 250 200 150 100

50

Figure 2-17. High Flux Powder Core, 125 Perm.


Kool Mu 125 Perm

50 250 200 150 100

100 150 200 250

50

Figure 2-18. Sendust (Kool Mu) Powder Core, 125 Perm.

Tesla Iron Powder-52 75 Perm

50 100 150 200 250 200 150 100

50

Figure 2-19. Iron Powder (-52) Core, 75 Perm.


250

i i i i i i i MPP Powder Cores

100

i

i

i

DC Magnetizing Force (Oersteds) i i i i i i i i i i i i i

1000

100 Figure 2-20. Permeability Versus dc Bias for Molypermalloy Powder Cores.

i i

I I I I I I I I High Flux Powder Cores

DC Magnetizing Force (Oersteds)

1000 Figure 2-21. Permeability Versus dc Bias for High Flux Powder Cores. Copyright 2002 by Marcel Dekker, Inc. All Rights Reserved.

i I i i I I I I I I Sendust Powder Cores (Kool Mu)

100

80

OJ

60

I CM 13

40

a

20 DC Magnetizing Force (Oersteds) I I I I i I I I I I I I I

1.0

100

10

1000

Figure 2-22. Permeability Versus dc Bias for Sendust Powder Cores.

I 1 I I I I II Iron Powder Cores

100

80

S v.

60

o» CH

13 40

20 DC Magnetizing Force (Oersteds) I

I I

I I I

1.0

L

10

i

i

1 I 1 I I

100

Figure 2-23. Permeability Versus dc Bias for Iron Powder Cores. Copyright 2002 by Marcel Dekker, Inc. All Rights Reserved.

I

I

I I I I I

1000

Core Loss The designer of power magnetic components, such as transformer and inductors, requires specific knowledge about the electrical and magnetic properties of the magnetic materials used in these components. There are two magnetic properties that are of interest to the design engineer, the dc and the ac. The dc, B-H hysteresis loop is a very useful guide for comparing the different types of magnetic materials. It is the ac, magnetic properties that are of interest to the design engineer. One of the most important ac properties is the core loss. The ac core loss is a function of the magnetic material, magnetic material thickness, magnetic flux density Bac, frequency f, and operating temperature. The choice of the magnetic material is, thus, based upon achieving the best characteristic, using the standard trade-off, such as cost, size, and performance.

All manufacturers do not use the same units when describing their core loss. The user should be aware of the different core loss units when comparing different magnetic materials. A typical core loss graph is shown in Figure 2-24. The vertical scale is core loss and the horizontal scale is flux density. The core loss data is plotted at different frequencies, as shown in Figure 2-24.

100

Frequency #1 Frequency #2 Frequency #3

10

o -J

o U 1.0

0.1 0.01

0.1

1.0 Flux Density

Figure 2-24. Typical Graph for Plotting Core Loss at Different Frequencies. Copyright 2002 by Marcel Dekker, Inc. All Rights Reserved.

Vertical Scale Here is a list of core loss units used by manufacturers: 1.

watts per pound

2.

watts per kilogram

3.

milliwatts per gram

4.

milliwatts per cubic centimeter (cm1)

Horizontal Scale Here is a list of flux density units used by manufacturers: 1.

gauss

2.

kilogauss

3.

tesla

4.

millitesla

The data can be plotted or presented in either hertz or kilohertz. Manufacturers are now presenting the core loss in an equation form such as: watts/kilogram = k f ( m ] B ( " ] Here, again, the units will change from one manufacturer to another.


Chapter 3

Magnetic Cores


Table of Contents

1. Introduction 2. Core Type and Shell Type Construction 3. Types of Core Materials 4. Eddy Currents and Insulation 5. Laminations 6. Annealing and Stress-Relief 7. Stacking Laminations and Polarity 8. Flux Crowding 9. Exciting Current 10. Tape Wound C, EE, and Toroidal Cores 11. Tape Toroidal Cores 12. Toroidal, Powder Core 13. Dimensional Outline for El Laminations 14. Dimensional Outline for UI Laminations 15. Dimensional Outline for LL Laminations 16. Dimensional Outline for DU Laminations 17. Dimensional Outline for Three Phase Laminations 18. Dimensional Outline for Tape Wound C, EE, and Toroidal Cores 19. Dimensional Outline for EE and El, Ferrite Cores 20. Dimensional Outline for EE and El Planar, Ferrite Cores 21. Dimensional Outline for EC, Ferrite Cores 22. Dimensional Outline for ETD, Ferrite Cores 23. Dimensional Outline for ETD/(low profile), Ferrite Cores 24. Dimensional Outline for ER, Ferrite Cores 25. Dimensional Outline for EFD, Ferrite Cores 26. Dimensional Outline for EPC, Ferrite Cores 27. Dimensional Outline for PC, Ferrite Cores 28. Dimensional Outline for EP, Ferrite Cores 29. Dimensional Outline for PQ, Ferrite Cores 30. Dimensional Outline for PQ/(low profile), Ferrite Cores 31. Dimensional Outline for RM, Ferrite Cores 32. Dimensional Outline for RM/(low profile), Ferrite Cores


33. Dimensional Outline for DS, Ferrite Cores 34. Dimensional Outline for UUR, Ferrite Cores 35. Dimensional Outline for UUS, Ferrite Cores 36. Dimensional Outline for Toroidal, Ferrite Cores 37. Dimensional Outline for Toroidal, MPP Powder Cores 38. Dimensional Outline for Toroidal, Iron Powder Cores


Introduction The key ingredient in a magnetic device is the magnetic field (flux) created when current is passed through a coiled wire. The ability to control (channel, predict, conduct), the magnetic field (flux) is critical to controlling the operation of the magnetic device.

The ability of a material to conduct magnetic flux is defined as permeability. A vacuum is defined as having a permeability of 1.0 and the permeability of all other materials is measured against this baseline. Most materials, such as air, paper, and wood are poor conductors of magnetic flux, in that they have low permeability. If wire is wound on a dowel, it exhibits a magnetic field exactly, as shown in Figure 3-1. There are a few materials, such as iron, nickel, cobalt, and their alloys that have high permeabilities, sometimes ranging into the hundreds of thousands. These materials and their alloys are used as the base materials for all core materials.

Coil

Dowel

Figure 3-1. Air Core with an Intensified Magnetic Field. The main purpose of the core is to contain the magnetic flux and create a well defined, predictable path for the flux. This flux path, and the mean distance covered by the flux within the magnetic material, is defined as the magnetic path length (MPL) (see Figure 3-2). The magnetic path length and permeability are vital keys in predicting the operation characteristic of a magnetic device. Selection of a core material and geometry are usually based on a compromise between conflicting requirements, such as size, weight, temperature rise, flux density, core loss, and operating frequency.


Magnetic Path Length

Flux,

Magnetic Core

Figure 3-2. Magnetic Core Confines the Magnetic Field. Core Type and Shell Type Construction There are two types of construction for magnetic cores, core type and shell type. The shell type construction is shown in Figure 3-3 and the core type construction is shown in Figure 3-4. In the shell type of construction, shown in Figure 3-3, the core surrounds the coil. In the shell type of construction the magnetic fields are around the outside of the coil. The advantage of this configuration is that it requires only one coil. In the core type of construction, shown in Figure 3-4, the coils are outside of the core. A good example of this is a toroid, where the coil is wound on the outside of a core.

E-I Core

Flux,

Coil

Figure 3-3. Shell Type Construction: the Core Surrounds the Coil.

C Core

Flux, Coils

\

\

Figure 3-4. Core Type Construction the Coil Surrounds the Core. Copyright 2002 by Marcel Dekker, Inc. All Rights Reserved.

Types of Core Materials Magnetic cores are made of three basic materials. The first is the bulk metal, the second is the powdered materials, and the third is ferrite. The bulk metals are processed from the furnace into ingots. Then, the material is put into a process of hot and cold rolling. The rolling process produces a sheet of material with a thickness ranging from 0.004 to 0.031 mils that can be punched into laminations. It can be further rolled to a thickness ranging from 0.002 to 0.000125 mils, then slit and wound into tape cores, such as C cores, E cores and toroids.

The powder cores, such as powder molypermalloy and powdered iron materials, are die-pressed into toroids, EE cores and slugs. Powder core processing starts at the ingot, then, goes through various steps of grinding until the powder is the right consistency for the required performance. Normally, powder cores are not machined after processing. Ferrites are ceramic material of iron oxide, alloyed with oxides or carbonate of manganese, zinc, nickel, magnesium, or cobalt. Alloys are selected and mixed, based on the required permeability of the core.

Then, these mixtures are molded into the desired shape with pressure of approximately 150-200 tons per square inch and fired at temperatures above 2000 degrees F. After the parts are made, they are usually tumbled to remove burrs and sharp edges, which are characteristic of this process. Ferrites can be machined to almost any shape to meet the engineer's needs.

Eddy Currents and Insulation Transformers operating at moderate frequency require the reduction of eddy current losses in the magnetic material. To reduce the eddy current losses to a reasonable value requires electrical steel to have adequate resistivity. Also, it needs to be rolled to a specific thickness, and it needs effective electrical insulation or coating of the magnetic material.

If an alternating voltage is applied to the primary winding, as shown in Figure 3-5, it will induce an alternating flux in the core. The alternating flux will, in turn, induce a voltage on the secondary winding. This alternating flux also induces a small alternating voltage in the core material. These voltages produce currents called eddy currents, which are proportional to the voltage. The magnitude of these eddy currents is also limited by the resistivity of the material. The alternating flux is proportional to the applied voltage. Doubling the applied voltage will double the eddy currents. This will raise the core loss by a factor of four. Eddy currents not only flow in the lamination itself, but could flow within the core as a unit, if the lamination is not properly stamped, and if the lamination is not adequately insulated, as shown in Figure 3-6. Copyright 2002 by Marcel Dekker, Inc. All Rights Reserved.

Magnetic Core Applied Voltage

Secondary Voltage

Flux, (() Figure 3-5. Applied Alternating Voltage Induces an Alternating Flux. There are two eddy currents, as shown in Figure 3-6, Ia and It,. The intralaminar eddy current, Ia, is governed by flux, per lamination and resistance of the lamination. It is, therefore, dependent on lamination width, thickness, and volume resistivity. Insulation, (Coating)

Figure 3-6. Using Insulation Between Laminations to Reduce Eddy Currents. The interlaminar eddy current, Ib, is governed by total flux and resistance of the core stack. It is primarily dependent upon stack width and height, the number of laminations, and the surface insulation resistance, per lamination. The magnetic materials used for tape cores and laminations are coated with an insulating material. The insulating coating is applied to reduce eddy currents. The American Iron and Steel Institute (AISI) have set up insulation standards for transformer steels used in different applications. High permeability nickel-iron cores are very strain sensitive. Manufacturers of these cores normally have their own proprietary, insulating material. Copyright 2002 by Marcel Dekker, Inc. All Rights Reserved.

Laminations Laminations are available in scores of different shapes and sizes. The punch press technology for fabricating laminations has been well developed. Most lamination sizes have been around forever. The most commonly used laminations are the El, EE, FF, UI, LL, and the DU as shown in Figure 3-7. The laminations differ from each other by the location of the cut in the magnetic path length. This cut introduces an air gap, which results in the loss of permeability. To minimize the resulting air gap, the laminations are generally stacked in such a way the air gaps in each layer are staggered.

El, Laminations

EE, Laminations

FF, Laminations

UI, Laminations

LL, Laminations

DU, Laminations

Figure 3-7. Commonly Used Lamination Shapes. There are bobbins and brackets for almost all standard stacking dimensions. Most of the El lamination is the scrapless. The name, scrapless, is derived from shapes that are punched with minimum waste, as shown in Figure 3-8.

A

El, Laminations

E, Laminations

A

I, Laminations

Figure 3-8. Typical, Scrapless El Lamination. Copyright 2002 by Marcel Dekker, Inc. All Rights Reserved.

Annealing and Stress-Relief One of the most important parameters in transformer steels is permeability. Any stress or strain of the magnetic materials will have an impact on the permeability. The resulting stress could cause higher magnetizing current, or a lower inductance. When the transformer is being assembled (in the stacking process) and a lamination is bent, (does not return to its original shape), that lamination has been stressed and should be replaced. Some of the important magnetic properties are due to stress and strain after stamping, shearing and slitting. These properties that have been lost or seriously reduced, can be restored to the magnetic materials by annealing. Basically, stress relief is accomplished by heating (annealing) the magnetic material to prescribed temperature, (depending on the material), followed by cooling to room temperature. The entire annealing process is a delicate operation. The annealing must be done under controlled conditions of time, temperature and the ambient atmosphere that will avoid, even minute, adverse changes in the chemistry of the steel.

Stacking Laminations and Polarity The edges of the magnetic material that have been stamped, sheared, or slit, will have a burr, as shown in Figure 3-9. The quality of the equipment will keep the burr to a minimum. This burr now gives the lamination a polarity. When a transformer is being stacked, the lamination build is normally sized by dimensions, or it just fills the bobbin. Lamination Worn Die

Expanded View

Bun-

=^

Figure 3-9. Expanded View, Showing Lamination Burr. If the laminations are stacked correctly, all of the burred ends will be aligned. If the laminations are stacked randomly, such as the burr ends facing each other, then, the stacking factor would be affected. The stacking factor has a direct impact on the cross-section of the core. The end result would be less iron. This could lead to premature saturation, as increase in the magnetizing current, or a loss of inductance. There are several methods used in stacking transformer laminations. The most common technique used in stacking laminations is the alternate method. The alternate method is where one set of laminations, such as an E and an I, are assembled. Then the laminations are reversed, as shown in Figure 3-10. This technique, used in stacking, provides the lowest air gap and the highest permeability. Another method for stacking Copyright 2002 by Marcel Dekker, Inc. All Rights Reserved.

laminations is to interleave two by two also shown in Figure 3-10. The second method of stacking would be in groups of two or more. This is done to cut assembly time. The loss in performance in stacking, other than one by one, is the increase in magnetizing current and a loss of permeability.

Laminations E and I

Interleave 1 x 1

Interleave 2 x 2

Figure 3-10. Methods for Stacking Laminations. Flux Crowding When laminations are stacked, as shown in Figure 3-11, there is flux crowding. This flux crowding is caused by the difference in spacing between the E, I, and the adjacent lamination. The adjacent lamination has a minimum air gap, which translates into a higher permeability.

Laminations E and I

Flux Crowding

)

\ \

Minute Air Gap Flux

Interleave 1 x 1 Figure 3-11. Flux Crowding when Lamination are Interleaved.


Exciting Current The flux will skirt the low permeability, air gap and migrate into the adjacent lamination, causing flux crowding in that lamination. Eventually, this crowding will cause saturation in that portion of the lamination, and the excitation current will rise. After that portion of the lamination has saturated, the flux will migrate back to the lower permeability segment of the lamination from, where it left. This effect can be easily viewed by observing the B-H loops at low and high flux densities and comparing them with a toroidal core of the same material, with a minimum air gap, as shown in Figure 3-12. The B-H loop along with the magnetizing current Im of a toroidal core, is shown in Figure 3-12A. The toroidal core, with its inherit minimum air gap, will have almost a square of current. Using the same material in lamination form will exhibit a B-H loop, and a magnetizing current, Im, similar to Figure 3-12B operating at low flux densities. Increasing the excitation will cause premature saturation of the lamination, as seen by the nonlinear, exciting current as shown in Figure 3-12C

B

rn AB H A

Figure 3-12. Comparing the Exciting Currents and Three B-H Loops. Copyright 2002 by Marcel Dekker, Inc. All Rights Reserved.

Most finished transformers or inductors will have some sort of bracket, such as an L bracket, end bells, a channel bracket or maybe a bolt through the mounting holes to the chassis. When transformers are being assembled, there is a certain amount of attention that has to be used to get proper performance. The insulation material used to coat the lamination is normally very durable, but it can be scratched off and degrade the performance. When brackets are used in the transformer assembly, as shown in Figure 3-13 care must be taken on how the bolts and brackets are put together. The transformer assembly bolts, shown in Figure 3-13 should be the recommended size for the mounting hole and use all of the required hardware. This hardware should include the correct bolt size and length, and correct surface washer, lock washer and nut. Also, included in this hardware, should be fiber shoulder washers and proper sleeving to cover the bolt threads. If insulating hardware is not used, there is a good chance of a partial, shorted turn. The continuity for this partial turn can be created through the bolts and bracket, or the bolts, bracket, and the chassis. This partial shorted turn will downgrade the performance of the transformer.

Sleeving

Laminations Shoulder Washer

Bolt Air Gap Material

Fringing Flux Mounting Bracket

Mounting Bracket Flux

Butt Stack

Figure 3-13. Lamination Mounting Hardware.


Tape Wound C, EE, and Toroidal Cores Tape wound cores are constructed by winding around a mandrel, a magnetic material in the form of a preslit tape, as shown in Figure 3-14. This tape material comes in all of the iron alloys, plus the amorphous materials. The tape thickness varies from 0.0005 inch (0.0127 mm) to 0.012 inch (0.305 mm). The advantage of this type of construction is that the flux is parallel with the direction of rolling of the magnetic material. This provides the maximum utilization of flux with the minimum of magnetizing force. There are two disadvantages in this type of construction. When the core is cut in half, as shown in Figure 3-15, the mating surface has to be ground, lapped, and then, acid-etched. This is done to provide a smooth mating surface with the minimum of air gap and the maximum of permeability. The other disadvantage is when the cores are reassembled, the method used is normally done with a band and buckle, and this procedure requires a little skill to provide the right alignment and correct tension, as shown in Figure 3-16. The C cores are impregnated for strength, prior to being cut. The cut C core can be used in many configurations in the design of a magnetic component, as shown in Figure 3-17. The EE cores are constructed in the same way as C cores, but they have an additional overwind, as shown in Figure 3-18. The assembled, three phase transformer is shown in Figure 3-19.

Magnetic Material (Tape)

Magnetic Material (Tape)

C Core Construction

Toroidal Core Construction

Mandrel

Mandrel

Figure 3-14. Tape Cores Being Wound on a Mandrel.

Cut C Core Mating Surface

Figure 3-15. Two Halves of a Cut C Core. Copyright 2002 by Marcel Dekker, Inc. All Rights Reserved.

Banding Material

Figure 3-16. Banding the Cut C Core.

Single Core Single Coil

(

Core

^ Coil

V

J

Single Core Dual Coils

r GO Coil

^

re

Dual Cores Single Coil

(

Core

^

\

Core

Coil

Coil

j

\[

V

A

}

Figure 3-17. Three Different C Core Configurations.

Overwind C Core Coil

Coil

Coil

WindowCore

Figure 3-18. Three Phase Cut EE Core.


J

Figure 3-19. Typical, Assembled EE Cut Core.

Tape Toroidal Cores Tape toroidal cores are constructed in the same way as tape C cores, by winding the magnetic material around a mandrel, in the form of a preslit tape. This tape material comes in all of the iron alloys, plus the amorphous materials. The tape thickness varies from 0.000125 inch (0.00318 mm) to 0.012 inch (0.305 mm). The tape toroid is normally offered in two configurations, cased and encapsulated as shown in Figure 3-20. The cased toroid offers superior electrical properties and stress protection against winding. The encapsulated cores are used when not all of the fine magnetic properties are important to the design, such as in power transformers.

Enclosure Cased Toroid

Caseless Toroid

Figure 3-20. Outline of a Cased and a Caseless Toroidal Core.

Toroidal, Powder Core Powder cores as shown in Figure 3-21 are very unique. They give the engineer another tool that speed the initial design. Powder cores have a built-in air gap. They come in a variety of materials and are very stable with time and temperature. The cores are manufactured with good engineering aids. Manufacturers provide catalogs for their cores that list, not only the size, but also permeability and Millihenrys per 1000 turns. The data is presented to the engineer in such a way that it takes the minimum amount of time to have a design that will function.

OD

Figure 3-21. Outline of a Powder Toroidal Core. Copyright 2002 by Marcel Dekker, Inc. All Rights Reserved.

Dimensional Outline for El Laminations Laminations are still one of the most widely-used cores in power conversion. The dimensional outline for El laminations is shown in Figure 3-22. The assembled transformer is shown in Figure 3-23. A listing of common El lamination sizes is shown in Table 3-1. E, Laminations

I, Laminations

w. G

D

Figure 3-22. El Lamination Outline.

V

Lamination

-

1

/ Channel Bracket

Coil Mounting Foot Dt *

Figure 3-23. El Lamination Assembled with Channel Bracket. Table 3-1. Standard 14 mil El Laminations.

El, Laminations Part Number EI-375 EI-021 EI-625 El-750 EI-875 EI-100 EI-112 EI-125 EI-138 EI-150 EI-175 EI-225

D cm 0.953 1.270 1.588 1.905 2.223 2.540 2.857 3.175 3.493 3.810 4.445 5.715

E cm 0.953 1.270 1.588 1.905 2.223 2.540 2.857 3.175 3.493 3.810 4.445 5.715

F cm 0.794 0.794 0.794 0.953 1.111 1.270 1.429 1.588 1.746 1.905 2.223 2.858


G cm 1.905 2.064 2.381 2.857 3.333 3.810 4.286 4.763 5.239 5.715 6.668 8.573

wa Ac

1.754 1.075 0.418 0.790 0.789 0.790 0.789 0.789 0.789 0.789 0.789 0.789

Ac

cm" 0.862

1.523 2.394 3.448 4.693 6.129 7.757 9.577 11.588 13.790 18.770 31.028

wa 2

cm 1.512 1.638 1.890 2.723 3.705 4.839 6.124 7.560 9.148 10.887 14.818 24.496

A

P 4

cm 1.303 2.510 4.525 9.384 17.384 29.656 47.504 72.404 106.006 150.136 278.145 760.064

K* cm 0.067 0.188 0.459 1.153 2.513 4.927 8.920 15.162 24.492 37.579 81.656 288.936

Dimensional Outline for UI Laminations The dimensional outline for UI laminations is shown in Figure 3-24. The assembled transformer is shown in Figure 3-25. A listing of common UI lamination sizes is shown in Table 3-2.

o

0

i ir i

•^*~

H

G

o F.

0

i ii i,H

D

F

F

Figure 3-24. UI Lamination Outline. Mounting Hardware Bolt, Washer, Nut Sleeving Shoulder Washer

0 UI, Laminations

Coil#l

Coil#2

e Side View

End View

Figure 3-25. UI Lamination Assembled with Coils and Hardware.

Table 3-2. Standard 14 mil UI Laminations.

Part

D

E

F

UI, Standard Laminations G H wa A c

2

wa

AP 4 cm 7.414

Kg 5 cm 0.592

Number

cm

cm

cm

cm

cm

Ac

50UI

1.270 1.429

1.270

3.810

1.270

60UI

1.270 1.429

2.223

5.398

1.429

3.159 6.187

1.939

cm 4.839 11.996

23.263

1.839

75UI

1.905

1.905

1.905

5.715

3.157

3.448

10.887

37.534

4.614

100UI

2.540

2.540

2.540

7.620

1.905 2.540

3.158

6.129

19.355

118.626

19.709

125UI

3.175

3.175

3.175

9.525

3.175

3.158

9.577

30.242

289.614

60.647

15 GUI 180UI

3.810 4.572

3.810 4.572

3.810 4.572

11.430

3.158 2.632

13.790

43.548

600.544

150.318

11.430

3.810 4.572

19.858

52.258

1037.740

313.636

240UI

6.096

6.096

6.096

15.240

6.096

2.632

35.303

92.903

3279.770

1331.997


cm 1.532

Dimensional Outline for LL Laminations The dimensional outline for LL laminations is shown in Figure 3-26. The assembled transformer is shown in Figure 3-27. A listing of common lamination sizes is shown in Table 3-3.

o

o I

1r H

It ^

G

o

o

-^ ^-

-^ ^-

-^ ^-

c

K

c

\r i 1! H D

Figure 3-26. LL Lamination Outline.

LL, Laminations

e -^>

e

Coil#l

e

Side View Figure 3-27.

Mounting Hardware Bolt, Washer, Nut Sleeving Shoulder Washer

Coil

Coil#2

e

End View

4 ^ LL Lamination Assembled with Coils and Hardware.

Table 3-3. Standard 14 mil LL Laminations.

LL, Standard Laminations Number

D cm

E cm

F cm

G cm

141L

0.635

0.635

2.858

108L 250L

1.031

1.031 1.031

1.270 0.874 0.874

5.239

101L 7L

1.111 1.270

1.111

1.588

2.858

1.270

1.270

3.810

4L 104L

1.270

1.270 1.270 1.270

3.810 5.555

105L

1.270 1.270

1.905 1.984

102L 106L

1.429 1.429

107L

1.588

Part

1.031

3.334

H cm 0.635

1.111 1.111

wa

Ac

Kg cm 0.043 0.201 0.316

9.473

cm 0.383

cm 3.629

2.884

1.010

4.532

1.010

2.913 4.577

AP 4 cm 1.390 2.943 4.624

3.867

4.536

5.322

0.340

4.839

7.414

0.592

Ac

7

7

1.111 1.270

3.159

1.173 1.532

1.270

4.737

1.532

7.258

11.121

0.785

1.270 1.270

7.193 8.488

1.532

11.020

16.885

1.176

1.532

13.004

19.925

4.419 6.187

1.939 1.939

8.569

16.617

1.407 1.462

11.996

23.263

1.839

5.474

2.394

13.105

31.375

2.946

1.905

6.826

1.429 1.429

1.588 2.223

5.398 5.398

1.429 1.429

1.588

2.064

6.350

1.588


wa

Dimensional Outline for DU Laminations The dimensional outline for DU laminations is shown in Figure 3-28. The assembled transformer is shown in Figure 3-29. A listing of common DU laminations sizes is shown in Table 3-4.

ii H

0

0

o

o

o

o

0

0

\r i

i

G ir t H

k

ir

-^ E

F

E -«-

Figure 3-28. DU Lamination Outline.

DU, Laminations

e — +*

Coil #1 Coil #2

e

Side View

is

c

e

Mounting Hardware Bolt, Washer, Nut Sleeving Shoulder Washer

Coil

e

fc

C

End View

Figure 3-29. DU Lamination Assembled with Coils and Hardware. Table 3-4. Standard 14 mil DU Laminations.

DU, Standard Laminations Part Number DU-63 DU-124 DU-18 DU-26 DU-25 DU-1 DU-39 DU-37 DU-50 DU-75 DU-1 125 DU-125

D cm

E cm

G cm

H cm

wa

cm

0.159 0.318 0.476 0.635 0.635 0.635 0.953 0.953 1.270 1.905 2.858 3.175

0.159 0.318 0.476 0.635 0.635 0.635 0.953 0.953 1.270 1.905 2.858 3.175

0.318 0.476 0.635 0.635 0.953 0.953 0.953 1.905 2.540 3.810 5.715 3.175

0.794 1.191 1.588 1.905 2.064 3.810 2.858 3.810 5.080 7.620 11.430 9.525

0.318 0.635 0.953 1.270 1.270 1.270 1.905 1.905 2.540 3.810 5.715 3.175

10.500 5.906 4.688 3.159 5.133 9.634 3.158 8.420 8.422 8.420 8.421 3.158

F


Ac

Ac

cm2 0.024 0.096 0.215 0.383 0.383 0.383 0.862 0.862 1.532 3.448 7.757 9.577

wa cm 0.252 0.567 1.008 1.210 1.966 3.690 2.722 7.258 12.903 29.032 65.322 30.242

Ap 4

cm 0.006 0.054 0.217 0.463 0.753 1.390 2.346 6.256 19.771 100.091 506.709 289.614

Kg cm5 0.00003 0.0009 0.0057 0.0180 0.0260 0.0479 0.1416 0.2992 1.2524 9.7136 74.8302 60.6474

Dimensional Outline for Three Phase Laminations The dimensional outline for three phase laminations is shown in Figure 3-30. The assembled transformer is shown in Figure 3-31. A listing of common three phase laminations sizes is shown in Table 3-5. 1t

E

*r i

F

0

i

O

^r

O v v """ W a

Ar

0

G

D

E

o

O

Figure 3-30. El Three Phase Laminations Outline.

Laminations -

Side Vie w

e —^

e

e

Coil #1

Coil #2

©

©

Coil #3

©

A

B

E

1r

\ ir

V\

r

^ 'W,

\

-* H -»

AcP

(

Figure 3-41. Dimension Outline for EFD Ferrite Cores. Table 3-12. Standard EFD Ferrite Cores.

EFD, Ferrite Cores Part Number EFD-10 EFD-15 EFD-20 EFD-25 EFD-30

A

B

C

D

E

G

H

wa

cm

cm

cm

cm

cm

cm

Ac

1.050

0.765

0.145

1.610

1.100 1.540

1.500 2.000

0.455 0.530 0.890

0.750

1.500 2.000 2.500 3.000

0.270 0.465 0.665

cm 1.040

1.100 1.540

0.240 0.360

2.090 1.610

1.870

0.910 0.910

1.140 1.460

1.860 2.240

0.520 0.490

1.170

2.240

2.500 3.000


1.266

^A c cm"

wa cm"

0.072 0.150 0.310

0.116

0.580 0.690

0.679 0.874

0.314 0.501

A

P

Kg

4

cm cm" 0.00836 0.00013

0.04702 0.00105 0.15515 0.00506 0.39376 0.01911 0.60278 0.03047

Dimensional Outline for EPC, Ferrite Cores SMD The dimensional outline for EPC ferrite cores is shown in Figure 3-42. A listing of ECP cores is shown in Table 3-13.

C

D

A

B

Ar

H

G

Figure 3-42. Dimension Outline for EPC Ferrite Cores. Table 3-13. Standard EPC Ferrite Cores.

EPC, Ferrite Cores (TDK) Part

A

B

C

D

E

G

H

wa

Number

cm

cm

cm

cm

cm

cm

cm

EPC- 10

1.020

0.760

0.340

0.810

0.500

0.530

0.190

Ac 0.734

cm" cm 0.0939 0.0689

cm 0.006470

EPC-13

1.325

1.050

0.460

1.320

0.560

0.900

0.205

1.765

0.1250 0.2205

0.027562

0.000549

EPC- 17

1.760

1.430

0.600

1.710

0.770

1.210

0.280

1.751

0.2280 0.3993

0.091040

0.002428

Ac 1

wa

Ap

Kg cm 0.000128

EPC- 19

1.910

1.580

0.600

1.950

0.850

1.450

0.250

2.334

0.2270 0.5293

0.120140

0.002981

EPC-25

2.510

2.080

0.800

2.500

1.150

1.800

0.400

1.804

0.4640 0.8370

0.388368

0.014533

EPC-27

2.710

2.160

0.800

3.200

1.300

2.400

0.400

1.890

0.5460 1.0320

0.563472

0.024036

EPC-30

3.010

2.360

0.800

3.500

1.500

2.600

0.400

1.832

0.6100 1.1180

0.681980

0.030015


Dimensional Outline for PC, Ferrite Cores The dimensional outline for PC ferrite cores is shown in Figure 3-43. A listing of PC cores is shown in Table 3-14.

C

B

A

X G

Figure 3-43. Dimension Outline for PC Ferrite Cores. Table 3-14. Standard PC Ferrite Cores.

PC, Ferrite Cores (Magnetics) Part

A

B

C

E

G

wn

wa

A

0.650

crrf 0.100

cm 0.065

P cm4 0.00652

K \ cm" 0.000134

0.559

0.631

0.249

0.157

0.03904

0.001331

0.720

0.697

0.429

0.299

0.11413

0.005287

0.940

0.920

0.612

0.639

0.391

0.24985

0.014360

1.610

1.148

1.102

0.576

0.931

0.536

0.49913

0.035114

1.880

1.350

1.300

0.549

1.360

0.747

1.01660

0.088001

2.200

1.610

1.460

0.498

2.020

1.007

2.03495

0.220347

2.960

1.770

2.040

0.686

2.660

1.826

4.85663

0.600289

Number

cm

cm

cm

cm

cm

Ac

PC-40905

0.914

0.749

0.562

0.388

0.361

PC-41408

1.400

1 . 1 60

0.848

0.599

PC-41811

1.800

1.498

1.067

0.759

PC-42213

2.160

1.790

1.340

PC-42616

2.550

2.121

PC-43019

3.000

2.500

PC-43622

3.560

2.990

PC-44229

4.240

3.560


AC

2

Dimensional Outline for EP, Ferrite Cores The EP ferrite cores are typically used in transformer applications. The shape of the assembly is almost cubical, allowing high package densities on the PCB. The dimensional outline for EP ferrite cores is shown in Figure 3-44. A listing of EP cores is shown in Table 3-15.

D

A

B

A

c

Figure 3-44. Dimension Outline for EP Ferrite Cores. Table 3-15. Standard EP Ferrite Cores.

EP, Ferrite Cores Part Number EP-7 EP-10 EP-13 EP-17 EP-20

A cm 0.940 1.150 1.280 1.800 2.400

B cm 0.720 0.940 0.970 1.200 1.650

C cm 0.650 0.760 0.900 1.100 1.500

D cm 0.750 1.020 1.300 1.680 2.140


E cm 0.340 0.330 0.450 0.570 0.880

G cm 0.500 0.740 0.900 1.140 1.440

wa Ac

0.987 1.997 1.344 1.066 0.704

Ac

wa

cm" 0.1080 0.1130 0.1950 0.3370 0.7870

cm 0.1066 0.2257 0.2622 0.3591 0.5544

AP cm4 0.01151 0.02550 0.05112 0.12101 0.43631

KB cm 0.00027 0.00053 0.00165 0.00540 0.03261

Dimensional Outline for PQ, Ferrite Cores The PQ ferrite cores (Power Quality) feature round center legs with rather small cross-sections. The dimensional outline for PQ femte cores is shown in Figure 3-45. A listing of PQ cores is shown in Table

3-16.

Ar

Figure 3-45. Dimension Outline for PQ Ferrite Cores. Table 3-16. Standard PQ Ferrite Cores.

PQ, Ferrite Cores Part

A

B

D

C

E

G

wa

Number

cm

cm

cm

cm

cm

cm

Ac

PQ20/16 PQ20/20 PQ26/20 PQ26/25 PQ32/20 PQ32/30 PQ35/35 PQ40/40 PQ50/50

2.050 2.050 2.650 2.650 3.200 3.200 3.510 4.050 5.000

1.800 1.800 2.250 2.250 2.750 2.750 3.200 3.700 4.400

1.620 2.020 2.015 2.475 2.055 3.035 3.475 3.975 4.995

1.400 1 .400 1.900 1.900 2.200 2.200 2.600 2.800 3.200

0.880 0.880 1.200 1.200 1.345 1.245 1.435 1 .490 2.000

1.030 1.430 1.150 1.610 1.150 2.130 2.500 2.950 3.610

0.764 1.061 0.507 0.724 0.475 0.930 1.126 1.622 1.321


Ac 2

cm 0.620 0.620 1.190 1.180 1.700 1.610 1 .960 2.010 3.280

wa cm" 0.474 0.658 0.604 0.854 0.808 1.496 2.206 3.260 4.332

AP 4 cm4 0.294 0.408 0.717 0.997 1.373 2.409 4.324 6.552 14.210

Kg cm 0.0167 0.0232 0.0161 0.0855 0.1401 0.2327 0.4511 0.6281 1.8123

Dimensional Outline for PQ/(low profile), Ferrite Cores The PQ/lp cores are a cut down version of the standard PQ cores. The PQ/lp cores have a substantially reduced total height. The dimensional outline for PQ/lp ferrite cores is shown in Figure 3-46. A listing of PQ/lp cores is shown in Table 3-17.

G

r— j i

~\ 1L

^\

\\

B

\

1r

A \ \

\r -^ (—i

Figure 3-46. Dimension Outline for PQ Ferrite Cores.

Table 3-17. Standard PQ Ferrite Cores.

PQ/lp, Ferrite Cores (Ferrite International) Part Number PQ20/20/lp PQ26/20/lp PQ32/20/lp PQ35/35/lp PQ40/40/lp

A cm 2.125 2.724 3.302 3.612

B cm 1.801

C cm 2.702

D cm 1.400

E cm 0.884

G cm 1.524

2.250 2.751 3.200

1.900 2.200 2.601

1.199

1.524 1.524 1.524

4.148

3.701

3.260 3.342 3.474 3.566

2.799


1.348 1.435 1.491

1.524

wa

Ac

2

Ac 1.127

cm 0.620

0.673 0.629 0.686 0.838

1.190 1.700 1.960 2.010

wa 2

cm 0.699 0.801 1.069 1.345 1.684

A

P cm4 0.433 0.953 1.817 2.636 3.385

K

g cm5 0.024 0.080 0.185 0.275 0.324

Dimensional Outline for RM, Ferrite Cores The RM cores (Rectangular Modular) were developed for high printed circuit board (PCB) packing densities. The dimensional outline for RM ferrite cores is shown in Figure 3-47. A listing of RM cores is shown in Table 3-18.

A

Figure 3-47. Dimension Outline for RM Ferrite Cores.

Table 3-18. Standard RM Ferrite Cores.

RM, Ferrite Cores B cm

C cm

D

E

G

H

wa

Ac

wa

cm

cm

cm

cm

Ac

0.815

1.04

NA

0.38

0.72

NA

1.12

cm' 0.157

1.04 1.265

1.04 1.24

NA

0.48

NA

0.63

0.65 0.82

NA

RM-6

0.963 1.205 1.44

cm" 0.14

0.768 0.71

RM-7

1.685

1.34

NA

0.71

NA

1.64

NA

0.84

1.86

NA

2.35 2.88

NA NA

1.07 1.26 1.47

0.865 1.1 1.27

Part Number RM-4 RM-5

A cm

RM-8

1.935

1.508 1.73

RM-10

2.415

2.165

RM-12 RM-14

2.925 3.42

2.55 2.95


1.71 2.1 1

A

P 4 cm 0.0219

Kg cm" 0.0006

0.0431 0.0953

0.0016 0.0044

0.159

0.00802 0.01911

NA

0.75 0.76

0.237 , 0 . 1 8 2 0.26 0.366 0.46 0.345 0.64 0.489

NA

0.71

0.695

0.681

1.103 1.556

1.544 2.77

NA

NA NA

0.98 0.788 ' 1.4 0.874 1.78

0.313

0.05098 0.139 0.2744

Dimensional Outline for RM/(low profile), Ferrite Cores SMD The RM/lp ferrite cores are a cut down version of the standard RM cores. The dimensional outline for RM/lp ferrite cores is shown in Figure 3-48. A listing of RM/lp cores is shown in Table 3-19.

A

Figure 3-48. Dimension Outline for RM/lp Ferrite Cores.

Table 3-19. Standard RM/lp Ferrite Cores.

RM/lp, Ferrite Cores (Ferrite International) Part Number PQ20/20/!p PQ26/20/!p PQ32/20/lp PQ35/35/lp PQ40/40/lp

A cm 2.126 2.725 3.302 3.612 4.148

B cm 1.801 2.250 2.751 3.200 3.701

C cm 2.702 3.260 3.342 3.474 3.566

D cm 1.400 1.900 2.200 2.601 2.799


E cm 0.884

1.199 1.348 1.435 1.491

G cm 1.524 1.524 1.524 1.524 1.524

wa Ac 1.127 0.673 0.629 0.686 0.838

Ac 7

cm" 0.620 1.190 1.700 1.960 2.010

wa

•> cm" 0.699 0.801 1.069 1.345 1.684

A

Kg P 4 cm5 cm 0.43323 0.02412 0.95303 0.08022 1.81744 0.18514 2.63606 0.27494 3.38488 0.32431

Dimensional Outline for DS, Ferrite Cores The DS ferrite cores are similar to standard Pot Cores. These cores have a large opening to bring out many strands of wire, which is convenient for high power and multiple outputs. The dimensional outline for DS ferrite cores is shown in Figure 3-49. A listing of DS cores is shown in Table 3-20.

Figure 3-49. Dimension Outline for DS Ferrite Cores.

Table 3-20. Standard DS Ferrite Cores.

DS, Ferrite Cores (Magnetics) Part

A

B

C

D

E

G

wa

Ac

wa

Number

cm

cm

cm

cm

cm

cm

AC

DS-42311

2.286

1.793

1.108

1.540

0.990

0.726

0.568

DS-42318

2.286

1.793

1.800

1.540

0.990

1.386

0.960

DS-42316

2.550 3.000

2.121

1.610 1.880

1.709

1.148

1.102

2.500

1.709

1.351

1.300

0.696 0.638

3.561

2.985

2.170

2.385

1.610

1.458

0.672

4.240

3.561

2.960

2.840

1.770

2.042

0.875

cm 0.512 0.580 0.770 1.170 1.490 2.090

cm" 0.291 0.557 0.536 0.747 1.002 1.828

DS-42319 DS-42322 DS-42329


AP 4 cm 0.14920 0.32275 0.41281 0.87381 1.49354 3.82179

Kg cm 0.00674 0.01624 0.02402 0.06506 0.11942 0.37109

Dimensional Outline for UUR, Ferrite Cores The UUR ferrite cores feature round legs with rather small cross sections. The round legs allow easy winding with either wire or foil. U cores are used for power, pulse and high-voltage transformers. The dimensional outline for UUR ferrite cores is shown in Figure 3-50. A listing of UUR cores is shown in Table 3-21.

C

A

Figure 3-50. Dimension Outline for UUR Ferrite Cores. Table 3-21. Standard UUR Ferrite Cores.

UUR, Ferrite Cores (Magnetics) Part Number UUR-44121 UUR-44119 UUR-44125 UUR-44130

A cm 4.196 4.196 4.196 4.196

C cm 4.120 4.180 5.080 6.100

D cm 1.170 1.170 1.170 1.170


F cm 1.910 1.910 1.910 1.910

G cm 2.180 2.680 3.140 4.160

wa Ac

4.215 5.619 6.070 8.O43

AC

wa

cm" 0.988 0.911 0.988 0.988

cm" 4.164 5.119 5.997 7.946

AP cm 4 4.114

4.663 5.925 7.850

Kg cm 0.202 0.211 0.291 0.386

Dimensional Outline for UUS, Ferrite Cores The UUS ferrite cores feature square or rectangular legs. U cores are used for power, pulse and highvoltage transformers. The dimensional outline for UUS ferrite cores is shown in Figure 3-51. A listing of UUS cores is shown in Table 3-22.

A

D

G

Figure 3-51. Dimension Outline for UUS Ferrite Cores.

Table 3-22. Standard UUS Ferrite Cores.

UUS, Ferrite Cores (Philips) Part Number UUS-10 UUS-20 UUS-25 UUS-30 UUS-67 UUS-93

F

G

wa

AC

wa

cm

D cm

cm

cm

Ac

1.640 3.120 3.920

0.290 0.750 1.270

0.435 0.640 0.840

1.000

5.179

1.660

1.896 1.841

cm" 0.084 0.560

cm" 0.435 1.062

5.060 5.400 15.200

1.600 1.430

1.050

A

C

cm 1 .000

2.080 2.480 3.130

6.730 9.300

4.800

3.880 3.620


2.280 2.980 2.540 9.600

1 .040

1.915

4.831

1.610 2.040

3.129 9.855

7.757

4.480

34.752

1.943

A

P 4 cm 0.0365

Kg 5 cm 0.000549

0.5949 1.9918 5.0377 20.1046 155.6889

0.030612 0.135668 0.430427 1.321661 12.808331

Dimensional Outline for Toroidal, Ferrite Cores The toroidal ferrite core has the best possible shape from the magnetic point of view. The magnetic flux path is completely enclosed within the magnetic structure. The toroidal structure fully exploits the capabilities of a ferrite material. The dimensional outline for toroidal ferrite cores is shown in Figure 3-52. A listing of toroidal cores is shown in Table 3-23.

HT.

I.D

Figure 3-52. Dimension Outline for Toroidal Ferrite Cores.

Table 3-23. Standard Toroidal Ferrite Cores.

Toroidal, Ferrite Cores (Magnetics) Part Number TC-40705 TC-41206 TC-42206 TC-42908 TC-43806 TC-43610 TC-43813 TC-48613

OD cm 0.762 1.270 2.210 2.900

3.810 3.600 3.810 8.570

ID cm 0.318 0.516

1.370 1.900 1.900 2.300

1.900 5.550

HT cm 0.478 0.635 0.635 0.749 0.635 1.000 1.270 1.270


wa

wa

Ac 0.806 0.946 5.896 7.919 4.974

cm 0.098

cm" 0.079

0.221

0.209

AP cm4 0.007783 0.046215

0.250

1.474

0.368528

0.358

2.835

1.015032

Kg cm5 0.000222 0.002011 0.013237 0.041312

0.570

2.835

1.616112

0.090668

6.616

0.628

0.146301

1.150 1.870

4.155 2.835 24.192

2.609185

2.465

3.260577

0.295249

45.239426

3.807278

12.937

Ac

2

7

Dimensional Outline for Toroidal, MPP Powder Cores The dimensional outline for toroidal MPP powder cores is shown in Figure 3-53. A listing of toroidal cores is shown in Table 3-24.

OD

Figure 3-53. Dimension Outline for Toroidal Powder Cores.

Table 3-24. A Small List of Standard Toroidal MPP Powder Cores.

MPP Powder Cores, Magnetics 60 mu (coated) ID

HT

MPL

wa

Ac

wa

Number

OD cm

cm

cm

cm

Ac

cm"

cm"

55021

0.699

0.229

0.343

1.36

0.877

0.047

0.041

55281

1.029

0.427

0.381

2.18

1.900

0.075

0.143

55291

1.029

0.427

0.457

2.18

1.512

0.095

0.143

55041

1.080

0.457

0.457

2.38

1.640

0.100

0.164

55131

1.190

0.589

0.472

2.69

3.013

0.091

0.273

55051

1.346

0.699

0.551

3.12

3.360

0.114

0.383

55121

1.740

0.953

0.711

4.11

3.714

0.192

0.713

55381

1.803

0.902

0.711

4.14

2.750

0.232

0.638

55848

2.110

1.207

0.711

5.09

5.044

0.226

1.140

55059

2.350

1.339

0.838

5.67

4.260

0.331

1.410

55351

2.430

1.377

0.970

5.88

3.840

0.388

1.490

55894

2.770

1.410

1 . 1 99

6.35

2.385

0.654

1.560

55071

3.380

1.930

1.161

8.15

4.360

0.672

2.930

55586

3.520

2.260

0.983

8.95

8.833

0.454

4.010

55076

3.670

2.150

1.128

8.98

5.369

0.678

3.640

55083

4.070

2.330

1.537

9.84

3.983

1.072

4.270

55439

4.760

2.330

1.892

10.74

2.146

1.990

4.270

55090

4.760

2.790

1.613

11.63

4.560

1.340

6.110

55716

5.170

3.090

1.435

12.73

5.995

1.251

7.500

55110

5.800

3.470

1 .486

14.300

6.565

1.444

9.480

Part


A

P 4 cm 0.001930 0.010746 0.013504

0.016400 0.024734 0.043662 0.136896 0.148016 0.257640 0.466710 0.578120 1 .020240 1.968960 1.820540 2.467920 4.577440 8.497300 8.187400 9.382500 13.689120

Kg cm 0.000040 0.000252

0.000357 0.000432 0.000603 0.001225 0.004787 0.006030 0.009730 0.020153 0.028639 0.070381 0.124886 0.087356 0.154082 0.389064 1.034038 0.717932 0.762526 1.212742

AL mh/lOOON 24 25 32 32 26 27 35 43 32 43 51 75 61 38 56 81 135 86 73 75

Dimensional Outline for Toroidal, Iron Powder Cores The dimensional outline for toroidal iron powder cores is shown in Figure 3-54. A listing of toroidal cores is shown in Table 3-25.

OD

Figure 3-54. Dimension Outline for Toroidal Iron Powder Cores.

Table 3-25. A Small List of Standard Toroidal Iron Powder Cores.

Iron Powder Cores, Micrometals 75 mu (coated) OD Part Number cm T20-26 0.508 T25-26 0.648 T26-26 0.673 T30-26 0.780 T37-26 0.953 T38-26 0.953 T44-26 1.120 T50-26 1.270 1.520 T60-26 1.750 T68-26 2.020 T80-26 T94-26 2.390 T90-26 2.290 T 1 06-26 2.690 T 130-26 3.300 T132-26 3.300 T131-26 3.300 T141-26 3.590 T 150-26 3.840 Tl 75-26 ^4.450

ID cm 0.224

0.305 0.267 0.384 0.521 0.445 0.582 0.770 0.853 0.940 1.260 1.420 1.400 1.450 1.980 1.780 1.630 2.240 2.150 2.720

HT cm

MPL cm

wa

0.178 0.244 0.483 0.325 0.325 0.483 0.404 0.483 0.594

1.15 1.50 1.47 1.84 2.31 2.18 2.68 3.19 3.74

1.713 1.974 0.622

0.483 0.635 0.792 0.953 1.110 1.110 1.110 1.110 1.050 1.110 1.650

4.23 5.14

3.875 5.395 4.373 3.895 2.504 4.409 3.090 2.357 5.844 4.091 4.334

5.97 5.78 6.49 8.28 7.96 7.72 9.14 9.38 11.200


Ac

1.929 3.329 1.364 2.686 4.156 3.054

Ac

cm 0.023 0.037 0.090 0.060 0.064 0.114 0.099 0.112 0.187 0.179 0.231 0.362 0.395 0.659 0.698 0.805 0.885 0.674 0.887 1.340

wa 2

cm 0.039 0.073 0.056 0.116 0.213 0.155 0.266 0.465 0.571 0.694 1.246 1.583 1.539 1.650 3.078 2.487 2.086 3.939 3.629 5.808

A

P 4 cm 0.000906 0.002702

0.005037 0.006945 0.013637 0.017721 0.026324 0.052128 0.106809 0.124159 0.287887 0.573000 0.607747 1.087655 2.148105 2.002191 1.845815 2.654762 3.218624 7.782377

Kg cm5 0.000014

0.000053 0.000154 0.000158 0.000308 0.000572 0.000760 0.002174 0.003938 0.007187 0.010389 0.016164 0.030827 0.113914 0.141056 0.151249 0.152983 0.164023 0.246891 0.715820

AL mh/lOOON 18.5 24.5 57 33.5 28.5 49 37 33 50 43.5 46 60 70 93 81 103 116 75 96 105

Chapter 4

Window Utilization, Magnet Wire, and

Insulation


Table of Contents

1.

Window Utilization Factor, K u

2.

Sj, Wire Insulation

3.

S2, Fill Factor

4.

S3, Effective Window

5.

S4, Insulation Factor

6.

Circular mil and Square mil

7.

Summary

8.

Magnet Wire

9.

Magnet Wire, Film Insulation

10. Wire Table 11. Solderable Insulation 12. Bondable Magnet Wire 13. Base Film Insulation 14. Bonding Methods 15. Miniature Square Magnet Wire 16. Multistrand Wire and Skin Effe 17. Multistrand Litz Wire 18. Specialty Wire 19. Triple Insulated Wire 20. Triple Insulated Litz 21. Polyfilar Magnetic Wire 22. Standard Foils 23. The Use of Foils 24. Calculating, MLT 25. Calculating, MLT (toroid) 26. Copper Resistance 27. Copper Weigh 28. Electrical Insulating Materials 29. References


Window Utilization Factor, Ku

The window utilization factor is the amount of copper that appears in the window area of the transformer or inductor. The window utilization factor is influenced by five main factors:

1.

Wire insulation, S|.

2.

Wire lay fill factor, layer or random wound, S2.

3.

Effective window area (or, when using a torrid, the clearance hole for passage of the shuttle), S3.

4.

Insulation required for multiplayer windings, or between windings, S4.

5.

Workmanship, (quality).

These factors, multiplied together, will give a normalized window utilization of Ku = 0.4, as shown in Figure 4-1.

Core Window Area Area Taken By: Bobbin Tube Margin Wrapper Insulation Layer Insulation Magnet Wire Insulation Fill Factor

Copper Area

Figure 4-1. Window Area Occupied by Copper.

The window utilization factor, K u , of the available core window space, that will be occupied by the winding, (copper), is calculated from areas, S,, S2, S3, and S4:

J^jj — O| X 02 X 03 X 04

Where: 51 - conductor area/wire area 52 = wound area/usable window area 53 = usable window area/window area 54 = usable window area/usable window area + insulation Copyright 2002 by Marcel Dekker, Inc. All Rights Reserved.

In which: Conductor area, A W ( B ) = copper area. Wire area, Avv = copper area + insulation area. Wound area = number of turns x wire area of one turn. Usable window area - available window area - residual area, that results from the particular winding technique used. Window area = available window area. Insulation area = area used for winding insulation.

Si, Wire Insulation In the design of high-current or low-current transformers, the ratio of the conductor area to the total wire area can vary from 0.941 to 0.673, depending on the wire size. In Figure 4-2, the thickness of the insulation has been exaggerated to show how the insulation impacts the overall area of the wire.

It can be seen, in Figure 4-2, that, by using multi-strands of fine wire to reduce the skin effect, it will have a significant impact on the window utilization factor, K u . Si is not only dependent upon wire size, but it is also dependent upon insulation coating. Table 4-1 shows the ratio of bare magnet wire to the magnet wire with insulation for single, heavy, triple, and quad insulation. When designing low-current transformers, it is advisable to re-evaluate, Si, because of the increased amount of insulating material. Si = A W( B/A W

Insulation 0.00965 cm

0.00787 cm

AWG #40

0.268 cm

0.259 cm

AWG #10

Figure 4-2. Comparing Insulation with Different Wire Gauges.


Table 4-1

Magnetic Wire Data (Nominal) Size AWG 10 15 20 25 30 35 40

Bare Area (cm") 0.1019 0.0571 0.0320 0.0179 0.0100 0.0056 0.0031

Ratio Bare/Single 0.961 0.939 0.917 0.878 0.842 0.815 0.784

Ratio Bare/Heavy 0.930 0.899 0.855 0.793 0.743 0.698 0.665

Ratio Bare/Triple 0.910 0.867 0.812 0.733 0.661 0.588 0.544

Ratio Bare/Quad 0.880 0.826 0.756 0.662 0.574 0.502 0.474

S2, Fill Factor S2 is the fill factor, or the wire lay, for the usable window area. When winding a large number of turns tightly on a smooth surface, the winding length exceeds the calculated value from the wire diameter by 10 to 15%, depending on the wire gauge. See Figure 4-3. The wire lay is subjected to wire tension, and wire quality, such as continuous wire diameter and the winding technique depending on the skill of the operator. The wire lay factor relationship for various wire sizes is shown in Table 4-2, for layer wound coils, and in Table 4-3, for random wound coils. The tables list the outside diameter for heavy film magnetic wire, 10 -

44 AWG.

Table 4-2

Wire Lay Factor For Layer Wound Coils Insulated Wire OD (inch) 0.1051 -0.0199 10 to 25 0.0178-0.0116 26 to 30 0.0105-0.0067 31 to 35 0.0060 - 0.0049 36 to 38 39 to 40 0.0043 - 0.0038 41 to 44 0.0034 - 0.0025 Heavy film magnetic wire

AWG

Insulated Wire OD (cm) 0.2670-0.0505 0.0452 - 0.0294 0.0267-0.0170 0.0152-0.0124 0.0109-0.0096 0.00863 - 0.00635

Wire Lay Factor 0.90 0.89 0.88 0.87 0.86 0.85

Table 4-3

Wire Lay Factor For Random Wound Coils Insulated Wire OD (inch) 10 to 22 0.1051 -0.0276 23 to 39 0.0623-0.0109 40 to 44 0.0038 - 0.0025 Heavy film magnet wire.

AWG


Insulated Wire OD (cm) 0.267 - 0.0701 0.0249 - 0.0043 0.0096 - 0.00635

Wire Lay Factor 0.90 0.85 0.75

Calculated turns Actual turns

ooooooooooooo Winding Length

Figure 4-3. Capable Turns per Unit Length.

There are two ideal winding arrangements shown in Figure 4-4 and Figure 4-5. The square winding is shown in Figure 4-4 and the hexagonal winding is shown in Figure 4-5. The simplest form of winding is done by a coil being wound, turn-by-turn and layer-upon-layer, as shown in Figure 4-4. The square winding pattern has a theoretical fill factor of 0.785.

Wire Area = 0.785

Winding Build

Figure 4-4. Theoretically, the Square Winding Pattern Fill Factor 0.785.

A seemingly better fill factor can be achieved by using the hexagonal winding in Figure 4-5, compared to the square winding in Figure 4-4. In this type of winding, the individual wires do not lie exactly above each other, as in the square winding pattern. Instead, the wires lie in the grooves of the lower layer, as shown in Figure 4-5. This style of winding produces the tightest possible packing of the wire. The hexagonal style of winding will yield a theoretical fill factor of 0.907.

The fill factor, using the square winding pattern of 0.785, would be nearly impossible to achieve by hand winding without some layer insulation. Any layer insulation will reduce the fill factor even further. The fill factor, using the hexagonal winding pattern of 0.907, is just as hard to get. Hand winding, using the hexagonal technique, will result in the following: The first layer goes down with almost complete order. In


0.866(D)

Winding Build

Figure 4-5. Theoretically, the Hexagonal Winding Pattern Fill Factor 0.907.

the second layer, some disordering has occurred. With the third and fourth layer, disordering really sets in and the winding goes completely awry. This type of winding performs well with a small number of turns, but, with a large number of turns, it becomes randomly wound.

The ideal winding on a rectangular bobbin is shown in Figure 4-6. Then, when winding rectangular bobbins or tubes, the actual winding height in the region covered by the core, will be greater than the calculated winding height or build, due to the bowing of the windings. See Figure 4-7. The amount of bowing depends on the proportions of the winding and the height of the winding. Usually, the available winding build should be reduced by 15 to 20%, or 0.85x the winding build. When winding on a round bobbin or tube, this bowing effect is negligible.

The conclusion is, in comparing the square winding pattern used in the layer wound coil with its insulation, with the hexagonal winding pattern and its awry winding pattern, both seem to have a fill factor of about 0.61. But there is always the hundred to one exception, such as, when a design happens to have the right bobbin, the right number of turns, and the right wire size. This normally only happens when the design is not critical.


Winding Build

Figure 4-6. Ideal Winding on a Rectangular Bobbin. To minimize this bowing effect and to insure a minimum build for either random or layer winding, the round bobbin, shown in Figure 4-8, will provide the most compact design. It can be seen, in Figure 4-8 that the round bobbin provides a uniform tension, all 360 degrees around the bobbin, for both layer and random windings. The other benefit, in using a round bobbin, is the reducing and minimizing of the leakage inductance caused from the bowing.

Winding Build Rectangular Core

Bowing

Figure 4-7. Bowing in Transformer Windings. Winding Build

Round Core

Figure 4-8. A Round Bobbin Insures Minimum Bowing.


S3, Effective Window The effective window, S3, defines how much of the available window space may actually be used for the winding. The winding area available to the designer depends on the bobbin or tube configuration. Designing a layer winding that uses a tube will require a margin, as shown in Figure 4-9. The margin dimensions will vary with wire size. See Table 4-4. It can be seen, in Figure 4-9 and Table 4-4, how the margin reduces the effective window area. When transformers are constructed, using the layer winding technique, there is an industry standard for layer insulation thickness. This thickness is based on the diameter of the wire as shown in Table 4-5. Tube

Layer Insulation Wrapper

Winding Length Margin Figure 4-9. Transformer Windings with Margins. Table 4-4

Winding Margins Versus AWG Margin

AWG 10-15 16-18 19-21 22-31 32-37 38-up

cm 0.635 0.475 0.396 0.318 0.236 0.157

inch 0.25 0.187 0.156 0.125 0.093 0.062

Table 4-5

Layer Insulation Thickness AWG 10- 16 17- 19 20-21 22-23 24-27 28-33 34-41 42-46 Copyright 2002 by Marcel Dekker, Inc. All Rights Reserved.

Insulation Thickness cm inch 0.02540 0.01000 0.01780 0.00700 0.01270 0.00500 0.00760 0.00300 0.00510 0.00200 0.00381 0.00150 0.00254 0.00100 0.00127 0.00050

A single bobbin design, as shown in Figure 4-10, offers an effective area, Wa, between 0.835 to 0.929 for laminations, and 0.55 to 0.75 for ferrites, while a two bobbin configuration, as shown in Figure 4-11, offers an effective area, W a , between 0.687 to 0.873 for the tape C cores.

The toroid is a little different. The term, S3, defines how much of the available window space can actually be used for the winding. In order to wind the toroidal core, there has to be room to allow free passage of the shuttle. If half of the inside diameter is set aside for the shuttle, then, there will be 75% of the window area, (Wa), left for the design which is a good value for the effective window area factor, S3 — 0.75, as shown in Figure 4-12. The toroid would fall into all of the above categories.

Bobbin

Channel Bracket

Coil Mounting Foot

Lamination

Figure 4-10. Transformer Construction with Single Bobbin.

Bobbin

Coil #1

Coil #2

Tape C Core Mounting Bracket

Figure 4-11. Transformer Construction with Dual Bobbins.


0.5 ID Clearance For Shuttle Effective Window Area Core Effective Window area Wa(eff) = (0.75)(7i)(ID)2/4 Figure 4-12. Effective Winding Area of a Toroidal Core.

S4, Insulation Factor The insulation factor, S4, defines how much of the usable window space is actually being used for insulation. If the transformer has multiple secondaries with significant amounts of insulation, S4 should be reduced by 5 to 10% for each additional secondary winding, partly because of the added space occupied by insulation and, partly because of the poorer space factor.

The insulation factor, S4, is not taken into account in Figure 4-12. The insulation factor, S4, is to be 1.0. The window utilization factor, Ku, is highly influenced by insulation factor, S4, because of the rapid buildup of insulation in the toroid, as shown in Figure 4-13.

In Figure 4-13, it can be seen that the insulation buildup is greater on the inside, than, on the outside. For example, in Figure 4-13, if 1.27 cm (1/2") wide tape was used with an overlap of 0.32 cm (1/8") on the outside diameter, the overlap thickness would be four times the thickness of the tape. It should be noted that the amount of overlap depends greatly on the size of the toroid and the required tape. In the design of toroidal components, and using the 0.5 ID remaining for passage of the shuttle, there is normally enough room for the wrapper.


Dacron Wrapper

Wound Ht.

Figure 4-13. Wrapped Toroid.

Circular mil and Square mil There are engineers that use circular mils/amp or square mils/amp. This is the reciprocal current density. The norm is to use amps/cm", which is a true current density. There have been some requests to define circular mils and square mils. First of all, let's define a mil, which is .001 inch. Figure 4-14 shows the area of a square mil, and the area of a circular mil.

One Square mil =

0.001

0.001

One Circular mil =

Figure 4-14. Comparing Circular-Mils and Square-Mils.

To convert Square mils to Circular mils , multiply by 1.2732. To convert Circular mils to Square mils , multiply by 0.7854. To convert Circular mils to Square centimeters , multiply by 5.066x10""To convert Square mils to Square centimeters , multiply by 6.45x10""


Summary I hope I have cleared up some of the mystery of how the window utilization factor, Ku, was derived. I hope the magic of 0.4 is now clear. I have tried to bring together all of the different parts that make up the window utilization and then, explain each one. I hope I have simplified for you the complexity of the window utilization factor. I hope I have not confused you more. As stated at the beginning of this chapter, a good approximation for the window utilization factor is Ku = 0.4.

S, = conductor area/wire area = 0.855, #20 AWG S2 = wound area/usable window area = 0.61 83 = usable window area/window area - 0.75 S4 = usable window area/usable window area + insulation = 1

Ku — S] S2 8^ S4 Ku = (0.855)(0.61)(0.75)(1.0) = 0.391« 0.4 Being a very conservative number, it can be used in most designs. It is an important factor in all designs of magnetic components.

Magnet Wire Standard magnet wire is available in three different materials, as shown in Table 4-6. The most common is copper, but aluminum and silver are available. Aluminum magnet wire is one-third the weight of copper for the same size conductor and one-half the weight for the same conductivity. Aluminum magnet wire is a little more difficult to terminate, but it can be done. Silver magnet wire has the highest conductivity, easy to solder to, and weighs 20% more than copper. Table 4-6

Magnet Wire Material Properties Density

Resistivity

Weight

Resistance

Temperature

Material

Symbol

grams/cm5

uQ/cm

Factor

Factor

Coefficient

Copper Silver Aluminum

Cu

8.89

1.72

1

1

Ag Al

10.49 2.703

1.59

1.18

0.95

2.83

0.3

1.64

0.00393 0.00380 0.00410


Magnet Wire, Film Insulation It is the design engineer's responsibility to ensure that the selected magnet wire, used in the design, is compatible with the environmental and design specification. The environmental specification will set the ambient temperature. The maximum operating temperature of the magnet wire is obtained by summing the maximum, ambient temperature, plus the temperature rise of the magnetic component. After the maximum temperature has been obtained then see Table 4-7 for the Temperature Class. The magnet wire insulation guide listing, in Table 4-7, is only a partial list from NEMA, Standard MW 1000. The maximum operating temperature is the, "Achilles Heel" to the magnet wire. Standard magnet wire is rated by temperature. The range is from 105°C to 220°C, as shown in Table 4-7. The insulation film of the magnet wire is on the surface of the copper wire. This insulation film is the most vulnerable to thermal overloads, so the selection of the insulation film is very critical for long life. When magnet wire is subjected to thermal overloads, or a high, ambient temperature above its rated temperature, the life of the magnet wire is greatly reduced, as shown in Figures 4-15 and 4-16. The engineer must be very careful of hot spots so as not to degrade the service life of the magnetic component. Table 4-7

Magnet Wire Insulation Guide NEMA Temperature

Insulation

Class

Type

Dielectric Constant

105°C 105°C

Polyurethane* Formvar

6.20 3.71

130°C

Polyurethane -Nylon*

MW-79-C

Standard MW 1000

MW-2-C MW-15-C MW-28-C

155°C

Polyurethane- 155

6.20 6.20

180°C

Polyester Solderable*

3.95

MW-77-C

200°C

Polyester-amid-imide

4.55

MW-35-C

220°C

Polyimide (ML)

3.90

MW-16-C

*Solderable insulations

Wire Table Table 4-8 is the wire table for AWG, 10 to 44, heavy film wire. The bare wire area is given in cm2, in column 2, and the circular mils is given in column 3 for each wire size. The equivalent resistance in microohms per centimeter (uQ/cm or 10"6 Q/cm and in wire length for each wire size. Columns 5 through 13 relate to heavy, insulated film coating. The weight of the magnet wire is found in column 13, in grams, per centimeter. Table 4-9 provides the maximum outside diameter for magnet wire with single, heavy, triple, and quad film insulation. The dimensional data is in centimeters and inches, for AWG 10 through 44.


Table 4-8

Wire Table Resistance Bare Area Area AWG f^Q/cm cm2(10~3) cir-mil 20°C cm2(10~3) cir-mil 1 2 4 3 5 6 32.7 55.9000 11046.00 10 52.6100 10384.00 41.4 44.5000 8798.00 11 41.6800 8226.00 12 33.0800 6529.00 52.1 35.6400 7022.00 65.6 28.3600 5610.00 13 26.2600 5184.00 14 20.8200 4109.00 82.8 22.9500 4556.00 104.3 18.3700 3624.00 15 16.5100 3260.00 16 13.0700 2581.00 131.8 14.7300 2905.00 17 10.3900 2052.00 165.8 11.6800 2323.00 18 209.5 9.3260 1857.00 8.2280 1624.00 263.9 7.5390 1490.00 19 6.5310 1289.00 20 332.3 6.0650 1197.00 5.1880 1024.00 21 812.30 418.9 4.8370 954.80 4.1160 531.4 3.8570 761.70 22 3.2430 640. 1 0 23 666.0 3.1350 620.00 2.5880 510.80 24 842.1 2.5140 497.30 2.0470 404.00 1062.0 2.0020 396.00 25 320.40 1.6230 1345.0 1.6030 316.80 26 1.2800 252.80 27 1687.0 1.3130 259.20 1.0210 201.60 28 158.80 2142.0 1.0515 207.30 0.8046 29 2664.0 0.8548 169.00 0.6470 127.70 30 100.00 3402.0 0.6785 134.50 0.5067 79.21 31 4294.0 0.5596 110.20 0.4013 32 5315.0 0.4559 90.25 64.00 0.3242 50.41 6748.0 0.3662 33 72.25 0.2554 34 8572.0 0.2863 39.69 56.25 0.2011 10849.0 0.2268 35 31.36 44.89 0.1589 36 13608.0 0.1813 36.00 25.00 0.1266 37 16801.0 0.1538 30.25 20.25 0.1026 38 16.00 21266.0 0.1207 24.01 0.0811 39 27775.0 0.0932 18.49 12.25 0.0621 14.44 40 9.61 35400.0 0.0723 0.0487 41 7.84 43405.0 0.0584 11.56 0.0397 42 54429.0 0.0456 6.25 9.00 0.0317 4.84 43 70308.0 0.0368 7.29 0.0245 44 0.0202 4.00 85072.0 0.0316 6.25


Heavy Synthetics Turns-Per Diameter cm Inch cm Inch 7 10 9 8 10 0.2670 0.105 3.9 0.094 0.2380 4.4 1 1 0.2130 0.084 4.9 12 0.1900 0.075 5.5 13 0.1710 0.068 6.0 15 0.1530 0.060 6.8 17 0.054 0.1370 19 7.3 0.1220 0.048 8.2 21 0.1090 0.043 9.1 23 0.0980 0.039 10.2 26 0.0879 0.035 11.4 29 0.0785 0.031 12.8 32 0.0701 0.028 14.3 36 0.0632 0.025 15.8 40 0.0566 0.022 17.6 45 0.0505 0.020 19.8 50 0.0452 0.018 22.1 56 0.0409 0.016 24.4 62 0.0366 0.014 27.3 69 0.0330 0.013 30.3 77 0.0294 0.012 33.9 86 0.0267 0.011 37.5 95 0.0241 0.010 41.5 105 0.0216 0.009 46.3 118 0.0191 0.008 52.5 133 0.0170 0.007 58.8 149 0.0152 0.006 62.5 167 0.0140 0.006 71.6 182 0.0124 0.005 80.4 204 0.0109 0.004 91.6 233 0.0096 0.004 103.6 263 0.0086 0.003 115.7 294 0.0076 0.003 131.2 333 0.0069 0.003 145.8 370 0.0064 0.003 157.4 400

Turns-Per crrf Inch2 11 12 11 69 13 90 17 108 21 136 26 169 211 33 41 263 331 51 64 415 515 80 638 99 800 124 1003 156 1234 191 239 1539 300 1933 2414 374 2947 457 3680 571 702 4527 884 5703 6914 1072 8488 1316 10565 1638 13512 2095 17060 2645 21343 3309 25161 3901 4971 32062 6437 41518 53522 8298 10273 66260 13163 84901 16291 105076 18957 122272

Weight gm/cm 13 0.46800 0.37500 0.29770 0.23670 0.18790 0.14920 0.11840 0.09430 0.07474 0.05940 0.04726 0.03757 0.02965 0.02372 0.01884 0.01498 0.01185 0.00945 0.00747 0.00602 0.00472 0.00372 0.00305 0.00241 0.00189 0.00150 0.00119 0.00098 0.00077 0.00059 0.00046 0.00038 0.00030 0.00023 0.00020

Table 4-9

Dimensional Data for Film Insulated Magnetic Wire Wire Size AWG 10 II 12 13 14 15 16 17 18 19 20 21 22 23 24 25 26 27 28 29 30 31 32 33 34 35 36 37 38 39 40 41 42 43 44

Single-Insulation Inches Centimeters 0.1054 0.2677 0.9410 2.3901 0.2134 0.0840 0.0750 0.1905 0.1702 0.0670 0.0599 0.1521 0.0534 0.1356 0.1214 0.0478 0.0426 0. 1 082 0.0382 0.0970 0.0341 0.0866 0.0306 0.0777 0.0273 0.0693 0.0244 0.0620 0.0554 0.0218 0.0195 0.0495 0.0174 0.0442 0.0156 0.0396 0.0139 0.0353 0.0126 0.0320 0.0112 0.0284 0.0100 0.0254 0.0091 0.0231 0.0081 0.0206 0.0072 0.0183 0.0064 0.0163 0.0058 0.0147 0.0052 0.0132 0.0047 0.0119 0.0041 0.0104 0.0094 0.0037 0.0084 0.0033 0.0030 0.0076 0.0066 0.0026 0.0024 0.0061

Maximum Diameter Triple-Insulation Heavy-Insulation Centimeters Inches Centimeters Inches 0.2720 0.1084 0.2753 0.1071 0.0969 0.2461 0.0957 0.2431 0.2172 0.2202 0.0855 0.0867 0.1971 0.0765 0. 1 943 0.0776 0.1765 0.0684 0.1737 0.0695 0.1557 0.0624 0.1585 0.0613 0.1392 0.1417 0.0548 0.0558 0.0502 0.1275 0.0492 0.1250 0.0440 0.1143 0.1118 0.0450 0.1026 0.0395 0. 1 003 0.0404 0.0362 0.0919 0.0897 0.0353 0.0317 0.0805 0.0326 0.0828 0.0742 0.0284 0.0721 0.0292 0.0668 0.0255 0.0648 0.0263 0.0229 0.0582 0.0602 0.0237 0.0544 0.0206 0.0523 0.0214 0.0192 0.0488 0.0185 0.0470 0.0165 0.0419 0.0172 0.0437 0.0394 0.0155 0.0148 0.0376 0.0134 0.0340 0.0141 0.0358 0.0120 0.0305 0.0127 0.0323 0.0292 0.0274 0.0115 0.0108 0.0249 0.0105 0.0267 0.0098 0.0224 0.0241 0.0095 0.0088 0.0084 0.0198 0.0213 0.0078 0.0193 0.0070 0.0178 0.0076 0.0160 0.0175 0.0063 0.0069 0.0062 0.0157 0.0057 0.0145 0.0142 0.0051 0.0130 0.0056 0.0114 0.0127 0.0045 0.0050 0.0044 0.0102 0.0112 0.0040 0.0036 0.0091 0.0040 0.0102 0.0032 0.0037 0.0094 0.0081 0.0084 0.0074 0.0033 0.0029 0.0027 0.0076 0.0069 0.0030


Quad-Insulation Inches Centimeters 0.1106 0.2809 0.2517 0.0991 0.0888 0.2256 0.2022 0.0796 0.1816 0.0715 0.0644 0.1636 0.0577 0.1466 0.0520 0.1321 0.0468 0.1189 0.0422 0.1072 0.0379 0.0963 0.0342 0.0869 0.0782 0.0308 0.0709 0.0279 0.2520 0.6401 0.0579 0.0228 0.0206 0.0523 0.0185 0.0470 0.0166 0.0422 0.0152 0.0386 0.0137 0.0348 0.0124 0.0315 0.0287 0.0113 0.0102 0.0259 0.0091 0.0231 0.0082 0.0208 0.0074 0.0188 0.0067 0.0170 0.0152 0.0060 0.0053 0.0135 0.0047 0.0119 0.0043 0.0109 0.0097 0.0038 0.0035 0.0089 0.0032 0.0081

20,000

10,000 52

Formvar 105°C Insulation MW15-C

o ffi

-a c W

1,000

100 100

200

300

Film Insulation Temperature, °C Figure 4-15. Thermal Endurance, for 105°C Formvar Insulation. 20,000 10,000 o ffi if

Polyimide (ML) 220°C Insulation MW16-C

8 I

w Is

1,000

100 100

200

300

Film Insulation Temperature, °C Figure 4-16. Thermal Endurance for 220°C Polyimide Insulation (ML).

Solderable Insulation Solderable insulation is a special film insulation that is used on magnet wire in low cost, high volume applications. The magnet wire, with this solderable insulation, is wrapped around the terminal or pin, as shown in Figure 4-17. Then the terminal can be dip-soldered at the prescribed temperature, without prior stripping. The ambient temperature range for this type of film insulation is 105°C to 180°C. Copyright 2002 by Marcel Dekker, Inc. All Rights Reserved.

There are drawbacks in using some of the solderable insulation magnet wire. Prior to using, check your application with the wire manufacturer. Some solderable film insulation is not recommended where severe overloads may occur. Some solderable film insulations are susceptible to softening, due to prolonged exposure to strong solvents, such as alcohol, acetone, and methylethy Ike tone.

Terminal Dipped Solder Connection Strain Relief Solderable Insulation

Figure 4-17. Solderable Insulation on a Dip Solder Terminal.

Bondable Magnet Wire Bondable, magnet wires are a film-coated, copper or aluminum, with an additional coating of a thermoplastic adhesive. See Figure 4-18. They are used in applications where it is desirable to have the bonding agent such as a solvent, which will hold the coil form, until it is oven-baked. Most adhesive coatings can be softened with solvents or heat. If a coil is wound with an irregular shape, held in a form, and then, raised to the appropriate temperature, the coil will retain its shape. Bondable magnet wires, have applications, such as armatures, field coils, and self-supporting coils.

Bondable Thermoplastic Adhesive Film Insulation Copper Wire

Figure 4-18. Typical Cross-Section of a Bondable Magnet Wire.

Base Film Insulation All conventional film insulations may be adhesive-coated to achieve a bondable wire. However, care should be taken in selecting wires, which are insulated with high temperature films, since the adhesive coating may not withstand the equally high temperatures. See Table 4-10. The temperatures in Table 4-10 are for reference only. It is wise to always check with the manufacturer for the latest in materials and application notes. The addition of the adhesive coating over the film insulation will result in an increase in the finished diameter, by the same magnitude, as if going from a single to a heavy insulation.


Table 4-10

Bondable Overcoats Operating Temperature

Heat Activation Temperature

105°C

120°- 140°C

Epoxy

130°C

130°- 150°C

Polyester

130°C

130°- 150°C

Methylethylketone Acetone Methylethylketone

Nylon

155°C

180°-220°C

None

Type Polyvinyl Butryal

Solvents Activating Agents Alcohol

Bonding Methods Heat Bonding may be accomplished by the use of a temperature-controlled oven. Small components can use a controlled hot air blower to bond the wires. In either case, caution should be used, when handling the coil while it is still hot, since deformation can take place.

Resistance Bonding is a method where a current is passed through the winding to achieve the desired bonding temperature. This method generates a very even, heat distribution resulting in a good bonding throughout the winding. Many coils can be resistance-bonded at the same time. The current required for one coil, will be the same current required when many are connected in series. Just solder the coils in series, then, adjust the applied voltage, until the same current is reached. Solvent Bonding is a method where the solvent activates the bonding material. This can be done, by passing the wire through a solvent-saturated felt pad, or a light spray application. There are many activating solvents that can be used: denatured ethyl alcohol, isopropyl alcohol, methylethylketone and acetone. The solvents should always be checked on with the manufacturer for the latest in materials and application notes.

Miniature Square Magnet Wire When product miniaturization calls for more copper in a given area, MWS Microsquare film, insulated magnet wire allows design of compact coils that deliver more power in less space. See Table 4-11. Microsquare magnet wire is available in both copper and aluminum. It is also available in a range of solderable and high temperature, film insulation. A cross-section of a number 26, heavy build, microsquare magnet wire is shown in Figure 4-19.


Film Insulation Copper Wire

0.0445 cm

0.0445 cm

Figure 4-19. Cross-Section of a 26, Heavy, Microsquare Magnet Wire.

Table 4-11

Micro-Square Magnetic Wire (Nominal Dimension) Wire

Bare

Bare

Wire

Size

Width

Width

Area

AWG

cm

Inch

15 16

cm" 0.1450 0.0571 0.019614

Wire

Copper

Aluminum

Single

Heavy

Area

Resistance

Resistance

Width

Width

Q/cm 0.000144

cm

3041

Q/cm 0.0000879

cm

0.1483 0.1514

sq-mils

0.1290 0.0508

0.015228

2361

0.0001132

0.000186

0.1323 0.1354

17

0.1151 0.0453

0.011816

1832

0.0001459

0.000239

0.1184 0.1212

18

0.1024 0.0403

0.009675

1500

0.0001782

0.000293

0.1054 0.1080

19

0.0912 0.0359

0.007514

1165

0.0002294

0.000377

0.0940 0.0968

20

0.0813 0.0320

0.006153

954

0.0002802

0.000460

0.0841 0.0866

21

0.0724 0.0285

0.004786

742

0.0003602

0.000591

0.0749 0.0772

22

0.0643

0.003935

610

0.0004382

0.000719

0.0668 0.0688

23

0.0574 0.0226 0.003096

480

0.0005568

0.000914

0.0599 0.0620

24

0.0511 0.0201

0.002412

374

0.0007147

0.001173

0.0536 0.0556

25

0.0455 0.0179

0.002038

316

0.0008458

0.001388

0.0480

26

0.0404 0.0159

0.001496

232

0.0011521

0.001891

0.0427 0.0445

27

0.0361

0.0142 0.001271

197

0.0013568

0.002227

0.0389

0.0409

28

0.0320 0.0126

0.001006

156

0.0017134

0.002813

0.0348

0.0366

29

0.0287 0.0113 0.000787

122

0.0021909

0.003596

0.0312 0.0330

0.0029372

0.004822

0.0277 0.0295

30

0.0253

0.0254 0.0100 0.000587

91

0.0498

Multistrand Wire and Skin Effect Electronic equipment are now operating at higher frequencies, and the predicted efficiency is altered, since the current carried by a conductor is distributed uniformly across the conductor, cross-section only, with direct current, and at low frequencies. The flux generated by the magnet wire is shown in Figure 4-20. There is a concentration of current near the wire surface at higher frequencies, which is termed the skin effect. This is the result of magnetic flux lines that generate eddy currents in the magnet wire, as shown in Figure 4-21.


External Flux, <j> Internal Flux, (j) Magnet Wire Current, I

Figure 4-20. Flux Distribution in a Magnet Wire.

Eddy currents setup by the internal flux, (j), field.

Magnet wire cross-section

Note: The main current shown in the center is being cancelled by the eddy currents. This forces the current to the surface, which causes surface crowding of the magnet wire.

Main current direction

Figure 4-21. Eddy Currents Generated in a Magnet Wire. Skin effect accounts for the fact that the ratio of effective, alternating current resistance to direct current is greater than unity. The magnitude of this effect, at high frequency on conductivity, magnetic permeability, and inductance is sufficient to require further evaluation of conductor size, during design. The skin depth is defined as the distance below the surface, where the current density has fallen to 1/e or 37 percent of its value at the surface.

£=

6.62

k

V7J

cm

e, is the skin depth /, is frequency in hertz K, is equal to 1 for copper When selecting the wire for high frequency, select a wire, so that the relationship between the ac resistance and the dc resistance is 1. /?„

R-de Copyright 2002 by Marcel Dekker, Inc. All Rights Reserved.

i

Using this approach, select the largest wire, operating at 100 kHz.

e = -T=r k' [cm]

8 =

6.62 /100,000

(1), [cm]

8 = 0.0209, [cm] Then, the wire diameter is: £ W c = 2 ( e ) , [cm] 0^=2(0.0209), [cm] 0^=0.0418, [cm] Then, the bare wire area A w(B) is

-,

[cm2]

(3.14)(0.0418) A^B) =0.00137,

,

[cm2 ]

[cm 2 ]

A graph of skin depth, as a function of frequency, is shown in Figure 4-22. The relationship of skin depth to AWG radius is shown in Figure 4-23, where R ac /R dc =l is plotted on a graph of AWG versus frequency.

1.0 :

o.i C

0.01

0.001 IK

10K

100K

Frequency, Hz Figure 4-22. Skin Depth Versus Frequency.


IMeg

50 40

Skin depth is more than the wire radius.

30

O

20 10

Skin depth is less than the wire radius.

0

10K 100K Frequency, Hz

IK

IMeg

Figure 4-23. AWG Versus Frequency at Which Skin Depth Equals the Radius.

To illustrate how the AWG, ac/dc resistance ratio changes with frequency, see Table 4-12.

Table 4-12

AWG ac/dc Resistance Ratio at Common Converter Frequencies 25kHz D(AWG)

8

50kHz Rac

AWG cm cm Rdc 12 0.041868 1.527 0.20309 14 0.16132 1.300 0.041868 0.12814 16 0.041868 1.136 18 0.041868 1.032 0.10178 1.001 20 0.041868 0.08085 1.000 22 0.041868 0.06422 1.000 24 0.05101 0.041868 1.000 0.04052 26 0.041868 1.000 0.041868 28 0.03219 0.041868 1.000 30 0.02557 AWG Copper, skin depth is at 20°C

200kHz

100 kHz

8 cm

Rac

8

Rac

8

Rac

Rdc

cm

Rdc

cm

Rdc

0.029606 0.029606 0.029606 0.029606 0.029606 0.029606 0.029606 0.029606 0.029606 0.029606

2.007 1.668 1.407 1.211 1.077 1.006 1.000 1.000 1.000 1.000

0.020934 0.020934 0.020934 0.020934 0.020934 0.020934 0.020934 0.020934 0.020934 0.020934

2.704 2.214 1.829 1.530 1.303 1.137 1.033 1.001 1.000 1.000

0.014802 0.014802 0.014802 0.014802 0.014802 0.014802 0.014802 0.014802 0.014802 0.014802

3.699 2.999 2.447 2.011 1.672 1.410 1.214 1.078 1.006 1.000

In Table 4-12, it can be seen that when a converter operates at 100 kHz, the largest wire that should be used is a number 26, with an ac/dc resistance ratio of 1.001.


Multistrand Litz Wire The term litz wire is extracted from the German word, meaning woven wire. Litz wire is generally defined, as a wire constructed of individually, film insulated wires, braided together in a uniform pattern of twists and length of lay. This multistrand configuration minimizes the power losses, otherwise encountered, in a solid conductor, due to the skin effect. The minimum and maximum number of strand for standard litz wire is shown in Table 4-13. Magnet wire suppliers will supply larger, twisted magnet wire on request.

Table 4-13

Standard Litz Wire AWG 30 32 34 36 38 40 41 42 43 44 45 46 47 48

Minimum Strands j 3

Maximum Strands 20 20 20 60 60 175 175 175 175 175 175 175 175 175

Approximate AWG 25 27 29 31 33 35 36 37 38 39 40 41 42 43

-> -> -5

3 -> J>

3 3 3 3 3 3 3

Approximate AWG 17.0 19.0 21.0 18.5 20.5 18.0 18.5 19.5 21.0 21.5 22.5 23.5 25.0 25.5

Specialty Wire There are a lot of new ideas out in the wire industry, if only the engineer had the time to evaluate these new concepts to build confidence and apply them.

Triple Insulated Wire Transformers designed to meet the IEC/VDE safety specification requirements for creepage and clearance must adhere to one of the following specifications: 1. VDE0805

2. IEC950

3. EN60950

4. UL1950-3e

5. CSA 950-95

The engineer must be aware that one specification does not encompass all applications. For example the IEC has specifications for office machines, data-processing equipment, electromedical equipment, appliances, and others.


Originally these IEC specifications were developed around linear 50 and 60 Hz transformers, and were not, always, conducive to optimal designs for high frequency, such as switching power transformers. The complexity of a standard, high frequency switching type transformer, designed to the IEC/VDE safety specification, is shown in Figure 4-24. In any switching transformer, coupling has the highest priority because of the leakage flux.

Wrapper Insulation Winding Area "^ ~

Bobbin Flange Secondary

Winding Area

Primary 3 Layers Insulation Minimum Positive Tape Barrier

Figure 4-24. Bobbin Cross-Section Design to Meet IEC/VDE Specifications.

The triple, insulated wire was developed to meet the above specification and eliminate the need for three layers of insulating tape between primary and secondary. Also, the triple, insulated wire eliminates the need for the creepage margin, and now, the whole bobbin can be used for winding. This wire can also be used as hook up wire, from the primary or secondary, to the circuits, without the use of sleeving or tubing.

The construction of the triple, insulated wire is shown in Figure 4-25. The temperature range for this type of wire is from 105°C to 180°C. The dimensions for triple, insulated wire are shown in Table 4-14, using a 0.002 inch coat per layer. Other thicknesses are available. The manufacturer, Rubadue Wire Company, is listed in the Reference section on page 4-34.

^v

x

Copper conductor 1st insulation layer 2nd insulation layer 3rd insulation layer

Figure 4-25. Triple, Insulated Wire Construction.


Table 4-14

Triple Insulated Wire (.002) Material Area

AWG 16 18 19 20 21 22 23 24 25 26 27 28 29 30 32 34 36 38

cm 2 (10~ 3 ) 13.0700 8.2280 6.5310 5.1880 4.1160 3.2430 2.5880 2.0470 1.6230 1.2800 1.0210 0.8046 0.6470 0.5067 0.3242 0.2011 0.1266 0.0811

Bare Wire Diameter Diameter inch mm 0.0508 1.2903 0.0403 1.0236 0.0359 0.9119 0.0320 0.8128 0.0285 0.7239 0.0253 0.6426 0.0226 0.5740 0.0201 0.5105 0.0179 0.4547 0.0159 0.4039 0.0142 0.3607 0.0126 0.3200 0.0113 0.2870 0.0100 0.2540 0.0080 0.2032 0.0063 0.1600 0.0050 0.1270 0.0040 0.1016

Resistance |LiQ/cm 132 166 264 332 419 531 666 842 1062 1345 1687 2142 2664 3402 5315 8572 13608 21266

With Insulation Diameter Diameter inch mm 1.5951 0.0628 1.3284 0.0523 1.2167 0.0479 1.1176 0.0440 0.0405 1.0287 0.9474 0.0373 0.0346 0.8788 0.0321 0.8153 0.7595 0.0299 0.7087 0.0279 0.0262 0.6655 0.6248 0.0246 0.0233 0.5918 0.0220 0.5588 0.0200 0.5080 0.0183 0.4648 0.0170 0.4318 0.4064 0.0160

Triple Insulated Litz High frequency litz wire, shown in Figure 4-26, is also available, triple insulated wire from manufacturers. The insulation, layers' thickness for litz wire comes in 0.002 and 0.003 inches.

1st insulation layer 2nd insulation layer 3rd insulation layer

Figure 4-26. Triple, Insulated Litz Wire.


Copper conductor Film Insulation

Polyfilar Magnetic Wire Poly or multiple strands of magnet wire, bonded together, can be used in many high frequency transformer and inductor applications. Round polyfilar magnet wire is shown in Figure 4-27, and square polyfilar is shown in Figure 28. Both can be used in place of foil in some applications. Polyfilar magnet wire can be used as a foil type winding, such as a low voltage, high current, or even a faraday shield. The polyfilar, magnet wire strip width can be easily increased or decreased, by adding, or removing wires to provide the proper strip width to fit a bobbin. It is relatively easy to wind. Polyfilar wire has complete insulation, and it does not have the sharp edge problem that could cut insulation in the way foil does. It is not recommended to wind a transformer with polyfilar magnet wire, in order to have an exact center tap, unless it is just a few turns, because of the penalty in capacitance. If the use of polyfilar is necessary, then use a magnet wire with a film insulation that has a low dielectric constant. See Table 4-7.

Bondable Thermal Adhesive Copper conductor Film Insulation

Figure 4-27. Polyfilar, Strip-Bonded, Round Magnet Wire.

Bondable Thermal Adhesive Copper conductor Film Insulation

Figure 28. Polyfilar, Strip-Bonded, Square Magnet Wire. Standard Foils The biggest advantage for using foil over magnet wire is the fill factor. The design of a high current, high frequency, dc to dc converter is common place. The main reason for going to high frequency is the reduction in size. The power transformer is the largest component in the design. When designing high frequency transformers, the design equations relate to a very small transformer. When operating transformers at high frequencies, the skin effect becomes more and more dominate, and requires the use of smaller wire. If larger wire is required, because of the required current density, then, more parallel strands of wire will have to be used (litz wire). The use of small wire has a large effect on the fill factor.


When using foil, the gain in the fill factor is the biggest improvement over htz. To make a comparison, a litz design is shown in Figure 4-29, and a foil design is shown in Figure 4-30. In the litz design, there is a percentage of the winding area, which cannot be used for the conductors. This lost area is made up of voids, space between the wires, and the insulation film on the wire. The foil wound coil, shown in Figure 4-35 can be designed to make optimum use of the available winding area. Each turn of the foil can extend within limits, edge-to-edge of the bobbin or tube. The insulation required between layers is at a minimum, as long as the foil has been rolled to remove the sharp burr.

Winding Build

Winding Length

Figure 4-29. Layer Winding, Using Litz Magnet Wire.

Winding Build

t

Winding Length

Figure 4-30. Layer Winding, Using Foil with Insulation.

The Use of Foils Designing transformers and inductors, with foil, is a very laborious task, especially if the engineer only does it now and then. A monumental job, in itself, is finding out where to get the materials. Foil has its advantages, mainly, in high current, high frequency, and a high density environment.

The window utilization factor, K u , can be greater than 0.6, under the right conditions, without a lot of force. The standard foil materials used, by transformer engineers, are copper and aluminum. The engineer has a good selection of standard thicknesses as shown: 1.0 mil, 1.4 mil, 2.0 mil, 5.0 mil, and 10 mil


The engineer will find other thicknesses, available, but standard thicknesses should be considered first. Be careful of using a nonstandard thickness. What you might be using could be from an overrun, and could create problems for you. Foil comes in standard widths, in inches, as shown:

0.25, 0.375, 0.50, 0.625, 0.75, 1.0, 1.25, 1.50, 2.00, 2.50, 3.00, 4.00

(inches)

Standard widths are the widths that are most readily available. There are also different styles of pre-fab foils, as shown in Figures 4-31, 4-32, and 4-33.

Cuffed Conductor

Backed Conductor

Figure 4-31. Pre-fab Foils.

p-TsSssssssss.i-3

Backed Multiple Conductor

Sandwiched Conductor


Jacketed Conductor

Jacketed Multiple Conductor


Although special slitting is done all the time, there is normally a minimum buy. When slitting is done, special care must be attended to, with the sharp edges, as shown in Figure 4-34. The cut edge should be Copyright 2002 by Marcel Dekker, Inc. All Rights Reserved.

rolled after slitting it, at least two times, to remove the sharp burrs that could cut through the insulation. It is wise, not to use insulation between layers of less than 1 mil.

Sharp edge caused by slitting.

Figure 4-34. Foil with Sharp Edge Burrs after Slitting.

When winding transformers or inductors with foil, special care must be taken with lead finishing. One of the biggest problems about using foil is solder wicking. This wicking will puncture the insulation, resulting in a shorted turn. The normal insulation used for foil is very thin. Winding with foil, the coil is still subjected to bowing, only more so, as shown in Figure 4-7.

Foil used for winding transformers and inductors should be dead soft. There is another shortcoming about using foil, and that is, the inherit capacitance build-up, as shown in Figure 4-35.

Wrapper

nnnmrn Layer Capacitance

Figure 4-35. Foil Capacitance Equation.


The capacitance build up is expressed:

'K(N-I)(MLT:)(G)} } ^ -^-f-

C = 0.0885] —^

K - Dielectric Constant

d

)I

[pfd]

MLT = Mean Length Turn

N = Number of Turns

G = Foil Width, cm

d = Layer Insulation Thickness, cm

The dielectric constant K for different materials can be found in Table 4-15.

Table 4-15

Dielectric Constants Material Kapton Mylar Kraft Paper Fish Paper Nomex

K 3.2-3.5 3-3.5 1.5-3.0 1.5-3.0 1.6-2.9

Calculating, MLT The mean length turn, (MLT), is required to calculate the winding resistance and weight for any given winding. The winding dimensions, relating to the mean length turn, (MLT), for a tube or bobbin coil are shown in Figure 4-36.

Calculating, MLT (toroid) It is very difficult to calculate the mean length turn (MLT) for a toroidal core that would satisfy all conditions. There are just too many ways to wind a toroid. If the toroid were designed to be wound by machine, then, that would require a special clearance for a wire shuttle. If the toroid were designed to be hand-wound, then, the wound, inside diameter would be different. The fabrication of a toroidal design is weighted heavily on the skill of the winder. A good approximation for a toroidal core, mean length turn, (MLT), is shown in Figure 4-37.


F = Winding tube thickness (MLT)j, first winding (MLT), single winding (MLT)2, second winding

MLT

2F)+TiA,

MLT,

single winding

+ 7rB, first winding

MLT

C),

second winding

Figure 4-36. Dimensions, Relating to the Winding Mean Length Turn, (MLT).

Mean Length Turn (MLT) Wound Toroid Toroidal Core

}Ht

\ '

38mm * *

Cover Wall Thickness Thickness 0.02 inches 0.0 15 inches 0.51 mm 0.51 mm 0.02 inches 0.025 inches 0.64 mm 0.51 mm 0.031 inches (min) 0.03 1 inches (min) 0.80 mm (min) 0.80 mm (min) * Magnetic device mounted by bracket, a clamp, or a similar device.

Type Fastener Screw Screw Screw Screw Bracket Bracket

Selecting the Enclosure The enclosure must be selected to best fit the magnetic device. There must be ample room for the terminal board, and space to route the leads. The selected enclosure should provide ease of assembly and inspection. If the selected enclosure is larger than it needs to be, then additional embedment would be required to fill these voids. See Figure 7-4 and Figure 7-5. Always select an enclosure that requires a minimum of embedment. Too much embedment will put undue stress on the magnetic device.

Fastener Tube Fasteners The fastener tube wall thickness shall be 0.031 of an inch (0.08 cm), minimum, for all magnetic devices. The fastener tube length shall be identical to the height of the enclosure and extend through the base and cover or spacer, as applicable in all applications. The ID of the fastener tube shall be 0.125 of an inch when, a 4-40 screw is specified and 0.144 of an inch when a 6-32 is the specified screw for single fastener tube application. Fastener tubes are shown in Figure 7-1. Where two or more fastener tubes are used, the internal diameter shall be 0.138 of an inch when, a 4-40 screw is specified and 0.151 of an inch when a 632 is the specified screw. Excessive Amount of Embedment Material

L: 3:

C Core

J Inserts Acceptable

Unacceptable

Figure 7-4. Comparing Enclosures C Cores. Copyright 2002 by Marcel Dekker, Inc. All Rights Reserved.

Excessive Amount of Embedment Material

Toroid

Inserts

Fastener Tube Acceptable

Unacceptable Figure 7-5. Comparing Enclosures C Cores.

Threaded Fasteners Threaded Fasteners Threaded Fasteners, embedded in the encapsulation material, shall be of the blind type. The threaded fasteners will be secured in place with a 360 degrees bead of epoxy adhesive. The threaded fasteners or blind type inserts are shown in Figure 7-6. Epoxy Adhesive

Threaded Fastener

Enclosure Wall Figure 7-6. Enclosures with Blind Type Threaded Fasteners.

Terminal Board Terminal Board Material The internal terminal boards shall be fabricated from epoxy-glass laminate, per MIL-P-18177, Type GEE, flame retardant Grade 4, or MIL-P-13949, Type GF (flame retardant).


Terminal Board Position The terminal board shall be positioned as follows: 1.

Bonded to the wall of a rectangular cup, as shown in Figure 7-7.

2.

Bonded to the wall of a round cup, as shown in Figure 7-8.

3.

Bonded to the core, as shown in Figure 7-9.

Bifurcated Terminals

C Core

Terminal Board

\

Fiberglass Enclosure

Inserts Figure 7-7. Terminal Boards, Bonded to the Wall of Rectangular Cup.


,/

.Terminal Board

Fastener Tube Toroid Fiberglass Enclosure

Figure 7-8. Terminal Boards, Bonded to the Wall Round Cup. Bifurcated Terminals

Terminal Board

C Core

Inserts

Fiberglass Enclosure Figure 7-9. Terminal Board, Bonded to the C Core.


Terminal Board Outline The terminal board shall have embedment, flow-through holes. The holes shall be 0.125 to 0.25 of an inch in diameter and shall number four to each square inch of board surface. The holes shall be located a minimum of 0.050 of an inch from any edge or installed terminal. See Figure 7-10.

Bifurcated Terminals Terminal Board

0.015 of an inch minimum 15-20 mils Thick

0.060 of an inch minimum

\

Q p O QIQ O 0.050 of an inch minimum Embedment Flows Through Holes.

End View This part of the board is cut away to clear breakout of interconnecting leads.

Figure 7-10. Terminal Board.

Terminals and Leads Terminal Description Terminals shall be bifurcated or turret, solderable, and capable of being permanently fastened to epoxy glass board. Terminals shall be procured, to the latest Mil Spec. It is common for a single bifurcated terminal to handle multiple terminations. See Figure 7-11. The terminal selected must be able to handle the required number of connecting lead wires.

Terminal Installation Terminal Installation Terminals shall be swaged using the force, specified in Table 7-2.


The body of the last stranded wire, added to the bifurcated terminal, shall not protude more than 50% above the tines.

Terminal Board

Figure 7-11. Terminal Board. Copyright 2002 by Marcel Dekker, Inc. All Rights Reserved.

Table 7-2. Swage Force for Terminals.

Swaging Force for Terminals Units

*Nominal Force (pounds/kilograms)

Maximum Force (pounds/kilograms)

pounds

80

100

kilograms

36

45

pounds

130

150

Approximate Size Swage Barrel

0.041 0.062 0.078

0.09 0.112

kilograms

59

68

pounds

200

225

kilograms

102

pounds

91 250

300

kilograms

113

136

pounds

500

800

kilograms

227

363

This is the force which is required to just meet minimum, roll-over requirements.

Terminal Flange The swage flange of the terminal shall be seated and then, there will be sufficient tightness to assure that the terminal will not move. Maximum permissible height of the terminal swage above the plane of the wiring board shall be 0.012 of an inch and the edge of the rollover shall not be more than 0.004 of an inch above the board surface.

Damaged Terminals Damaged Terminals Damage to the funnel type swage and loose terminals is unacceptable. See Figure 7-12. 1.

Acceptable

2.

More than two cracks in the terminal flange are unacceptable.

3.

A loose terminal, as a result of insufficient swage force, is unacceptable. An edge of rollover, more than 0.004 of an inch above the board surface is unacceptable.

4.

Funnel type swage is unacceptable.

Bifurcated Terminals Installation The bifurcated terminals shall show no evidence of damage caused by the swaging tools. See Figure 7-13. 1.

Acceptable

2.

If the installation is not perpendicular to the plane of terminal area, it is unacceptable.

3.

Bent tines are unacceptable.


Figure 7-12. Terminal Swaging.

Figure 7-13. Bifurcated Terminal Installation. Turret Terminals Installation The turret terminals shall show no evidence of damage caused by the swaging tools. See Figure 7-14. 1.

Acceptable

2.

If the installation is not perpendicular to the plane of terminal area, it is unacceptable.

3.

Bent terminals are unacceptable.

1

2

3

Figure 7-14. Turret Terminal Installation.


Measles Terminal Installation (measles) Small white spots, or "measles", caused by terminal installation, shall be acceptable provided they do not form a continuous path between terminals, as shown in Figure 7-15. The small white spots, or "measles" can appear after time or temperature. 3

Unacceptable

Acceptable Ideal

Figure 7-15. Terminal Board with Spots Called, "measles."

Leads Terminal Leads A stranded, insulated terminal lead, with a minimum length to facilitate testing and assembly, shall be used for connection to the magnetic device. External Wire Size The external lead wire size shall be equal to, or greater than, the area of the magnet wire used in the magnetic device. The minimum external conductor size shall be 26 AWG, stranded wire. Bifurcated terminals, or solder ferrules shall provide the solder interconnection between the coil and external leads for Wire 15, AWG, and smaller. The solder interconnection for wire sizes, 14 AWG, and larger, shall be soldered directly to the external lead by the use of a ferrule appropriately sized. Stranded Lead Wire

Ferrule

Magnetic Wire X

Solder

Solder

Figure 7-16. Ferrule Connection Using Stranded and Magnetic Wire. Copyright 2002 by Marcel Dekker, Inc. All Rights Reserved.

Magnetic Wire Termination Small Leads Winding leads of 33 AWG, shall be wrapped around a tine of the terminal, a maximum of 180°. See Figure

7-17.

180° Hook Bifurcated Terminals

Magnet Wire Figure 7-17. Winding Lead Termination for a 33 AWG. Large leads Winding leads of 32 AWG up to 15 AWG, shall be terminated without wrapping. See Figure 7-18.

Straight Lead


Magnet Wire Figure 7-18. Winding Lead Termination for a 32 to 15 AWG. Magnetic Component Lead Preparation (Pattern) If the spec control drawing (SCD) does not call out the length of the finish leads, then do the following: Using an enclosure with terminals as a pattern, place the magnetic device in the enclosure, and align the leads with the terminals. With the magnetic device in place, route the leads to provide suitable strain relief, plus sufficient length to rework the solder joint once. See Figure 7-19.

Tinned Magnet Wire (finished leads) Dacron Wrapper

Woven Glass Sleeving

Figure 7-19. Toroidal Winding Leads Breakout. Copyright 2002 by Marcel Dekker, Inc. All Rights Reserved.

External Leads External Lead Connection External leads connected to an internal board shall extend through a separate opening in the enclosure, or encapsulation material, with spacing of 0.125 of an inch minimum on the centers, as shown in Figure 7-20. The external leads shall emerge, evenly spaced within a 90° sector or side, unless minimum spacing limits require a larger angle, as shown in Figure 7-21. If the number of leads is greater than that which can be accommodated around the periphery, the leads may be aligned in two rows. The external lead length will be six inches long, unless otherwise specified in the spec control drawing (SCD). 0.125 Inch Maximum Cover O

O

Lead Exit Holes 0.125 Inch Minimum

O

t


Figure 7-20. External Leads Breakout Location.

Bifurcated Terminals Fiberglass Enclosure Terminal Board

External Insulated Leads

Figure 7-21. Top View Showing Leads Breakout.

Installing the Magnetic Component Installation of Magnetic Device The magnetic component shall be placed into the enclosure in the location specified on the drawing. If the location is not specified, the magnetic device shall be located, as centrally in the enclosure cavity as practicable. The magnetic device may be spot-bonded in place, when properly located. See Figure 7-22. For an approved spot bonding material, refer to Chapter 8. Copyright 2002 by Marcel Dekker, Inc. All Rights Reserved.

Bifurcated Terminals Sleeved Magnet Wire


Fastener Tube

(4) Spot Bonding

Toroid

Figure 7-22. Spot Bonding the Toroid.

Terminating the Leads Terminating the Leads After the magnet wire leads are terminated, with suitable strain relief, the external leads are attached to the terminals and soldered. The design of the enclosure and internal terminal boards shall be such that flexing of external lead wires, prior to encapsulation, shall not apply appreciable strain to the terminals. The standard length for lead wires is 6 inches; if the external lead wire is to be longer, it must be called out in the spec control drawing (SCD). After the lead wires have been soldered, a verification is done of the lead wire numbers, with the numbers on the magnetic device being the same, then the solder joints will be inspected. After inspection of solder joints and lead numbers, the magnetic device is ready for a pre-pot test. See Figure 7-23.

Fiberglass Enclosure Bifurcated Terminals (4) Spot Bonding Dacron, Covered Toroid Lead Markers (5) Teflon Leads, 6 Inches Long

Figure 7-23. Magnetic Device in Final Assembly.


Fastener Tube

Surface Mounts for High-Rel, Power Magnetics The surface mount carrier (SMC) is a means of attaching a component to a printed circuit board (PCB). There are many types of packages, and headers for mounting magnetic components to printed circuit boards. These mountings come in different configurations and styles. There are horizontal, vertical, open, and surface mounting, all of which are designed for the printed circuit board. The design engineer must select which configuration will best fulfill the design requirement. There are five areas to investigate when selecting a surface mount carrier (SMC) for use in a Hi-Rel environment: (1) molding material; (2) mechanical integrity; (3) terminal material; (4) solderability; (5) inspectability.

Introduction Mounting and packaging for magnetic components have become more important in recent years because the size of the power converter has become smaller. The reduction in size of the magnetic component is due to the higher operating frequency, and the power demand required by new scientific instruments and microprocessors. The surface mount carrier (SMC) is ideally suited for high frequency converters. However, using standard packaging has its drawbacks, such as a limited number of sizes for a given configuration. This could lead to design trade-offs, in order to get an adequate fit. Another factor is the current carrying capacity of the surface mount carrier pins. The output power of the converter has drop, but the output current could remain the same. The conductor material of the pins should be of high conductance for a minimum voltage drop. Even with copper pins, the cross section of the pin is not enough to handle the current capacity, and pins have to be paralleled to minimize the voltage drop.

Selecting the Best Plastic for Your Application Plastics used in molding toroid mounts and headers come in two broad categories: thermoset and thermoplastic. Thermoset plastics include epoxies, phenolics, and diallyl phthalate, (DAP), which are known for their environmental stability, and ability to tolerate over 400°C (750°F) without melting. Thermoplastics include nylon, polypropylene, polycarbonate, polyester, (Valox, Rynite), LCP (Vectra), and PPS (Ryton), which will begin to melt if they experience temperatures much above 260°C (500°F) for an extended period. The chemistry that gives thermoplastics a lower melting point also makes it less expensive to mold, giving it a cost advantage over the thermoset plastics. Thermoplastics are widely used in applications that do not experience temperatures above 260°C (500°F), except for a few seconds during the winding lead to terminal, and component to a PCB soldering process. Thermoset plastics, on the other hand, are popular in magnetic applications when they are used in conjunction with self-stripping magnetic wire. The unstripped and untinned magnetic wire is wrapped around the terminal, molded into a thermoset header or toroid mount, and then, dipped into a 400°C (750°F) solder pot. The high temperature solder will burn off the wire's insulation, tin the wire, and solder it to the terminal in a cost-effective way, without melting the mount. Copyright 2002 by Marcel Dekker, Inc. All Rights Reserved.

There are trade-offs between the two plastic types that must be considered. Parts molded from thermoplastic, will require pretinning the winding leads, and careful heat management while soldering the leads to the mount, and soldering the mount to the circuit board. The thermoset parts can be used with selfstripping magnetic wire. Several terminations can be soldered at once. This type of termination makes it ideal for fine insulated, magnet wire.

Through-Hole Toroid Mounts Through-hole headers and mounts connect components to a printed-circuit board by inserting a terminal or lead through a hole in the board and soldering it to the opposite side. Through-hole headers and mounts have two basic configurations, horizontal or vertical.

Horizontal Toroid Mounts Horizontal through-hole headers or toroid mounts are widely used as a platform or holder for mounting wound toroids on their side. They are usually molded from plastic, with the size, shape, and number of termination points specific to the wound toroid. They are most often either a platform, as shown in Figure 7-24, or cup-shaped, like Figure 7-25. The molding of either configuration will typically include standoffs, which allows the printed circuit board's, cleaning solutions to easily flow under the component. The minimum standoff is usually 0.0015 in.

The leads from the toroidal winding are attached to the mount's terminals, usually by soldering. Once the toroid is attached to the mount, this component is ready for insertion into a printed circuit board. Magnetic components are heavy, and the mechanical characteristics of the solder connection are as important as the electrical integrity. Printed circuit boards, with unplated through holes using heavy components, may require a clinched terminal, as described in Figure 7-26. Printed circuit boards, with printed through holes, offer good mechanical integrity without clinching, providing a successful intermetallic bond. This bond is created during the board solder process, as shown in Figure 7-27.

The toroids can be attached to the mount with either adhesives or mechanical means. Cup-shaped toroid mounts can be filled with a potting or encapsulation compound to both adhere and protect the wound toroid. Horizontal mounting offers both a low profile and a low center of gravity in applications that will experience shock and vibration. As the toroid's diameter gets larger, horizontal mounting begins to use up valuable circuit board real estate. If there is room in the enclosure, vertical mounting is used to save board space.


Header

Top View

Toroid

Solder Terminals

mm/

Side View

Standoffs Pads

Figure 7-24. Horizontal Platform with Through Hole.

Toroid Bottom View

Solder Pins

S •N • •

»* * »* *

|'

iriV

Side View

s ,X „ ~

rapIF

IT~l

Standoffs Pads

Figure 7-25. Horizontal Cup with Through Hole. Toroid

\

1

Header •M^H

J

Terminals Cut to Length

>

^r

^

^

V^/

A

V/'

1

? -^ — Printed Circuit Boarc Solder

Plated Solder Pad

Figure 7-26. Clinched Terminals. Copyright 2002 by Marcel Dekker, Inc. All Rights Reserved.

Toroid

K

Header Terminals Cut to Length

/

Printed Circuit Board Plated Through Holes and Pad

Solder Figure 7-27. Clinched Terminals.

Vertical Toroid Mounts Toroids using vertical through-hole headers and vertical cups, are used to save circuit board real estate. As, with horizontal mounts, vertical mounts are usually molded from plastic, with the size, shape, and number of termination points specific to the application. The molding of either configuration will typically include standoffs, which allow the printed circuit board cleaning solutions to flow easily under the component. The minimum standoff is usually 0.015 in. Vertical toroid mounts come in many configurations, several of which are shown in Figures 7-28 and Figure 7-29. Much of their structure is devoted to supporting the vertical toroid and creating a stable base for connection to the printed circuit board.

Vertical Mount

Toroid Vertical Support

Standoff Pads

Solder Pins

Figure 7-28. Vertical Mount with Through Hole.

Vertical Cup

Standoff Pads

Toroid

Solder Pins

Figure 7-29. Vertical, Open Cup with Through Hole.


The leads from the toroidal winding are attached to the mount's terminals, by soldering, as shown in Figure 7-28. The toroids can be attached to the mount with either adhesives or mechanical means, or by encapsulation. The cup-shaped toroid mounts shown in Figure 7-29 and Figure 7-30, can be filled with a potting or encapsulation compound to both adhere and protect the wound toroid.

Toroid Vertical Cup Standoff Pads Solder Pins Figure 7-30. Vertical Cup with Through Hole. Vertical mounting saves circuit board real estate when a toroid's diameter gets larger, but it creates a component height issue. Vertical mounting also raises the component's center of gravity, making it vulnerable to shock and vibration.

Surface Mount, Toroid Mounts Surface mount components are a direct response to smaller size magnetic components and improved circuit board real estate. Instead of a pin or terminal passing through a printed circuit board, and being soldered on the opposite side, surface mount components utilize a flat solderable surface that is soldered to a flat solderable pad on the face of the printed circuit board. See Figures 7-31 and 7-32. For ease of manufacturing, the circuit board is usually coated with a paste-like formulation of solder and flux. With careful placement, surface mount style components on solder paste will stay in position until temperatures are elevated, usually from an infrared oven. The temperature melts the solder paste and solders the mount's flat terminals to the circuit board's pad. Surface Mount Header

Component Solder Pins

Standoff Pads

Figure 7-31. Gull Wing Surface Mount Carrier (SMC). Copyright 2002 by Marcel Dekker, Inc. All Rights Reserved.

r

^-^ Component Solder Pm —— *-—. Gull Wing Surface Mount

'

.} (

7^

11

11

\ 1

^N.

Printed Circuit Board /^= \r^> , ., mSS\\\\\1

L[amps] FJ

The voltage drop, V2, should be very small compared to Vac:

= , [volts]

22

100

Then: = - - , [ohms] 100/_

The series network of Rl and Cl perform the integration of the applied voltage. The resistance should be very large compared to the impedance of the capacitor at the operating frequency:

Rl = —, coC

[ohms]

Then:

The measurement of the B-H loop is then:

'"

4A4NAC

And Tr

H =

n

,

r

—- L [oersteds]

The voltage, Vc, across the capacitor, Cl, is directly proportional to the, Bm, in tesla. The voltage, V2, across the resistor, R2, is directly proportional to H in oersteds. The oscilloscope will have to be calibrated with a known, magnetic material.


High Voltage Testing Transformer and Inductor Test (Magnetic Component) Test voltage shall be applied to the magnetic component undergoing the test in a vacuum chamber, at room pressure. Corona detection networks, as shown in Figure 10-19, shall be used in appropriate leads to monitor for corona or arcing. Typical corona and arcing waveforms are shown in Figure 10-20. With the voltage continuously applied, the air pressure shall be reduced to the lower limit, 5.0 x 10"4 torr, and then raised to 50 torr. This pressure shall be varied between the upper and lower limits several times for a minimum length of one hour in the critical pressure region. At the conclusion of the test, the voltage shall be removed, and the magnetic component shall be brought back to ambient room pressure. During the test, any evidence of corona or arcing shall be cause for rejection.

Input Return

; • ' Corona ia LI

}

11

1

LB2 Arc

L2

Figure 10-19. Corona Detection Network Schematic (CDN). Parts' List Cl = 300 pf 400 volt mica capacitor. LI = 2.6 mh+/-20%, 40 ohms air-core. L2 = 3.0 h+/-20%, 225 ohms. LB1= NE-2 Neon, AC/DC visual corona indicator. LB2= NE-2 Neon, AC/DC visual arc indicator. SI - SPST, Bypass switch. VTVM, AC/DC corona detector. Scope, AC corona indicator.


Oscilloscope

Corona Burst

AC Supply Frequency Abrupt breaks in the scope trace or burst amplitudes, >5 volt pk-pk, indicates arcing, rather than corona. Figure 10-20. Typical Oscilloscope, Corona Burst Pattern. Test Configuration The configuration for testing magnetic components shall be shown, as in Figure 10-21 and Figure 10-22. Electrical connectors and wire leads shall be corona proof, when the pressure is in the critical pressure region. Transformer Mounting Magnetic components, undergoing tests, shall be mounted in a similar manner to that in the subsystem, especially, with regards to, adjacent metallic surfaces, terminals, etc. Potting, coating or encapsulation shall be similar to that applied to the magnetic component part in the complete subsystem. Interwinding Insulation The insulation integrity between windings, between the winding and the core, and between the winding and the case, if one is used, or between windings and mounting inserts, if used, shall be tested by applying a voltage between the various windings, cores, etc., in accordance with Figure 10-21 and Table 10-1. The voltage shall be applied for a minimum time of 5 +/- 1 seconds. Table 10-1 Working Voltage (dc plus peak ac)

Test (rms)

250 to 700 volts

2.8x working voltage

Above 700 volts

1.4x working voltage plus 1000

Intrawinding Insulation Magnetic components shall be subjected to a voltage to cause twice the rated voltage to appear across all windings at the critical pressure region. The test voltage may be applied to any winding, as shown in Figure 10-22. Copyright 2002 by Marcel Dekker, Inc. All Rights Reserved.

Vacuum Chamber or Bell Jar Transformer Under Test (TUT)

Mounting Plate

Insulator

High Voltage Feed Through Low Voltage Feed Through

High Voltage Return CDN = Corona Detection Network

Figure 10-21. Transformer Interwinding, Voltage Breakdown Test.

Note: 1.

Switch, SI, in the Corona Detection Network, shall be closed for this test.

2.

Grounding type selector switch may be used with one Corona Detection Network.

3.

CDN = Corona Detection Network. See Figure 10-19.


Vacuum Chamber or Bell Jar Transformer Under Test (TUT) \

Mounting Plate

Insulator

High Voltage Feed Through

/

Low Voltage Feed Through

CDN = Corona Detection Network

Figure 10-22. Transformer Intrawinding, Voltage Breakdown Test. Note: 1.

Resistors are loading, R's, for the secondary winding. (They may be located outside of the chamber).

2.

Switch, SI, in the Corona Detection Network, shall be closed for this test.

3.

The power supply, ac voltage, shall be twice-rated voltage for the winding, energized with the frequency raised, so that the ac current flowing is equal to, or less than the rated current.

4.

The grounding type selector switch may be used with one Corona Detection Network.

5.

CDN = Corona Detection Network. See Figure 10-19.


Care must be taken to terminate all of the magnetic components terminals so that external corona or arcing is prevented. Mountings and windings shall be grounded as they would be in service. The test frequency shall be far enough from any resonant frequency, so that voltages, more than twice rated, will not occur in any winding. Twice the rated voltage shall be applied across a winding at approximately twice the normal frequency, or in a manner that will not exceed twice rated current.

Examination During and After Test Magnetic components, undergoing the tests, shall show no internal corona or arcing during the test. After the test, the magnetic component shall be examined for evidence of arcing, flashover, breakdown of insulation, and damage. Visible damage or detection of voltage breakdown or corona, by insulation, shall be cause for rejection.

High Voltage Test Equipment Corona Detection Network Detection of corona or arcing shall be by a current, or series type network, as shown in Figure 10-19. Insert the Corona Detection Network in series with the ground, or return of the high voltage circuit being tested. Indicators, LB-1, and LB-2, shown in Figure 10-19, serve the dual purpose of corona and arc indication, and over voltage protection. Inductance LI and L2 are in series and provide a significant ac impedance, from audio frequencies to nearly 0.5 MHz respectively, which is the significant frequency range of corona voltage. The function of the capacitor, Cl, is to attenuate the ac supply frequency to a sufficient degree, but pass the corona burst pulses, so the maximum sensitivity of the oscilloscope may be utilized. The power supply waveform, appearing on the oscilloscope, shall serve as a reference for corona bursts, as shown in Figure 10-20. Thus corona bursts can be distinguished from extraneous noise in the circuit. Vacuum Chamber The vacuum equipment shall have sufficient capacity to pump down to the critical region in 20 minutes with the chamber air and outgassing loads present. Switching Switching of the magnetic component, high voltage leads shall be accomplished externally to the vacuum system. Oscilloscope The frequency response of the vertical amplifiers of the oscilloscope shall be flat to 1.0 MHz. Deflection sensitivity of the trace shall be 10 millivolts/cm or less. The zero trace of the oscilloscope shall be blanked out visually by opaque tape, so that the intensity can be turned up sufficiently to see the trace.


Quality Assurance


Introduction Pursuing reliability in the manufacturing of transformers and inductors primarily involves attention to details, coupled with close controls in all phases of manufacturing. The manufacturing cycle should be controlled and monitored by a conscientious Quality Assurance (QA) program, which includes appropriate in-process inspection points, and testing activities to prevent workmanship defects and assures delivery of a high reliable end product.

Quality Assurance Requirements Assumptions Prior to Fabrication Vendor Survey A vendor survey had been performed and all open items had been closed.

Facility and Work Stations Facility (Clean Room) The general assembly and soldering area shall have a controlled environment, which limits the entry of contaminations. The temperature and humidity in the soldering area shall be monitored and maintained within the comfort zone, as shown in Figure 10-23. The enclosed soldering facility will maintain a positive pressure, unless the soldering area is not in an air-conditioned, clean room. 90°F 30°C

80°F 2 25°C

ex S 70°F 20°C

60°F

20

40 60 Relative Humidity, (%)

Figure 10-23. Temperature and Humidity in the Soldering Area. Lighting The lighting at the working surface for soldering and solder pot operations shall have a minimum illumination of 100 foot-candles.


Handling Parts Prior to handling parts and/or materials, the operator shall thoroughly clean his/her hands; the use of any hand lotion is forbidden. Anyone working or handling parts and/or materials must wear clean gloves and/or finger cots. Gloves must be changed when they show signs of contamination, and finger cots must be replaced when they are torn or contaminated. Work Area The work areas and workbenches shall be maintained in a clean and orderly manner. At the start of each workday, the work stations shall be free of visible dirt, grime, grease, flux or solder splatters, and other foreign materials. Restrictions There will be no smoking, eating, or drinking permitted at the work stations. Cosmetics Hand cream, ointments, perfumes, cosmetics, and other materials unessential to the fabrication operation shall not be permitted at the work station. ESP Protection Requirement Supplier shall establish and maintain a documented program for the control of Elect-Static Discharge, (ESD), during fabrication and handling of such devices. The program shall comply with the requirements of MIL-STD- 1686. Certified Personnel All certified personnel must have up-to-date, valid training certificates before fabrication can begin. In-House Fabrication Procedures All fabrication drawings and procedures must be signed off by the cognizant engineer before work can begin. There shall no red line drawings in the magnetic component assembly area. Purchase Order The purchase order or contract has to be issued between the company and vendor, before any parts are ordered or the beginning of fabrication. The purchase order, or contract, defines the test that will be performed, the manufacturing, and the quality assurance requirements. Fabrication Review A fabrication review will have been conducted between the company and the vendor to assure that the vendor is ready to begin fabrication and testing.


Engineering Model (EM) or Prototype After the engineering model, (EM), has been tested for form, fit, and function, a review panel that will involve the vendor and the Company Quality Assurance, and the cognizant engineer will be setup to access the magnetic component for practicality, reliability, and fabrication.

Before the Start of Fabrication Procedures Quality Assurance personnel shall review, inspect, and give their concurrence on: (a) assembly drawings, (b) test procedures, (c) potting procedures, (d) inspections, (travelers), and (e) shipping. Materials Quality Assurance personnel shall review, inspect, and issue a Part Acceptance Tag (PAT tag), as shown in Figure 10-24, on all materials such as: (a) wire, both magnet and insulated, (b) insulation material, (c) magnetic cores, (d) enclosures, (e) terminals, and (f) solder type. Equipment Quality Assurance personnel shall review, inspect, and give their concurrence on: (a) the winding machine, (b) the tension device, (c) the soldering iron, (d) the solder pot, (e) hand tools, and (f) aids.

Part Acceptance Tag

No. 35002

Part Number

Revision

Lot Number

P.O./W.O. Number

Inspection Report No.

Cert. Number

Supplier

Quantity

Date Received

Cert. Number

Date Inspected

Inspection Stamp

Figure 10-24. Typical, Quality Assurance, Part Acceptance Tag.

Documentation Materials Certification Manufacturers of the materials shall supply certification of conformance for the required and applicable specification. Traceability 100% traceability of all parts and materials shall be maintained throughout the process from the receiving, or source inspection, to the final tests.


Reverse-Traceability The information content of each document shall be sufficient to provide reverse-traceability. Manufacturing and Inspection Records All manufacturing and inspection checks shall be recorded on an approved, (Traveler). See Figures 10-25 and 10-26. The approved fabrication instructions will accompany each deliverable item, which will provide an accurate history of the part. Deliverable Package A documentation package shall be maintained for each deliverable piece of electronic equipment, and will include approved fabrication instructions, inspection reports, deviation reports, and all Material Review Board, (MRB), evaluations. This package will also include a Certificate of Compliance, serialized test data, and the traceability information.

In-Process Inspection In-Process Inspection The Company Quality Assurance personnel shall set up mandatory in-process inspection points after the vendor supplies assembly and test flow charts. Discrepancies Any discrepancies, with respect to the specification, drawing or inspection standards, defined in the contract, shall be written up on an Inspection Report, (IR). The, (IR), will be submitted to the company, cognizant engineer for disposition. Parts, that have been written up on an Inspection Report, (IR), will be assessed for impact to form fit or function. Common Problems In the fabrication of magnetic components, the most common problems found over the years are: (a) cold solder joints, (b) nicked magnet wire, (c) magnet wire lead dressing, and (d) magnet wire lead fatigue.

Unit Specification Verification Test Demonstration Verification, by testing, is accomplished by subjecting the magnetic component to a set of conditions under the control of the approved test plan, procedures, and test equipment which will provide the accurate test data. The results of the test are compared with the specification control drawings.


Test Discrepancies Any discrepancies in the test results, when compared to the specification requirements, shall be written up by the company Quality Assurance. This information will be submitted to the cognizant, company engineer for disposition. The parts will be assessed for impact to form fit or function.

Visual Inspection The magnetic component shall be measured/inspected to verify that the construction, the physical dimensions, the correct markings, cleanliness, and the workmanship are in accordance with the specification, control drawings.

Traveler-Transformers, Inductors and Coil Assemblies (Front) Assembly No. Drawing No. & Rev. Serial No.

Prograrr Machine Specific ation No.

Material Part Number

IR/PAT

Type

Tech

Date

QA

Core Bobbin / Tube Wire Hook Up Tape Adhesive Tape Cloth Poly Shielding Banding Strap Seal Strap Air Gap Material Mylar Housing Terminal Board Sleeving Remarks Figure 10-25. Typical, Transformer, Inductor Inspection Traveler Card, (Front).


Date

Traveler-Transformers, Inductors and Coil Assemblies (Back) Wire AWG

*Test Tech. IR/PAT Turns I 2 3 4 5 6 7 8 9 10 Inspection Prior to Soldering Solder Wires and Inspect Electrical Test Encapsulation Serial No. Marking Part No. Assembly No. A. Magnetizing Current B. Turns Ratio Test to Perform Winding Number

Date

QA

C. See Winding Specification

Figure 10-26. Typical, Transformer, Inductor Inspection Traveler Card, (Rear).


Date

High Reliability Magnetic Devices (Electrical Engineering & Electronics, 115)

High Reliability Magnetic Devices

Electrical devices

Advanced Electrical and Electronics Engineering: Volume 2

Radio Receiver Design (Electrical Engineering & Electronics) (Electrical and Computer Engineering)

Electrical Engineering: Problems & Solutions

Electrical, Optical, Magnetic Properties

Engineering reliability

Engineering Reliability

Engineering Electronics

Electronics Engineering

High Magnetic Fields

Electrical Engineering

Electrical Engineering

Vacuum Electronics: Components and Devices

High-speed heterostructure devices

High Speed Heterostructure Devices

Hydrogen Power Electricity Electrical Electronics

The Electronics Handbook, Second Edition (Electrical Engineering Handbook)

Engineering Design Reliability Handbook

Reliability Physics and Engineering

Engineering Design Reliability Handbook

Life Cycle Reliability Engineering

Handbook of Reliability Engineering

Circuits, devices, and systems : a first course in electrical engineering

Handbook of reliability engineering

Wiley Encyclopedia of Electrical and Electronics Engineering, 24 Volume Set

High Voltage Circuit Breakers: Design and Applications (Electrical Engineering and Electronics, Vol 100)

Power-Switching Converters, Second Edition (Electrical Engineering and Electronics)

Handbook of Reliability Engineering

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High Reliability Magnetic Devices (Electrical Engineering & Electronics, 115)