Electronic Circuit Design: From Concept to Implementation

ELECTRONIC CIRCUIT DESIGN From Concept to Implementation

Certain materials contained herein are reprinted with the permission of Microchip Technology Incorporated. No further reprints or reproductions may be made of said materials without Microchip Technology Inc.’s prior written consent. CRC Press Taylor & Francis Group 6000 Broken Sound Parkway NW, Suite 300 Boca Raton, FL 33487-2742 © 2008 by Taylor & Francis Group, LLC CRC Press is an imprint of Taylor & Francis Group, an Informa business No claim to original U.S. Government works Printed in the United States of America on acid-free paper 10 9 8 7 6 5 4 3 2 1 International Standard Book Number-13: 978-0-8493-7617-7 (Hardcover) This book contains information obtained from authentic and highly regarded sources. Reasonable efforts have been made to publish reliable data and information, but the author and publisher cannot assume responsibility for the validity of all materials or the consequences of their use. The authors and publishers have attempted to trace the copyright holders of all material reproduced in this publication and apologize to copyright holders if permission to publish in this form has not been obtained. If any copyright material has not been acknowledged please write and let us know so we may rectify in any future reprint. Except as permitted under U.S. Copyright Law, no part of this book may be reprinted, reproduced, transmitted, or utilized in any form by any electronic, mechanical, or other means, now known or hereafter invented, including photocopying, microfilming, and recording, or in any information storage or retrieval system, without written permission from the publishers. For permission to photocopy or use material electronically from this work, please access www.copyright.com (http://www.copyright.com/) or contact the Copyright Clearance Center, Inc. (CCC), 222 Rosewood Drive, Danvers, MA 01923, 978-750-8400. CCC is a not-for-profit organization that provides licenses and registration for a variety of users. For organizations that have been granted a photocopy license by the CCC, a separate system of payment has been arranged. Trademark Notice: Product or corporate names may be trademarks or registered trademarks, and are used only for identification and explanation without intent to infringe. Library of Congress Cataloging-in-Publication Data Kularatna, Nihal. Electronic circuit design : from concept to implementation / Nihal Kularatna. p. cm. Includes bibliographical references and index. ISBN 978-0-8493-7617-7 (alk. paper) 1. Electronic circuit design. I. Title. TK7867.K78 2008 621.3815--dc22 Visit the Taylor & Francis Web site at http://www.taylorandfrancis.com and the CRC Press Web site at http://www.crcpress.com

2007048859

This book is dedicated to Sir Arthur C. Clarke, who departed us recently.

With loving thanks to my wife Priyani, and the two daughters, Dulsha and Malsha, who always tolerate my addiction to tech-writing and electronics.

Contents Preface...........................................................................................................ix About the Author.........................................................................................xi Acknowledgments..................................................................................... xiii Contributors.............................................................................................. xvii 1.

Review of Fundamentals.................................................................... 1

2.

Design Process................................................................................... 57 Shantha Fernando

3.

Design of DC Power Supply and Power Management................. 77

4.

Preprocessing of Signals................................................................ 205

5.

Data Converters............................................................................... 277

6.

Configurable Logic Blocks for Digital Systems Design............ 343 Morteza Biglari-Abhari

7.

Digital Signal Processors............................................................... 363

8.

An Introduction to Oscillators, Phase Lock Loops, and Direct Digital Synthesis................................................................. 413 Sujeewa Hettiwatte, Coauthor

9.

System-on-a-Chip Design and Verification................................. 427 Chong-Min Kyung

Index........................................................................................................... 469

vii

Preface Electronics is probably one of the few subjects where the “knowledge halflife” of a professional is very short. Today it is probably only 3 to 4 years. Designing electronic systems today requires a unique combination of (1) fundamentals; (2) research and development directions in the latest semiconductors and passive components; (3) nitty-gritty aspects within the “mixed signal world;” (4) access to component manufacturers’ data sheets, design guidelines, and development environments; and, most importantly, (5) a timely and practical approach to overall aspects of a design project. Classen’s rule—“the usefulness of a product is proportional to log (technology)”—reminds us that engineers have to add a lot of technology and engineering teamwork to get more user-friendliness into electronics. With portability and miniaturization becoming buzzwords in electronics, design engineers have to concentrate on many additional aspects in the core of a design. Some of these are power supply design, packaging, thermal design, and reliability. In the new millennium, with very large-scale integrated (VLSI) circuits and system-on-a-chip (SOC) technologies maturing, designers have many options when designing electronic systems. My own career of more than 32 years in different environments such as aviation, telecommunications, and power electronics, with a long mid­career in research and development, has given me the opportunity to look at the world of electronics in a broader perspective. A budding electronics engineer with a degree-level qualification takes a few years to appreciate the breadth of the subject while learning the depth in limited specialized areas. The breadth and depth of the subject together are necessary to produce a commercially viable product or system, giving consideration to time to market (TTM). This work attempts to address several areas of analog and mixed signal design, including power supply design, signal conditioning, essentials of data conversion, and signal processing, while summarizing a large amount of information from theory texts, application notes, design bulletins, research papers, and technology magazine articles. In a few chapters I had the assistance of experts in different subject areas, as chapter authors. Because of page limitations, I had to summarize a large amount of useful subject matter extracted from more than 500 technical publications. I suggest that readers refer to the cited references for more details. I would also appreciate your assistance in notifying me of any errors found in the book. Nihal Kularatna Auckland New Zealand ix

About the Author Former CEO of the Arthur C. Clarke Institute for Modern Technologies (ACCIMT) in Sri Lanka, Nihal Kularatna is an electronics engineer with more than 30 years of experience in professional and research environments. He is the author of two Electrical Measurement Series books for the IEE (London), titled Modern Electronic Test and Measuring Instruments (1996) and Digital and Analogue Instrumentation: Testing and Measurement (2003), and two Butterworth (USA) titles, Power Electronics Design Handbook: Low Power Components and Applications (1998) and Modern Component Families and Circuit Block Design (2000). He coauthored Essentials of Modern Telecommunications Systems for Artech House Publishers (April 2004). From 1976 to 1985 he worked as an electronics engineer responsible for navigational aids and communications projects in civil aviation and digital telephone exchange systems. In 1985 he joined the ACCIMT as a research and development engineer and earned a principal research engineer status in 1990; he was appointed CEO/Director in 2000. From 2002 to 2005 he was a senior lecturer at the Department of Electrical and Electronic Engineering, University of Auckland, New Zealand. He has participated in many specialized training programs with equipment manufacturers, universities, and other organizations in the United States, United Kingdom, France, and Italy. He was an active consultant for two U.S. companies, including the Gartner Group, and many Sri Lankan organizations. He is currently a member of the expert reviewer panel of the Foundation for Research, Science and Technology, New Zealand. He is currently active in research in transient propagation and power conditioning in power electronics, embedded processing applications for power electronics, and smart sensor systems. He has contributed more than 60 papers to academic and industry journals and international conference proceedings. He was the principal author of the McGraw-Hill (Datapro) report “Sri Lanka Telecoms—An Industry and Market Analysis” (1997). A Fellow of the IEE (London), a Senior Member of IEEE (USA), and an honors graduate from the University of Peradeniya, Sri Lanka, during his research career in Sri Lanka, he was the winner of a Presidential Award for Inventions (1995), Most Outstanding Citizens Award (1999, Lions Club), and

xi

xii

About the Author

a TOYP Award for academic accomplishment (Jaycees) in 1993. He is a fellow of the Institution of Engineers, Sri Lanka. He is currently employed as a senior lecturer in the Department of Engineering, University of Waikato, New Zealand. His hobby is gardening cacti and succulents.

Acknowledgments Since my graduation in 1976 from the University of Peradeniya, Sri Lanka, I have spent more than 30 years in electronics and associated fields. Since 2002 I’ve had a full-time university teaching career in New Zealand, which has given me the time to think of the differences between industry and academia. My 16-year career (1985–2002) at the Arthur C. Clarke Institute for Modern Technologies (ACCIMT) in Sri Lanka gave me the opportunity to work closely with engineers from other countries where electronics technology has had rapid progress. Colleagues, project partners, students, friends and family always encouraged me to be closely involved with the world of electronics and enjoy the opportunities. There are too many people to mention by name, but I thankfully acknowledge them all with a very grateful heart. This book attempts to provide a reasonable link between the theoretical knowledge domain and the valuable practical information domain from the technology developers. The broad approach in this work is to understand the complete systems and appreciate their interfacing aspects, embracing many digital circuit blocks coupled with the mixed signal circuitry in complete systems. A large amount of published material from industry and academia has been used in this book. The work and organisations that deserve strong acknowledgments are:

1. Many published text books for the material in Chapter 1. 2. Many published articles in the Power Electronics Technology magazine (Penton Media) for Chapter 3. 3. Analog Devices, Inc., for a large amount of material in Chapters 4 and 5. 4. Industry magazines such as EDN, Electronic Design, Test & Measurement World and many IEEE/ IEE publications.

I am very thankful to the tireless attempts by the following for creating figures, word processing, etc.:



1. Chandrika Weerasekera, and Jayathu Fernando, ACCIMT staff members from Sri Lanka, who spent a lot of their private time coordinating with me across different time zones. 2. Pawan Shestra and David Nicholls, from our own technical staff at the University of Waikato, and Heidi Eschmann from the department administration. xiii

xiv

Acknowledgments

3. Postgraduate student Zhou Weiqian and undergraduate student Ben Haughey. 4. Thiranjith Weerasinghe, Kasun Talwatte, and Dilini Attanayake, my friends and neighbours. 5. Dulsha, Malsha, and Rajith, from the family.

Without the above persons it would have been impossible for me to deal with writing the 5½ chapters of the book and the overall manuscript preparation. My special gratitude is extended to my chapter authors Shantha Fernando (Shantha, after many years since our ACCMT days you made me feel part of a good team work again across the Tasman sea!), Morteza Biglari-Abhari, Sujeewa Hettiwatte and Prof. Chong-Min Kyung of KAIST, Korea, for their great team work with regards to the contents of the book. I am also very indebted to our postgraduate student Chandani Jinadasa for her assistance in checking my proof reading comments. For the copyright permission for certain contents in the book I am very thankful to the following:

1. David Morrison, Editor-in-Chief of the Power Electronics Technology Magazine, Penton Media, USA 2. Jon Titus (formerly with EDN Magazine), Kasey Clark and Maury Wright of the Editorial Group of the EDN Magazine 3. Mark David of Electronic Design magazine, USA 4. Rich Fassler of Power Integrations Inc., USA 5. Steve West of On Semiconductor, USA 6. John Hamburger of Linear Technology Inc., USA 7. Bill Hohl and Katherine Souter of ARM Limited, UK 8. Mike Phipps and Hamish Rawnsley of Altera 9. Bill Hutchins, Jaqueline Eischmann and Carol Popvich of Microchip Technologies, USA 10. Yvette Huygen of Synopsys Inc., USA 11. James Bryant from Analog Devices Inc, European Headquarters. 12. Deborah Sargent and Rick Nelson of Test and Measurement World Magazine, USA 13. Joseph P. Hayton and colleagues of Elsevier Science & Technology Books

It was an extremely pleasant experience to work with the staff of the CRC Press on this book, which occurred during a very busy time for my family due to my daughter’s wedding. I am particularly thankful to publisher Nora Konopka for her understanding and the support to get the project

Acknowledgments

xv

moving smoothly. Jessica Vakili, Katherine Colman, and the other members of ­Editorial are gratefully acknowledged for their support in collecting my manuscript in several stages, allowing me to balance my time between work, family and book writing, particularly during our stay in Sri Lanka for the wedding. I am very grateful to Robert Sims and the Production staff for their assistance in solving difficult problems during the production of the work. I am very glad to mention that the Editorial and Production group of the CRC Press is a very understanding team, and it was a great pleasure to work with them. Since moving to Hamilton, New Zealand, to work at the University of Waikato, I am privileged to work with a very friendly team of colleagues who give me lots of encouragement for my work. I particularly appreciate the spectacularly beautiful area in New Zealand and the friendly kiwi atmos­ phere in which I have been able to complete the final stages of this book. To Sir Arthur Clarke, Patron of the Clarke Institute, I was inspired by you during the 16 years of my work at the ACCIMT. Thank you so much for your encouragement in my work. Last, but not least, my special thanks go to Priyani, who made this work possible by taking over my family commitments ­—looking after the family needs and taking the full responsibility of planning a wedding in Sri Lanka. Nihal Kularatna November 2007

Contributors Dr. Morteza Biglari-Abhari  Department of Electrical and Computer Engineering, University of Auckland, Auckland, New Zealand Shantha Fernando  Advanced Technology Centre NSW Police Force, Australia Dr. Sujeewa Hettiwatte  Department of Electrical and Electronic Engineering, School of Engineering, Auckland University of Technology, Auckland, New Zealand Prof. Chong-Min Kyung  Department of Electrical Engineering and Computer Science, Korea Advanced Institute of Science and Technology, Daejeon, Republic of Korea

xvii

1 Review of Fundamentals Contents 1.1 Introduction..................................................................................................... 2 1.2 Ohm’s Law, Kirchoff’s Laws, and Equivalent Circuits.............................. 3 1.3 Time and Frequency Domains...................................................................... 3 1.3.1 The Fourier Transform....................................................................... 5 1.4 Discrete and Digital Signals.......................................................................... 6 1.4.1 Discrete Time Fourier Transform..................................................... 7 1.4.2 Discrete Fourier Transform............................................................... 7 1.4.3 Fast Fourier Transform....................................................................... 8 1.5 Feedback and Frequency Response............................................................. 9 1.5.1 Gain Desensitization........................................................................ 10 1.5.2 Noise Reduction................................................................................ 10 1.5.3 Reduction in Nonlinear Distortion, Bandwidth Extension, and Input/Output Impedance Modification by Feedback......... 11 1.6 Loop Gain and the Stability Problem........................................................ 13 1.6.1 The Nyquist Plot............................................................................... 17 1.6.2 Poles and Zeros, S-Domain, and Bode Plots................................. 17 1.6.3 Bode Plots and Gain and Phase Margins...................................... 20 1.7 Amplifier Frequency Response..................................................................22 1.7.1 BJT Equivalent Circuits....................................................................22 1.7.2 BJT Small Signal Operation and Models....................................... 26 1.7.3 High-Frequency Models of the Transistors and the Frequency Response of Amplifiers................................................ 26 1.7.3.1 Low-Frequency Response................................................. 28 1.7.3.2 High-Frequency Response................................................ 30 1.7.3.3 Use of Short-Circuit and Open-Circuit Time Constants for the Approximate Calculations of ωL and ωH.................................................................................. 30 1.8 Transistor Equivalent Circuits, Models, and Frequency Response of Common Emitter/Common Source Amplifiers................................... 32 1.8.1 Calculation of the Low-Frequency 3-dB Corner Frequency, ωL......................................................................................................... 32 1.8.2 Calculation of the High-Frequency 3-dB Corner Frequency, ωH....................................................................................34

1

2

Electronic Circuit Design: From Concept to Implementation

1.9 Noise in Circuits........................................................................................... 39 1.9.1 Noise in Passive Components......................................................... 39 1.9.2 Effect of Circuit Capacitance........................................................... 41 1.9.3 Noise in Semiconductors and Amplifiers..................................... 41 1.9.4 Circuit Noise Calculations and Noise Bandwidth.......................44 1.9.5 Noise Figure and Noise Temperature............................................ 47 1.10 Passive Components in Circuits................................................................. 49 1.10.1 Resistors............................................................................................. 50 1.10.2 Capacitors........................................................................................... 51 1.10.3 Inductors............................................................................................ 53 1.10.4 Passive Component Tolerances and Worst-Case Design............ 53 References............................................................................................................... 55

1.1 Introduction Circuit design can be considered an art based on the fundamental concepts we learn in electrical and electronic engineering. With the unprecedented advancement of semiconductors, today a designer has many choices of components. Although keeping track of all the new integrated circuits (ICs) appearing on the market is a difficult task, it may be particularly useful if design challenges include miniaturization of the overall product. Passive components such as resistors, capacitors, inductors, and transformers need to be mixed effectively and optimally with semiconductor components in building a particular circuit. In this exercise of “design and development,” the designer needs to work with the delicate balance between the real or the analog world, where signals can take any value within a given range, and the digital world, where we make use of processors, memories, and other peripheral devices to accurately process information. To summarize the need for the delicate balance required, it may be appropriate to cite Jim Williams, a well-known linear circuit designer: “Wonderful things are going on in the forgotten land between ONE and ZERO. This is real electronics.” Within the past quarter century, a new domain of semiconductors has appeared that links the analog and digital worlds. That is the world of mixed signal electronics, where analog-to-digital conversion, and vice versa, occurs. In this book, emphasis is placed on the analog world of electronics together with the mixed signal domain of design. In dealing with the challenges we face in the process of design and development, a few essential fundamentals need to be reviewed. This chapter reviews the essentials so that designers can comfortably link theory and practice. The reader can find details related to theory and analysis in standard textbooks used for undergraduate and postgraduate courses.

3

Review of Fundamentals Rs vs(t)

+ – (a)

is(t)

Rs (b)

Figure 1.1 Useful forms of equivalent circuits: (a) Thevenin form; (b) Norton form.

1.2 Ohm’s Law, Kirchoff’s Laws, and Equivalent Circuits Ohm’s law relates the voltage and current in a circuit element with the familiar relationship V = IR. From this relationship, and then applying Kirchoff’s laws to more complex circuits, a design engineer has a precise tool set to analyze a complete circuit with both active and passive components. Deriving from these basic laws the most commonly used equivalent circuits, such as the Thevenin and Norton forms, allows us to simplify many complex circuit blocks, provided that we use reasonable assumptions to simplify each case. For example, a transducer such as a microphone may be simplified as a voltage source in series with a resistor, giving us the familiar Thevenin or Norton form as in Figure 1.1. Similarly, a regulated direct current (DC) power supply may be represented by its Thevenin equivalent to explain the essential characteristics defined in its specifications (discussed later in Chapter 2). One essential reminder regarding the use of Thevenin’s equivalent is that it does not allow us to calculate the dissipation within a circuit. Another is that if we replace the voltage or current sources with their internal resistances to calculate the equivalent resistance, Rs, these sources need be independent ones. Another reminder is to take care when calculating Rs in circuits where dependent voltage or current circuits exist.

1.3 Time and Frequency Domains All electrical signals can be described as a function of either time or frequency. When we observe signals as a function of time, they are called time domain measurements. Sometimes we observe the frequencies present in signals, in which case they are called frequency domain measurements. The word spectrum refers to the frequency content of any signal. All practical components we use as building blocks perform according to design specifications only within a limited frequency range. Therefore, we can define a

4

Electronic Circuit Design: From Concept to Implementation

bandwidth for each circuit block and an overall bandwidth of operation for the entire product. When signals are periodic, time and frequency are simply related; namely, one is the inverse of the other. Then we can use the Fourier series to find the spectrum of the signal. For nonperiodic signals, a Fourier transform is used to obtain the spectrum. However, performing a Fourier transformation involves integration over all time, that is, from −∞ to +∞. Because this is not practicable, we approximate the Fourier transform by a discrete Fourier transform (DFT), which is performed on a sampled version of the signal. The computational load for direct DFT, which is usually computed in a processor subsystem, increases rapidly with the number of samples and sampling rate. As a way around this problem, Cooley and Tukey invented the fast Fourier transform (FFT) algorithm in 1954. With the FFT, computational loads are significantly reduced. Let f (t) be an arbitrary function of time. If f (t) is also periodic, with a period T, then f (t) can be expanded into an infinite sum of sine and cosine terms. This expansion, which is called the Fourier series of f (t), may be expressed in the form

a f t = 0+ 2

()

∞











∑a cos  2πTnt  + b sin  2πTnt  , 0

n

(1.1)

n=1

where Fourier coefficients an and bn are real numbers independent of t and which may be obtained from the following expressions:





2 an = T

2 bn = T

T









∫ f (t) cos  2πTnt  dt,  where n = 0, 1, 2, … ,

(1.2a)

0

T

∫ f (t) sin  2πTnt  dt,  where n = 0, 1, 2, … .

(1.2b)

0

Further, n = 0 gives us the DC component, n = 1 gives us the fundamental, n = 2 gives us the second harmonic, and so on. Another way of representing the same time function is

()

f t =

a0 + 2

∞





∑ d cos  2πTnt + φ  , n

n

(1.3)

n=1

where dn = an2 + bn2  and tan ϕn = −bn/an. The parameter dn is the magnitude and ϕn is the phase angle.

5

Review of Fundamentals Equation (1.3) can be rewritten as +∞

()

f t =



∑c e n

j 2πnt/T

,

(1.4)

n=−∞

where

1 cn = T

T

∫ f (t ) e

− j 2πnt/T

dt  and n = 0, ±1, ±2.

0

The series expansion of Equation 1.4 is referred to as the complex exponential Fourier series. The cn are called the complex Fourier coefficients. According to this representation, a periodic signal contains all frequencies (both positive and negative) that are harmonically related to the fundamental. The presence of negative frequencies is simply a result of the fact that the mathematical model of the signal given by Equation 1.4 requires the use of negative frequencies. Indeed, this representation also requires the use of complex exponential functions, namely ej2πnt/T, which have no physical meaning either. The reason for using complex exponential functions and negative frequency components is merely to provide a complete mathematical description of a periodic signal, which is well suited for both theoretical and practical work. 1.3.1 The Fourier Transform The transformation from the time domain to the frequency domain and back again is based on the Fourier transform and its inverse. When the arbitrary function f (t) is not necessarily periodic, we can define the Fourier transform of f (t) as

( )

F f =

∞

∫ f (t ) e

− j 2πft

dt.

(1.5)

−∞

The time function f (t) is obtained from F( f  ) by performing the inverse Fourier transform:

()

f t =

∞

∫ F( f )e

j 2πft

df .

(1.6)

−∞

Thus, f (t) and F( f  ) form a Fourier transform pair. The Fourier transform is valid for both period and nonperiod functions that satisfy certain minimum conditions. All signals encountered in the real world easily satisfy these

6

Electronic Circuit Design: From Concept to Implementation Waveforms

Time Domain

Frequency Domain

Sinewave

Triangle

Sawtooth

Rectangle

Pulse

Random noise

Bandlimited noise

Random binary sequence

Figure 1.2 Signal observation with oscilloscope and spectrum analyzer.

conditions. These conditions, known as the Dirichlet conditions, determine whether a function is Fourier expandable [1]. An oscilloscope displays the amplitude of a signal as a function of time, whereas a spectrum analyzer represents the same signal in the frequency domain. The two types of representations (which form Fourier transform pairs) are shown in Figure 1.2 for some common signals encountered in practice.

1.4 Discrete and Digital Signals A discrete signal is either discrete in time and continuous in amplitude, or discrete in amplitude and continuous in time. Discrete signals can occur in

7

Review of Fundamentals

charge-coupled device (CCD) arrays or switched capacitor filters. A digital signal, however, is discrete in both time and amplitude, such as those signals encountered in digital signal processing (DSP) applications. 1.4.1 Discrete Time Fourier Transform The discrete time Fourier transform (DTFT) maps a discrete time function h[k] into a complex function H(e jω), where inverse transforms are ∞

( ) = ∑ hk e



H e

jω

− jkω



(1.7)

k =−∞

1 h k  = 2π



π

∫ H (e ) dω . jω

(1.8)

−π

Note that the inverse transform, as in the Fourier case, is an integral over the real frequency variable. A complex inversion integral is not required. The DTFT is useful for manual calculations but not for computer calculations because of continuous variable ω [2]. The role of the DTFT in discrete time system analysis is very much the same as the role the Fourier transform plays for continuous time systems. 1.4.2 Discrete Fourier Transform An approximation to the DTFT is the discrete Fourier transform (DFT). The DFT maps a discrete time sequence of N-point duration into an N-point sequence of frequency domain values. Formally, the defining equations of the DFT are [2]





 j 2 πn  H  e N  = H n =   1 h k  = N

N −1

N −1

∑

h k  e

−j

2 πnk N

 for k, n = 0, 1, … , N − 1

(1.9)

k =0

∑ H n e

j

2πnk N

 for k, n = 0, 1, … , N − 1.

(1.10)

n=0

These equations are the DTFT relationships with the frequency discretized to N points spaced 2π/N radians apart around the unit circle, as in Figure 1.3. In addition, the time duration of the sequence is limited to N points. The DFT uses two finite series of N points in each of the definitions. Together they form the basis for the N points DFT. The restriction to N points

8

Electronic Circuit Design: From Concept to Implementation lm(z)

2π/N Re(z)

Figure 1.3 Unit circle with N = 8.

and reduction from integral to summation occur because we are only evaluating the DTFT at N points in the Z domain. The value of N is usually determined by constraints in the problems or the resources for analysis. The most popular values of N are in powers of 2. However, many algorithms do not require this. N can be almost any integer value [2]. 1.4.3 Fast Fourier Transform The FFT is not a transform but an efficient algorithm for calculating the DFT. The FFT algorithms remove the redundant computations involved in computing the DFT of a sequence by exploiting the symmetry and periodic nature of the summation kernel, the exponential term, and the signal. For large values of N, FFTs are much faster than performing direct computation DFTs. Table 1.1 shows the approximate computation loads for various sizes of directly calculated DFTs and the DFT computed using the FFT decimationin-time algorithm. Note that as N grows beyond 16, the savings are quite dramatic [2]. For details on practical FFT algorithms, in particular, the decimation-intime algorithm, see Lynn and Fuerst [3]. In modern digital storage scopes, the FFT function is a common feature, and most of these techniques and algorithms are used within the processor subsystem of the scope (see Kularatna [4], Chapters 6 and 9).

9

Review of Fundamentals Table 1.1 Comparison of Approximate Computational Loads for Direct DFT and Decimations–in-Time FFT Number of Points 4 8 16 32 64 128 256 512 1024 2048 4096

Multiplications DFT 16 64 256 1024 4096 16,384 65,536 262,144 1,048,576 4,194,304 16,777,216

FFT 4 12 32 80 192 448 1024 2304 5120 11,264 24,576

Additions DFT 12 56 240 992 4032 16,256 65,280 261,632 1,047,552 4,192,256 16,773,120

FFT 8 24 64 160 384 896 2048 4608 10,240 22,528 49,152

1.5 Feedback and Frequency Response Harold Black, an electronics engineer with Western Electric, invented the feedback amplifier in 1928. Since then, the technique has been so widely used that it is almost impossible to think of electronic circuits without some form of feedback. Feedback can be either negative or positive. In amplifier design, negative feedback is commonly used to achieve one or more of the following: • Desensitize the gain—make the gain less sensitive to variations in the value of circuit components. • Reduce nonlinear distortion—make the output gain constant and independent of the signal level. • Reduce the effect of noise—make improvements to the signal-tonoise ratio (SNR). • Control the input/output impedances. • Extend the bandwidth. Figure 1.4, which is a signal flow diagram where x indicates either voltage or current signals, indicates the concept of applying feedback to an amplifier. With the addition of the feedback block, the basic amplifier with an openloop gain of A now receives a modified input signal xi = xs − xf . The feedback

10

Electronic Circuit Design: From Concept to Implementation

Source

xs + –

xi

Σ

A

xo

Load

xf β Figure 1.4 General structure of the feedback amplifier.

block with a feedback factor of β makes xf = βxo, where xi is the input into the open-loop amplifier, xo is the output of the open-loop amplifier, and xf is the feedback signal. With the assumptions that the transmission of the signals through the basic amplifier is only in the forward direction and the transmission through the feedback block is only in the backward direction, it can be shown that the overall gain of the feedback system Af is given by the relationship

Af =

xo A . = xs 1 + Aβ

(1.11)

Quantity Aβ is called the loop gain and quantity (1 + Aβ) is called the amount of feedback. In a practical design, open-loop gain Aβ is large, Aβ >> 1, then from Equation 1.11, it follows that Af ≈ 1/β, which is a very practical and useful property in design, because the close loop gain is now entirely dependent on the feedback network. 1.5.1 Gain Desensitization From the above discussion, we can further derive many useful attributes of the feedback system in Figure 1.4. For example, we can show that the percentage change in Af (due to variations in some circuit parameters) is smaller than the percentage change in A by the amount of feedback (1 + Aβ). This is given by the relationship

dA f 1 dA = . (1 + Aβ) A Af

(1.12)

1.5.2 Noise Reduction Negative feedback can be employed to reduce the noise or interference in an amplifier and hence improve the SNR. This situation is summarized in Figure 1.5. If we refer the total noise in the amplifier to its input by a noise source of value Vn, we can show that the SNR is given by Vs/Vn for the case in Figure 1.5a. Now, if the system is modified by adding a theoretically

11

Review of Fundamentals

Vs + –

+–

Vn

+ Vo –

A (a) Vn

Vs

–

Ap

∑

∑

A

Vo

β (b) Figure 1.5 Concept of increasing the SNR of a circuit: (a) basic amplifier indicating the case of input referred noise; (b) with an additional noise-free amplifier at the front end for SNR improvement.

noise-free amplifier with gain Ap and a feedback block with a feedback factor of β, the total output Vo is given by Figure 1.5b:

Vo = Vs

AAp A + Vn . 1 + AApβ 1 + AApβ

(1.13)

Thus, the modified SNR becomes

SNRnew =

VS Ap . Vn

(1.14)

This is an improvement by a factor of Ap. For details, see Sedra and Smith [5]. 1.5.3 Reduction in Nonlinear Distortion, Bandwidth Extension, and Input/Output Impedance Modification by Feedback Another useful application of feedback is the linearization of transfer characteristics of an amplifier. For a discussion on this, see Chapter 8 in Sedra and Smith [5]. Similarly, for a low-pass filter, by applying feedback, the bandwidth of the circuit can be increased at the expense of a lower midband gain. To illustrate the case, consider a single-pole amplifier with a midband gain of A M and a 3-dB upper cutoff frequency of ωH, where the transfer function is given by

A( s) =

AM . 1 + s / ωH

(1.15)

12

Electronic Circuit Design: From Concept to Implementation

With a feedback circuit with a feedback factor of β added to the circuit, commencing from the relationships in Equation 1.11 and Equation 1.15, we can show that the new transfer function is given by A f ( s) =



AM /(1 + AMβ) . 1 + s / ω H (1 + AMβ)

(1.16)

This simply illustrates the basis for increasing or decreasing the bandwidth at the expense of decreasing or increasing the midband gain. Similar discussions are possible for other types of amplifiers [5]. Also, it reminds us that the gain bandwidth product for a single-pole low-pass circuit is constant. Feedback can be utilized to modify the input and output impedances of amplifier stages. At this point it may be appropriate to discuss the four common types of amplifiers: voltage, current, transconductance, and trans­ resistance. To ease the design process, for each case we can insert a feedback stage and suitable practical simplifications such as (1) an open-loop amplifier, which acts as a unilateral circuit (where no reverse feedback occurs through the amplifier) and (2) a β circuit (no forward transfer of signals occurs from the input to output side of the amp through the β circuit). Figure 1.6 shows the four topologies with the feedback networks. We give different names in each case, depending on the way we mix and sample the signals at the input and output sides, respectively. For example, in the case of a voltage amplifier, we

+ Vs –

Rs

– Vf +

Basic voltage amplifier

+

RL

Vo –

Is

Basic current amplifier

Rs Is

Is 1

Feedback network

2

Rs Vs + –

2

(b)

– Vf +

Basic transconductance amplifier

Io

RL

1

Feedback network

If 2

+

Basic transresistance amplifier

Rs

Is

Io

(c)

RL

Io Feedback network

1

(a)

Io

RL

Vo –

If 1

Feedback network

2

(d)

Figure 1.6 The four basic feedback topologies: (a) voltage-sampling series mixing (series-shunt) topology; (b) current-sampling shunt mixing (shunt-series) topology; (c) current-sampling series-mixing (series-series) topology; (d) voltage-sampling shunt-mixing (shunt-shunt) topology.

13

Review of Fundamentals

mix a voltage sample with the input (hence a series configuration at input) by sampling the output voltage (hence a shunt sampling case). Thus, this case is called “series-shunt” feedback. It may be worth noting that the cases come from the basic requirement for mixing identical types of signals, voltages, or currents. (Two ideal voltage sources can be mixed in series only, whereas two ideal current sources can be mixed in parallel only.) Although a complete treatment of the four different cases is beyond the scope of this chapter, Table 1.2 summarizes the four cases. For details, see Sedra and Smith [5]. Figure 1.7 indicates the use of a two-port h parameter feedback circuit with a voltage amplifier and derivation of the A circuit (a modified version of the basic amplifier) and the β circuit for the series-shunt feedback. In a practical design, one needs to simplify the case to have a unilateral β circuit, as shown in Figure 1.7b, after neglecting the parameter h21. Similarly, for the transconductance amplifier, Figure 1.8 indicates the derivation of the A circuit and β circuit where series-series feedback is used. The ideal structure of a series-shunt feedback amplifier is shown in Figure 1.9, consisting of a unilateral open-loop amplifier (the A circuit) and an ideal voltage sampling series mixing feedback network (the β circuit). The A circuit has an input resistance Ri, a voltage gain A, and an output resistance Ro. It is assumed that the source and load resistances have been included in the A circuit. Furthermore, note that the β circuit does not load the A circuit; that is, connecting the β circuit does not change the value of A, which is defined as Vo/Vi. Similarly, for the series-series feedback case, the ideal structure and the equivalent circuit are shown in Figure 1.10.

1.6 Loop Gain and the Stability Problem In Figure 1.4, the open-loop gain is a function of frequency, and the feedback factor β is dependent on frequency in a general case. This situation can be indicated by the generalized case for the feedback with a closed-loop transfer function Af (s) as in Equation 1.17:

A f ( s) =

A( s) . 1 + A( s)β( s)

(1.17)

If the amplifier is assumed to be a direct-coupled case with a constant DC gain of A0 with the poles and zeros occurring in the high-frequency band, the loop gain A(s)β(s) becomes a constant (A0β0) at low frequencies when the feedback factor β(s) at low frequencies reduces to a constant value (β0). This is a common situation we assume in design work for simplicity and convenience

Gain

Av = vo /vs

Ai = io /is

Gm = io /vs

Rm = vo /is

Basic Amplifier Type

Voltage amplifier

Current amplifier

Transconductance amplifier

Transresistance amplifier

Shunt-shunt

Series-series

Shunt-series

Series-shunt

Type of Feedback

y parameters

z parameters

g parameters

h parameters

Parameters Used for Two-Port Network-Based Analysis feedback network

〈〈 h12 feedback network

amplifier

〈〈 h21 basic

feedback network

feedback network

feedback network

〈〈 z12

feedback network

amplifier

〈〈 z21 basic

〈〈 g12

amplifier

〈〈 g 21 basic

y 21

feedback network

amplifier

y12 basic

feedback network

amplifier

〈〈 y 21 basic

〈〈 y12

(unilateral condition)

amplifier

z12 basic

z21

amplifier

g12 basic

g 21

(unilateral condition)

amplifier

h12 basic

h21

Assumptions for Simplifications

Ri/(1 + Aβ)

Ri*(1 + Aβ)

Ri/(1 + Aβ)

Ri*(1 + Aβ)

Input Impedance

RO/(1 + Aβ)

RO (1 + Aβ)

RO (1 + Aβ)

RO/(1 + Aβ)

Output Impedance

Modified Values

Different Amplifier Types, Feedback, and the Modification of Input and Output Impedances

Table 1.2

14 Electronic Circuit Design: From Concept to Implementation

15

Review of Fundamentals Rs +

Basic Amplifier

+ Vs –

RL

h11 1

+ + _

I1

Vo _

h12 V2

h21 I1

h22

2

_

Feedback Network (a)

A circuit

Rs Vs + –

_ V f

+ Vi + _

Basic Amplifier

h11

+ _

RL

+ Vo _

h22

+ h12 Vo

Vo _

(b)

β circuit

Figure 1.7 Derivation of the A circuit and β circuit for the series-shunt feedback amplifier: (a) the circuit in Figure 1.6a with the feedback represented by h parameters; (b) the circuit in Figure 1.7a, neglecting h21 and modifying the basic amplifier to form the A circuit.

and to achieve a negative feedback condition. However, at higher frequencies this need not be the case, and by substituting s = jω,

A f ( jω) =

A( jω) . 1 + A( jω)β( jω)

(1.18)

This clearly indicates that A(jω)β(jω) is a complex number L(jω) that can be represented by

L( jω) ≡ A( jω)β( jω) e jφ(ω) .

(1.19)

This indicates that at higher frequencies, loop gain L(jω) can be positive or negative, depending on the phase angle ϕ. To appreciate this fact, consider

16

Electronic Circuit Design: From Concept to Implementation RL

Rs

Io

Basic Amplifier

+ Vs – Rif

I1

z12I2

1

Io

z11

z22 + _

Rof

I2

z21I1

2

Feedback Network (a)

+ Vs –

– Vf +

A circuit

RL

Rs Basic Amplifier z11

Io z22 Io

z12Io

+ _

β circuit (b)

Figure 1.8 Derivation of the A circuit and the β circuit for a series-series feedback amplifier: (a) the circuit in Figure 1.6c with the feedback represented by z parameters; (b) the same circuit neglecting z21 and modifying the basic amplifier to form the A circuit.

the frequency at which ϕ(ω) becomes 180°. At this frequency, ω180, A(jω)β(jω), or the loop gain will be a negative real number. As per Equation 1.18, at this frequency the feedback will be positive. If at ω = ω180 the loop gain is less than unity, the closed-loop gain will be greater than the open-loop gain A(jω). Nevertheless, the feedback amplifier will be stable. In the special case where the loop gain becomes −1 at ω = ω180, the amplifier closed-loop gain will be infinite, with the practical situation of output rising to a very high value for no appreciable input signal, which represents the case of an oscillator with a frequency of oscillation equal to ω180. This is discussed further in Chapter 8. When the magnitude of the loop gain is greater than unity at ω180 it is not very obvious from Equation 1.18, but the circuit will oscillate with the amplitude growing gradually until the circuit nonlinearities limit the amplitude and the loop gain becomes unity.

17

Review of Fundamentals

S

Ro

Ii +

Vs

+ –

Vi – Vf +

Ri

AVi

+ _

_

A circuit O + Vo _ O´

S´

+ + –

βVo Vo _

β circuit Figure 1.9 The series-shunt feedback amplifier.

1.6.1 The Nyquist Plot Based on the preceding discussion, a formalized approach for testing the stability is the Nyquist plot, which is simply a polar plot of loop gain with frequency as a parameter. Figure 1.11 depicts a Nyquist plot where the radial distance is |Aβ| and the angle is the phase angle φ. The solid line indicates positive frequencies and the dotted line indicates negative frequencies, which forms a mirror image of the plot for the positive and negative ω. The Nyquist plot intersects the negative real axis at the frequency ω180, indicating that if the intersection occurs to the left of the point (−1,0), loop gain |Aβ| > 1, making the amplifier unstable. However, if the plot intersects the negative real axis to the right of (−1,0) the amplifier will be stable. In summary, if the Nyquist plot encircles the point (−1,0), the amplifier will be unstable, which is a simplified version of the Nyquist criterion. For the full theory, see Haykin [6]. 1.6.2 Poles and Zeros, S-Domain, and Bode Plots The frequency response of an amplifier can be analyzed by representing the gain as a function of the complex frequency s. In the s domain analysis, a capacitor of value C is represented by an equivalent impedance of 1/sC, and an inductance of value L is represented by Ls. Using common circuit analysis techniques, a voltage transfer function T(s) is derived as T(s) ≡ Vo(s)/Vi(s).

18

Electronic Circuit Design: From Concept to Implementation A circuit S

Ii

Io

O

+ Vs + _

Ri

Vi

AVi

_

– Vf +

Ro Io O´

S´

+ –

Io

+ βIo _

β circuit (a)

S

O

+ Rif

Vs

AfVs

Io

Rof

– O´

S´

(b)

Figure 1.10 The series-series feedback amplifier: (a) ideal structure; (b) equivalent circuit with modified input and output impedances.

Once this function is derived, by replacing s with jω we can evaluate its frequency behavior. In general, the transfer function T(s) in its own form (without substituting s = jω) can reveal many useful details about the stability of the circuit. Function T(s) can be expressed in many different forms, including

( s − z1 )( s − z2 )( s − z3 )…( s − zm ) ( s − p1 )( s − p2 )( s − p3 )…( s − pn )

(1.20)

am sm + am−1 sm−1 + am−2 sm−2 +…+ a0 . s n + bn−1 sn−1 + bn−2 sn−2 …+ b0

(1.21)

T ( s) = am

and

T ( s) =

19

Review of Fundamentals ω Negative and increasing in magnitude

Imaginary

ω=∞ ω = ω180

–1, 0

ω=0

ф

Real

|Aβ|

ω Positive and increasing Figure 1.11 The Nyquist plot of an unstable amplifier.

In Equation 1.20, z1, z2, … , zm are called transfer function zeros or transmission zeros, and p1, p2, … , pn are called transfer function poles or the natural modes of the network. The n of the transfer function is called the order of the network. The equivalent form of the transfer function in Equation 1.21, where coefficients a and b are real numbers, gives us the condition that the poles or zeros must occur as conjugate pairs; that is, if, for example, a pole or a zero occurs at 3 + 2j, there should be another pole or zero at 3 − 2j. A zero that is pure imaginary (±jωz) causes the transfer function to be exactly zero at ω = ωZ. For a system to be stable, its poles should lie in the left half of the s plane. A pair of complex-conjugate poles on the jω axis gives rise to sustained oscillations. Poles on the right-hand side of the s plane give rise to growing oscillations, which will be limited by nonlinearities in a practical system. Figure 1.12 indicates the three possibilities. From the closed-loop transfer function of Equation 1.17, we see that the poles of the feedback amplifier can be obtained by solving the equation

1 + A(s)β(s) = 0,

(1.22)

which is called the characteristic equation. This indicates that feedback to a system changes its poles. In an amplifier with an open-loop transfer function characterized by a single pole, the open-loop transfer function is

A( s) =

Ao . s 1+ ωp

(1.23)

20

Electronic Circuit Design: From Concept to Implementation jω

S Plane σ

Time (a)

jω

S Plane

σ

jω

Time (b)

S Plane σ

Time (c)

Figure 1.12 Relationship between pole location and transient response: (a) poles on the left-hand side of the imaginary axis; (b) poles on the right-hand side of the imaginary axis; (c) poles on the imaginary axis.

The closed-loop transfer function is given by

A f ( s) =

Ao /(1 + Aoβ) . 1 + s / ω p (1 + Aoβ)

(1.24)

Feedback moves the pole along the negative real axis to a frequency ωPf :

ωPf = ωP(1 + Aoβ)

(1.25)

This process is illustrated in Figure 1.13. With feedback, the new pole shifts to the left-hand side of the original pole. Figure 1.13b indicates its effect on the Bode plot, where at low frequencies the gain drops by an amount equal to 20log(1 + Aoβ) and the two curves coincide at high frequencies. Figure 1.13b indicates that applying feedback extends the bandwidth of the amplifier at the expense of loop gain. Because the pole of the closed-loop amplifier never enters the right-hand side of the s plane, the single-pole amplifier is unconditionally stable. 1.6.3 Bode Plots and Gain and Phase Margins From Section 1.6 we know that we can determine whether a feedback amplifier is or is not stable by examining its loop gain Aβ as a function of frequency.

21

Review of Fundamentals jω

dB 20 log (1 + Ao β)

ωp

S Plane

|A0|

0

|Af 0|

σ

|A| |Af |

ωpf ω (log scale)

ωp

ωpf = ωp (1 + Ao β) (a)

(b)

Figure 1.13 Effect of feedback on a single-pole amplifier: (a) pole location; (b) the frequency response (Bode magnitude plot).

Amplitude |Aβ|

dB

0

Phase Angle of Aβ

0°

ω1

ω1

Gain margin ω180 ω(log scale)

ω180

ω(log scale)

90° 180° 270°

Phase margin

360°

Figure 1.14 Bode plots for the loop gain illustrating the gain and phase margin.

The most effective and easiest way to do this is through the use of a Bode plot of Aβ, such as the one in Figure 1.14. This feedback amplifier will be stable, because at the frequency of the 180° phase shift, ω180, the magnitude of the loop gain is less than unity or negative in decibel (dB) terms. The difference between the value of |Aβ| at ω180 and unity is called the gain margin (usually expressed in decibels). This important parameter represents the amount by which the loop gain can be increased while stability is maintained. Feedback amplifiers are usually designed to have sufficient gain margins to allow

22

Electronic Circuit Design: From Concept to Implementation

for the inevitable changes in loop gain with physical parameters such as temperature, relative humidity, and the age of components. Another way to investigate stability and its degree is to examine the Bode plot at the frequency for which |Aβ| = 1, which is the location at which the magnitude plot crosses the 0-db line. If at this point the phase angle is less (in magnitude) than 180°, then the amplifier is stable. This situation is illustrated in Figure 1.14. The difference between the phase angle at this frequency and 180° is called the phase margin. In other words, if at the frequency of unity loop gain magnitude the phase lag is in excess of 180°, the amplifier will be unstable. An alternative and much simpler approach for investigating stability is to construct the Bode plots for the open-loop gain and, assuming that the β is independent of frequency, plot the horizontal line corresponding to 20log(1/β) as a horizontal straight line on the same amplitude plot, looking at the difference between the two curves. For details, see Chapter 8 and Appendix E of Sedra and Smith [5].

1.7 Amplifier Frequency Response As a review, let’s summarize a few important equivalent circuit models for bipolar junction transistors (BJTs) and metal oxide semiconductor field effect transistors (MOSFETs) and then briefly discuss the frequency response considerations of simple transistor amplifier circuits. Comprehensive discussions of this subject can be found in many textbooks, including Sedra and Smith [5]. It is important to note that for transistors we use different circuit models for large-signal and small-signal operations. Data sheet parameters clearly distinguish these two subsets. 1.7.1 BJT Equivalent Circuits The following sections review transistor models and equivalent circuits in low-frequency and high-frequency regions. Because there are many books and other publications that provide details on BJTs as well as FETs, the purpose of the section is to remind the reader of how and which models are useful for different applications. Sedra and Smith [5] is very comprehensive in the treatment of these devices within the discrete as well as the IC domains. Ayers [7] is a good source of information related to digital circuit implementations. An NPN BJT’s transfer characteristics can be simplified as shown in Figure 1.15. This iC-vBE characteristic can be summarized by the exponential relationship

iC = ISe vBE /VT ,

(1.26)

where vBE is the instantaneous base-emitter voltage; V T is the thermal voltage, which is approximately 25 mV at 25°C; and IS is the scale current, which

23

Review of Fundamentals ic

ic

I

0.5

0.7

T1 T2 T3 T1>T2>T3

vBE(V)

vBE(V)

(a)

(b)

Figure 1.15 Characteristics of the transistor: (a) iC-vBE plot; (b) temperature behavior indicating −2 mV/°C at a constant emitter current.

is dependent on the process parameters and the dimensions of the transistor. Similar exponential plots exist for iE-vBE and iB-vBE but with different scale currents, IS/α and IS/β, respectively. Because the constant of exponential characteristic, 1/V T, is high (≈ 40), the curve rises very sharply. For vBE smaller than 0.5 V, the current is negligibly small. Also, over most of the normal current range, vBE lies in the range of 0.6 to 0.8 V. For a PNP transistor circuit, the iC-vEB characteristic will look identical to that of Figure 1.15, except that vBE is replaced with vEB. In one of the most commonly used configurations, the common emitter configuration, iC is dependent on the value of the collector voltage; this is shown in Figure 1.16. In this common emitter mode, at low values of vCE, as the collector voltage goes below that of the base by more than 0.4 V, the collector-base junction becomes forward biased and the transistor leaves the active mode and becomes saturated. In the active region, iC-vCE curves are

iC + VBE _

+ VCE _

ic

Saturation region Active region

vBE = . . . vBE = . . . vBE = . . . vBE = . . .

–VA

vCE(V)

Figure 1.16 Common emitter characteristics indicating the practical behavior of transistors at different VBE values.

24

Electronic Circuit Design: From Concept to Implementation

B

iB

iC

C

+ vBE

B

iB

iC

C

+ DB (Is/β)

Is evBE/VT

DB (Is/β)

vBE

ro

–

βiB

– iE

iE E

ro

E (a)

(b)

Figure 1.17 Large signal model of the transistor (a) based on vBE as the control input; (b) based on iB as the control input.

reasonably straight and have nonzero slope values. In fact, when these are extrapolated, they meet at a point on the negative axis, at vCE = −VA. The voltage VA (a positive number) is a parameter of the particular BJT and is called the Early voltage. Typical values for VA are in the range of 50 to 100 V. The linear dependence of iC on vCE can be accounted for by assuming that IS remains constant and including a factor of 1 + (vCE/VA) in Equation 1.26. The nonzero slope of the iC-vCE straight lines indicates that the output resistance looking into the collector is not infinite. This gives us the relationship

ro = (VA + VCE)/IC.

(1.27)

It is rarely necessary to include the dependence of iC on vCE in DC bias calculations or analysis. However, the parameter ro can have a significant effect on the small signal gain of the transistor amplifiers. Based on the above characteristics, a large signal equivalent circuit model for the BJT is depicted in Figure 1.17. In this figure we have assumed that the transistor gain β is constant for all operational conditions of the transistor. However, this assumption is not accurate, and in practical design we usually define two different β values, namely the large signal β or DC β and the small signal β or AC β. Common emitter characteristics with base current as the control input are shown in Figure 1.18; compare this to the case of vBE used as the input control parameter in Figure 1.16. The operating point Q in the graph allows us to define these two parameters, βDC and βAC, respectively, as

βDC ≡ ICQ/IBQ βac ≡

∆iC ∆iB

vCE=constant

(1.28a) .

(1.28b)

25 ic

Saturation region

Review of Fundamentals

Active region

iB = IB1 iB = IB2

∆ic iC

IB

VCE

iB

iB = IBQ + ∆iB

ICQ

iB = IBQ

Q

iB = . . . 0

VCEQ

(a)

iB = 0 vCE(V) BVCEO

(b)

Figure 1.18 Common emitter characteristics showing the definition of hFE and hfe and also the primary breakdown limits at the vCE value of BVCEO. (a) Simplified circuit; (b) ic versus VCE characteristics with breakdown phenomenon at high VCE values. hFE (β) 400 T = 125°C 300

200

100

1 µA

T = 25°C

T = –55°C

10 µA 100 µA 1 mA 10 mA 100 mA

IC

Figure 1.19 Dependence of hFE for the DC collector current.

In data sheets, βDC is referred to as hFE and βAC is referred to as hfe, representing the two port h parameter equivalent circuits used. Figure 1.19 indicates the typical dependence of hFE on the quiescent DC over a scale of several orders. In Figure 1.18 another important limiting parameter, collector-emitter breakdown voltage BVCEO, is also indicated (on data sheets this parameter is sometimes referred to as sustaining voltage [LVCEO]). This is the limiting

26

Electronic Circuit Design: From Concept to Implementation

value at iB = 0 where the transistor collector current suddenly rises; under general operating conditions, reaching this collector-emitter voltage value should be avoided. 1.7.2 BJT Small Signal Operation and Models In most amplifier configurations, a transistor circuit is biased around a quiescent point, and we try to operate the amplifier around that point for better linearity. In such situations, instead of the large signal models shown in Figure 1.17, small signal models are used. The parameters based on the hybrid-π model are shown in Figure 1.20d and Figure 1.20e. Note that these parameters are based on a circuit with DC sources as bias voltage supplies, later reduced to the two cases separately for DC bias analysis and AC signal components only, as in Figure 1.20a to Figure 1.20c. Figure 1.20f shows linear operation of the transistor under small signal conditions. For the small signal models of the transistor, it can be shown that

gm = Ic/V T

(1.29a)



rπ = β/gm = V T/IB.

(1.29b)

A more in-depth analysis of these topics with other models can be found in Sedra and Smith [5]. 1.7.3 High-Frequency Models of the Transistors and the Frequency Response of Amplifiers Let us begin this review with the transfer function of a capacitively coupled amplifier. As indicated in Figure 1.21, the amplifier gain is almost constant over a wide frequency range called the midband. In this frequency range, all capacitances such as coupling, bypass, and device internal values are considered to have negligible effects. At the high-frequency end of the spectrum, the gain drops because of the effect of device internal capacitances, whereas the low-frequency end is mostly governed by the bypass and the coupling capacitances. The extent of the midband is defined by the two frequencies ωL and ωH, where the gain drops by 3 dB below the midband. The amplifier bandwidth (BW) is usually defined as

BW = ωH − ωL.

(1.30)

In many cases where ωL